AN1311 Single Cell Input Boost Converter Design Author: auto switching from a Pulse Skipping, or Pulse Frequency Modulation (PFM) mode to a continuous 500 kHz Fixed Frequency mode by using MCP1640/MCP1640C devices. For applications that cannot tolerate the low frequency Pulse Skipping mode or the output ripple voltage associated with it, the MCP1640B/D devices switch at a continuous fixed pulse width modulation frequency of 500 kHz. In addition to dual switching modes, the MCP1640/B/C/D family of devices offers two disable options. In the True Output Disconnect option (MCP1640/MCP1640B devices), the output of the synchronous boost converter is open and the typical diode path from input to output is removed, isolating the input from the output. In the Input Bypass option (MCP1640C/D devices), the input is connected to the output using the synchronous P-Channel switch. During this mode, the quiescent current draw from the battery is less than 1 µA typical. The Input Bypass mode provides voltage to power a load in deep sleep with the ability to boost the voltage up to the levels that are necessary for normal operation. Terry Cleveland Microchip Technology Inc. INTRODUCTION Currently, many portable battery-powered applications use multiple cell batteries for power. In some cases, the product form factor is driven by the size of the battery pack. This application note introduces and details design equations and trade-offs that facilitate the use of single cell input synchronous boost converters from the Microchip MCP1640/B/C/D family of devices. These single cell input boost converters enable startup from very low input voltage sources. The MCP1640/B/C/D converters will start from a 0.65 V source and operate down to 0.35 V, while boosting the output voltage from 2.0 V to 5.5 V. Two typical application schematics are shown in Figure 1. Efficiency is maximized over the entire load range by L1 4.7 µH VIN 0.9 V to 1.7 V VOUT 3.3 V @ 100 mA SW V OUT VIN Alkaline + CIN 4.7 µF 976 K VFB EN 562 K - GND COUT 10 µF L1 4.7 µH VIN 3.0 V to 4.2 V Li-Ion + CIN 4.7 µF - FIGURE 1: SW V OUTS VIN VOUTP EN VFB VOUT 5.0 V @ 200 mA 976 K COUT 10 µF 309 K PGND SGND Typical MCP1640 Applications. 2010 Microchip Technology Inc. DS01311A-page 1 AN1311 BOOST CONVERTER ANALYSIS Boost Converter Operation The Inductive Switch mode boost power converter is used to step up a lower voltage to a higher voltage. The boost topology requires an inductor, switch, diode, and output capacitor. To analyze the operation of a boost converter, it is assumed that the output voltage ripple is low or DC. In practice this assumption is normally valid for DC-DC converters. However, in many boost converters, the DC current flows from input to output through an inductor L1 and a diode. And, in typical applications, when the boost converter is turned off, this can drain the battery. In MCP1640/B/C/D devices, the diode is replaced with a P-Channel MOSFET that acts like a diode, i.e., it turns on to forward current from input to output and turns off to block reverse current from output to input. An internal switch blocks the forward diode path of the P-Channel while the converter is disabled. Figure 2 represents the basic components of a synchronous boost regulator. L1 VOUT VIN Q2 Q1 COUT Boost Converter FIGURE 2: Boost Converter Topology. SWITCH CLOSED At the beginning of the cycle, switch Q1 is turned ON. During this time, the output current is supplied by the output capacitor COUT, and magnetic field energy is stored in inductor L1. With Q1 ON, the inductor current ramps up at a constant rate of VIN (Input Voltage) divided by the inductance of L1. The diagram in Figure 3 represents the Switch Closed state. L1 VOUT SWITCH OPEN At the end of the Pulse Width Modulation (PWM) cycle, the boost switch Q1 turns off. The inductor current must—and will—continue to flow, finding a path through Q2. This current now supports the load, in addition to replenishing the current removed from COUT during the switch ON time. The diagram in Figure 4 represents the Switch Open state. L1 VOUT VIN Q2 COUT Boost Converter Q1 OFF FIGURE 4: Switch Q1 OFF. For steady state operation, the energy that is removed from COUT during the switch ON time must be replaced with exactly the same amount of energy during the switch OFF time. In addition to the charge-time balance on the output capacitor COUT, the inductor current ramp during the switch ON time must be exactly equal to the inductor current ramp during the switch OFF time to achieve steady state PWM switching. For steady state operation, the applied volt-time on the inductor must be balanced or equal in magnitude, and opposite in direction, for the switch ON and OFF time. This forms the basis for our first equation: EQUATION 1: INDUCTOR VOLT-TIME BALANCE V IN ton = V OUT – V IN t off Using the inductor volt-time balance and replacing the switch ON time with duty cycle D, and the switch OFF time with 1-D, the inductor volt-time balance can be used to derive the switch duty cycle D. EQUATION 2: DUTY CYCLE BALANCE D = V OUT –V IN V OUT VIN Q1 COUT Boost Converter/Q1 Closed FIGURE 3: DS01311A-page 2 Switch Q1 ON. 2010 Microchip Technology Inc. AN1311 Inductor Current Operating Modes CONTINUOUS INDUCTOR CURRENT MODE In the previous derivation, there are two inductor volt-time states. • State 1: VIN is applied across L1. • State 2: VOUT-VIN is applied across L1. For steady state operation, current must be flowing in L1 at all times. However, as the boost output current lowers, another state is entered. In this third state, the inductor current reaches zero. This adds another term to the volt-time balance equation. Figure 5 represents Continuous Inductor Current mode. VOUT - VIN VOUT VSW IIN IL DISCONTINUOUS INDUCTOR CURRENT MODE During Discontinuous Inductor Current mode, the inductor current reaches zero prior to the end of the cycle. This operating mode does not impact the regulation of the boost converter. Discontinuous mode is entered when the output power (VOUT * IOUT) is less than the amount of energy stored in the inductor multiplied by the switching frequency ((1/2*L*ILPK2)*FSW). As the load is reduced, the inductor current will eventually reach 0A. If the load is further reduced, the duty cycle must also be reduced to prevent overcharging the output capacitor or losing voltage regulation. To derive the duty cycle equation for Discontinuous mode, the same procedure (that was used for Continuous mode) applies. In the Discontinuous equation, there are three states, versus the two for Continuous mode. • State 1: switch is ON, the current is ramping in the inductor, and the voltage applied is +VIN. • State 2: switch is OFF, the current is ramping down, and inductor voltage is -(VOUT-VIN) • State 3: switch is OFF, the inductor current has reached zero, and the inductor voltage is zero. By adding the third state the duty cycle solution becomes more difficult; but it is solvable, through the use of two equations. Since the inductor current ramp up must be equal to the inductor current ramp down (see Figure 6), the following relationship can be derived: VIN VL EQUATION 3: VIN - VOUT V ID OUT INDUCTOR CURRENT BALANCE = V IN D1 + D2 D2 --------------------------- IOUT D1 1-D1 TS FIGURE 5: Waveforms. Continuous Inductor Current 2010 Microchip Technology Inc. DS01311A-page 3 AN1311 Figure 6 represents Discontinuous Inductor Current mode. VOUT-VIN VOUT VSW VIN IIN IL VIN 0V VL VIN- VOUT ID IOUT D1 D2 TS D3 D1 D2 D3 EQUATION 4: 1 1 I OUT = ----- --- I LPK D2 T S Ts 2 Substitute VIN/L* TON for ILPK to simplify. EQUATION 5: V 1 IN I OUT = --- ---------- D1 T S D2 2 L The derivation is reduced to two equations and two unknowns. Solving each equation for D2 and setting them equal to each other results in the following solution, after substituting VOUT/R for IOUT. Solving for VOUT results in two solutions. Disregarding the imaginary solution, and substituting VOUT and VIN back into the previous D2 equations, and solving for D1, results in the following discontinuous duty cycle equation: EQUATION 6: DISCONTINUOUS DUTY CYCLE 12 2 R T s V OUT L V OUT – V IN 1 LOAD D1 = --------------------------------- -------------------------------------------------------------------------------------------------------------------------------------R T V LOAD s IN TS FIGURE 6: Discontinuous Inductor Current Waveforms. For DC-DC converter analysis, the output energy is equal to the input energy, assuming efficiency is 100%. Using this relationship, the following equation can be written to determine the output current. The output current is equal to the average inductor current during the switch off time. DS01311A-page 4 2010 Microchip Technology Inc. AN1311 When the inductor current reaches zero at the same time the switch turns back on, it is defined as the boundary between continuous and discontinuous inductor current. To calculate the load for this boundary condition, use the energy stored per cycle and convert it to load current. Pulse Frequency Modulation (PFM) The MCP1640/MCP1640C devices can operate in a third mode, Pulse Frequency Modulation (PFM) mode. PFM mode is entered when the output current reduces below a predetermined threshold. In PFM mode, the inductor peak current is fixed at a value that is higher than required to keep the output in regulation. This pumps the output voltage up; pulsing stops when the output voltage reaches the maximum limit, and the device enters a low quiescent current state to minimize the current draw on the battery. Higher output voltage ripple is a result of the PFM mode. Figure 7 shows PFM mode waveforms versus Pulse-Width Modulation (PWM) mode waveforms for 1 mA load current. The MCP1640B/D devices do not enter PFM mode, and the peak inductor current continues to reduce with load while the devices operate in normal Discontinuous Inductor Current mode. Compared to PFM mode, the output ripple voltage is lower and the device switches at a constant frequency of 500 kHz. This is desirable for applications that have audio or low-frequency signals. The disadvantage of not entering PFM mode is the lower efficiency. Figure 8 compares PFM/PWM mode efficiency with PWM-only mode efficiency. V OUT = 3.3V 100 VIN = 2.5V 90 80 Efficiency (%) CONTINUOUS VS. DISCONTINUOUS BOUNDARY 70 VIN = 0.8V 60 VIN = 1.2V 50 40 30 PWM / PFM 20 PWM ONLY 10 0 0.01 0.1 1 10 100 1000 IOUT (mA) FIGURE 8: Operating Modes. Efficiency, PFM and PWM The P-Channel Synchronous rectifier switch turns off when the inductor current reaches zero, for all devices and modes of operation. This prevents current from flowing backwards from output to input, keeping the efficiency high. For ultra light loads, pulse skipping does occur when operating in PWM-only mode. The peak current in the inductor is low, keeping the ripple voltage low. Figure 9 graphs the current at which the MCP1640B/D devices begin to skip pulses versus the input voltage. PFM Mode 4.5 4 VOUT = 5.0V 3.5 VIN (V) 3 VOUT = 3.3V 2.5 2 VOUT = 2.0V 1.5 1 0.5 0 0 1 2 3 4 5 6 7 8 9 10 IOUT (mA) PWM Mode FIGURE 7: Operation. PFM Operation vs. PWM 2010 Microchip Technology Inc. FIGURE 9: Pulse Skipping Threshold Voltage vs. Load Current. DS01311A-page 5 AN1311 Peak current mode control compares the peak switch (or inductor current) with the output of the error amplifier. As the load demands change, the error amplifier (with integrated compensation) changes to set the proper peak current for voltage regulation. Peak Current Mode Control The MCP1640/B/C/D family of devices uses peak current mode control. This control method reduces the order of the power system to one versus two, when compared to voltage mode control. The device block diagram is represented in Figure 10. VOUT VIN Internal Bias Direction Control GND Gate Drive and Shutdown Control Logic Oscillator PWM /PFM Logic SOFT-START 0V ILIMIT + + - ISENSE Slope Compensation S - EN .3V IZERO + SW + - 1.21V FB + EA FIGURE 10: Peak Current Mode Control. For sudden changes in load, the peak current mode control provides a fast response. The response is a function of the inductor value and the output capacitor value. Since the compensation for the MCP1640/B/C/D family is integrated, there are limits on the range of inductance and output capacitance that can be used. For peak current mode control, applications that operate with over 50% duty cycle, slope compensation is necessary to maintain stability. Slope compensation is added to the current sense signal internally to the device. This also limits the variation in inductance that can be used. A peak current limit is set by limiting the height of the sensed switch current to a safe value. The MCP1640/B/C/D family of devices limits the peak current to 850 mA typically. DS01311A-page 6 FIGURE 11: Inductor Current Waveform, 850 mA Peak Limit. 2010 Microchip Technology Inc. AN1311 The range of the boost inductor and minimum output capacitor are limited. Table 1 provides some guidance for how much variation can be used. In most cases, a 4.7 µH inductor and 10 µF capacitor are recommended for boost inductance and output capacitance. APPLICATIONS AND CONSIDERATIONS Input capacitance should be a minimum of 4.7 µF. Additional capacitance should be added for applications that are located far from the battery, or source, and have high source impedance. For low input voltage and high output current applications, 10 µF is recommended. The MCP1640/B/C/D family of devices is capable of starting with a very low input voltage with a load applied. The low voltage startup begins with the P-Channel MOSFET turning on to charge the output voltage up to the input voltage. Once the output voltage is charged, the N-Channel begins to switch, pumping the output voltage up to approximately 1.6 V. At this voltage, the internal bias switches from the input to the output. Typically the device can start with 0.65 V applied to the input. Typical startup waveforms are shown in Figure 12. For very low load applications, smaller output capacitors can be used. The value depends on the input voltage, output voltage, and output current. TABLE 1: LIMITS ON BOOST INDUCTANCE AND OUTPUT CAPACITANCE VOUT LMIN LMAX CMIN 2.0 V 2.2 µH 4.7 µH 10 µF 3.3 V 4.7 µH 10 µH 10 µF 5.0 V 4.7 µH 15 µH 10 µF Low Voltage Startup EFFICIENCY AND PERFORMANCE Converter efficiency is highly dependent on the input and output voltage, and current conditions. The dominant loss for the MCP1640/B/C/D family is resistance, so lower input/output voltage efficiency is lower in efficiency than higher input/output voltage applications. Other factors that can impact efficiency are the losses in the inductor and capacitor, mostly the resistive losses of the inductor. Larger inductors result in lower resistance and higher efficiency, the trade-off being size and cost. QUIESCENT CURRENT, LEAKAGE CURRENT AND HOW IT RELATES TO BATTERY LIFE The MCP1640/B/C/D family of devices operate with very low quiescent current (IQ). The typical IQ for the devices, while operating in PFM mode, is 19 µA. For applications that have a low Sleep mode current, this can result in substantial average battery current. For some multi-cell or coin cell applications, a Bypass mode that uses the integrated P-Channel MOSFET to connect the input to the output can be used to provide bias power to the load. When regulated voltage is needed, the EN input pin is pulled high and the output is regulated to the desired voltage. In Shutdown mode, the bypass current consumption is less than 1 µA, extending battery life. The output true-disconnect option isolates the input from the output by reversing the integrated P-Channel MOSFET body diode. In Shutdown mode, the output voltage is 0V and the typical IQ is less than 1 µA. 2010 Microchip Technology Inc. FIGURE 12: Low Voltage Startup. Low Input Voltage High Output Current Operation While operating at low input voltage and high output current, the input current of a MCP1640/B/C/D device can reach its peak limit. The peak current is typically limited to 850 mA, but can be as low as 600 mA. The peak input current can be estimated by calculating the output power (VOUT * IOUT), dividing the product (output power) by the input voltage, and dividing the quotient by the estimated efficiency. The final result is the average input current. High Duty Cycle Operation While operating at low input voltage and high output voltage, the duty cycle of MCP1640/B/C/D devices can approach the maximum limit of 91% typical. For example, when operating at 0.9 V with a 5.0 V output, the calculated duty cycle ((VOUT-VIN)/VOUT) = 82%. When taking efficiency into account, the actual duty cycle can approach 90%. This results in some PWM jitter and even loss of output voltage regulation. A maximum duty cycle limit is necessary for any boost converter; practical limits from 90% to 92% allow for high step up ratios. DS01311A-page 7 AN1311 4.7 µF Output Capacitors Though 10 µF of output capacitance is recommended for most applications, 4.7 µF ceramic output capacitors can be used under certain restrictions. Converter stability and output voltage ripple will be affected by the reduction of output capacitance. STABILITY USING 4.7 μF OUTPUT CAPACITORS The MCP1640/B/C/D family of devices has peak current mode control with internal compensation and adaptive slope compensation to match the inductor down-slope. For 4.7 µH inductors and 10 µF capacitors, the devices offer high phase and gain margin over the entire input voltage, output voltage, and output current operating range. Figure 13 shows that the converter 0dB cross-over frequency is approximately 15 kHz with 60 degrees of phase margin and 15 dB of gain margin. When using a 4.7 µF output capacitor, the 0 dB crossover is pushed out to almost 30 kHz, providing a faster responding system. However, the phase margin is reduced to less than 40 degrees and the gain margin to approximately 10 dB. A phase margin of 40 degrees is considered marginal for stability; as the input voltage changes, the phase margin will continue to decrease to the point of instability. An unstable converter results in a low frequency AC content to the output ripple that can be in the audible frequency range. While operating in Discontinuous Inductor Current mode, the converter stability is changed, and the order of the system is reduced by one, resulting in an increase in phase margin. A bode plot of the converter while operating in Discontinuous mode is shown in Figure 15. The 0 dB crossover is approximately 28 kHz, the phase margin is approximately 60 degrees and the gain margin is high—greater than 20 dB. As shown, the converter is stable while operating in the Discontinuous mode. 80 60 60 40 40 20 20 0 0 VIN = 1.2V; VOUT = 3.3V, IOUT = 75 mA, L = 4.7µH, COUT = 10µF -20 -20 -40 -40 -60 -60 -80 -80 1000000 10 100 1000 10000 100000 GAIN (dB) 80 PHASE (DEGREES) GAIN (DB) 100 100 100 80 80 60 60 40 40 20 20 0 0 VIN = 1.2V; VOUT = 3.3V, IOUT = 50 mA, L = 4.7 µH, COUT = 4.7µF -20 -20 -40 -40 -60 -60 -80 10 100 1000 10000 100000 PHASE (DEGREES) BOOST PEAK CURRENT MODE BODE PLOT BOOST PEAK CURRENT MODE BODE PLOT 100 -80 1000000 FREQUENCY (HZ) FREQUENCY (HZ) FIGURE 13: Bode Plot 4.7 µH, 10 µF Output Capacitor Continuous Current Mode. FIGURE 15: Bode Plot 4.7 µH, 4.7 µF Output Capacitor Discontinuous Current Mode. Figure 14 shows the system bode plot for the same conditions as Figure 13, with the output capacitor changed to 4.7 µF. In summary, to reduce the output capacitor to 4.7 µF, the converter must be operating in Discontinuous Inductor Current mode, which limits the maximum output current. Table 2 can be used as a guide: TABLE 2: 100 80 80 60 60 40 40 20 20 0 0 VIN = 1.2V; VOUT = 3.3V, IOUT = 75 mA, L = 4.7µH, COUT = 4.7µF -20 -20 -40 -40 -60 -60 -80 10 100 1000 10000 100000 PHASE (DEGREES) GAIN (DB) BOOST PEAK CURRENT MODE BODE PLOT 100 -80 1000000 MAX IOUT FOR DISCONTINUOUS MODE 1 Cell Input VIN = 0.9 V to 1.6 V 2 Cell Input VIN = 1.8 V to 3.2 V 3.3 V Input 2.0 V 3.3 V 5.0 V IOUT< 25 mA IOUT< 35 mA IOUT< 50 mA IOUT< 15 mA IOUT< 80 mA IOUT< 150 mA FREQUENCY (HZ) FIGURE 14: Bode Plot 4.7 µH, 4.7 µF Output Capacitor Continuous Current Mode. DS01311A-page 8 2010 Microchip Technology Inc. AN1311 Sub 2V Output Applications CONCLUSION The MCP1640/B/C/D family of devices operates from an internal voltage that selects the maximum voltage between VIN and VOUT. During startup, the maximum voltage is VIN, While up and running, the maximum voltage is VOUT. For a single cell input, 1.8 V output applications, it is recommended that the inductor is changed from 4.7 µH to 2.2 µH and the output capacitor is changed to 20 µF. For single cell inputs, the output current range for 1.8 V VOUT applications is limited to 100 mA for operation down to 0.9 V. Figure 16 represents the device efficiency while operating with a 1.8 V output. The MCP1640/B/C/D family of devices enables operation from a single cell input, delivers high efficiency, is small in size, and provides excellent dynamic performance. Like most DC-DC converters, the details of topology operation can be understood by balancing the volt-time on the inductor (or charge-time on the capacitor). Integrated compensation (error amplifier and slope) make stabilizing the DC-DC converter straight forward while using the standard 4.7 µH inductor and 10 µF output capacitor. Under limited output current and input voltage range, the inductor and capacitor values can be changed to further reduce solution size, cost, and operating range. VOUT = 1.8V, L = 2.2 µH, COUT = 22 µF 100 VIN = 1.6V 90 Efficiency (%) 80 VIN = 1.2V 70 60 VIN = 0.9V 50 40 30 20 10 0 0.01 0.1 1 10 100 IOUT (mA) FIGURE 16: 1.8V Output Efficiency. PFM / PWM Threshold Current (mA) For 1.8 V output applications, the PFM/PWM current threshold will vary as a result of lower internal bias voltage and lower internal gate drive voltage. Figure 17 represents the PWM/PFM mode threshold current plotted versus input voltage. 25 VOUT = 1.8V, L = 2.2 µH, COUT = 22 µF 20 15 PWM Mode 10 5 PFM Mode 0 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5 1.6 Input Voltage (V) FIGURE 17: 1.8V Output PFM/PWM Threshold Current. Due to rising threshold voltages at cold temperatures, it is recommend that the MCP1640/B/C/D minimum output voltage is 1.8 V for ambient temperatures greater than 0°C. For output currents less than 40 mA, a 3.3 µH inductor and a 10 µF output capacitor can be used when operating from a single cell alkaline input. 2010 Microchip Technology Inc. DS01311A-page 9 AN1311 NOTES: DS01311A-page 10 2010 Microchip Technology Inc. Note the following details of the code protection feature on Microchip devices: • Microchip products meet the specification contained in their particular Microchip Data Sheet. • Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions. • There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property. • Microchip is willing to work with the customer who is concerned about the integrity of their code. • Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.” Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act. Information contained in this publication regarding device applications and the like is provided only for your convenience and may be superseded by updates. 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Trademarks The Microchip name and logo, the Microchip logo, dsPIC, KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART, PIC32 logo, rfPIC and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. FilterLab, Hampshire, HI-TECH C, Linear Active Thermistor, MXDEV, MXLAB, SEEVAL and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A. Analog-for-the-Digital Age, Application Maestro, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN, ECONOMONITOR, FanSense, HI-TIDE, In-Circuit Serial Programming, ICSP, Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB, MPLINK, mTouch, Octopus, Omniscient Code Generation, PICC, PICC-18, PICDEM, PICDEM.net, PICkit, PICtail, REAL ICE, rfLAB, Select Mode, Total Endurance, TSHARC, UniWinDriver, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries. SQTP is a service mark of Microchip Technology Incorporated in the U.S.A. All other trademarks mentioned herein are property of their respective companies. © 2010, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved. Printed on recycled paper. ISBN: 978-1-60932-035-5 Microchip received ISO/TS-16949:2002 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified. 2010 Microchip Technology Inc. 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