AN1311

AN1311
Single Cell Input Boost Converter Design
Author:
auto switching from a Pulse Skipping, or Pulse
Frequency Modulation (PFM) mode to a continuous
500 kHz Fixed Frequency mode by using
MCP1640/MCP1640C devices. For applications that
cannot tolerate the low frequency Pulse Skipping mode
or the output ripple voltage associated with it, the
MCP1640B/D devices switch at a continuous fixed
pulse width modulation frequency of 500 kHz. In
addition to dual switching modes, the MCP1640/B/C/D
family of devices offers two disable options. In the True
Output Disconnect option (MCP1640/MCP1640B
devices), the output of the synchronous boost
converter is open and the typical diode path from input
to output is removed, isolating the input from the
output. In the Input Bypass option (MCP1640C/D
devices), the input is connected to the output using the
synchronous P-Channel switch. During this mode, the
quiescent current draw from the battery is less than
1 µA typical. The Input Bypass mode provides voltage
to power a load in deep sleep with the ability to boost
the voltage up to the levels that are necessary for
normal operation.
Terry Cleveland
Microchip Technology Inc.
INTRODUCTION
Currently, many portable battery-powered applications
use multiple cell batteries for power. In some cases, the
product form factor is driven by the size of the battery
pack.
This application note introduces and details design
equations and trade-offs that facilitate the use of single
cell input synchronous boost converters from the
Microchip MCP1640/B/C/D family of devices.
These single cell input boost converters enable startup
from very low input voltage sources. The
MCP1640/B/C/D converters will start from a 0.65 V
source and operate down to 0.35 V, while boosting the
output voltage from 2.0 V to 5.5 V. Two typical
application schematics are shown in Figure 1.
Efficiency is maximized over the entire load range by
L1
4.7 µH
VIN
0.9 V to 1.7 V
VOUT
3.3 V @ 100 mA
SW V
OUT
VIN
Alkaline
+
CIN
4.7 µF
976 K
VFB
EN
562 K
-
GND
COUT
10 µF
L1
4.7 µH
VIN
3.0 V to 4.2 V
Li-Ion
+
CIN
4.7 µF
-
FIGURE 1:
SW V
OUTS
VIN
VOUTP
EN
VFB
VOUT
5.0 V @ 200 mA
976 K
COUT
10 µF
309 K
PGND SGND
Typical MCP1640 Applications.
 2010 Microchip Technology Inc.
DS01311A-page 1
AN1311
BOOST CONVERTER ANALYSIS
Boost Converter Operation
The Inductive Switch mode boost power converter is
used to step up a lower voltage to a higher voltage. The
boost topology requires an inductor, switch, diode, and
output capacitor. To analyze the operation of a boost
converter, it is assumed that the output voltage ripple is
low or DC. In practice this assumption is normally valid
for DC-DC converters.
However, in many boost converters, the DC current
flows from input to output through an inductor L1 and a
diode. And, in typical applications, when the boost converter is turned off, this can drain the battery.
In MCP1640/B/C/D devices, the diode is replaced with
a P-Channel MOSFET that acts like a diode, i.e., it
turns on to forward current from input to output and
turns off to block reverse current from output to input.
An internal switch blocks the forward diode path of the
P-Channel while the converter is disabled. Figure 2
represents the basic components of a synchronous
boost regulator.
L1
VOUT
VIN
Q2
Q1
COUT
Boost Converter
FIGURE 2:
Boost Converter Topology.
SWITCH CLOSED
At the beginning of the cycle, switch Q1 is turned ON.
During this time, the output current is supplied by the
output capacitor COUT, and magnetic field energy is
stored in inductor L1. With Q1 ON, the inductor current
ramps up at a constant rate of VIN (Input Voltage)
divided by the inductance of L1. The diagram in
Figure 3 represents the Switch Closed state.
L1
VOUT
SWITCH OPEN
At the end of the Pulse Width Modulation (PWM) cycle,
the boost switch Q1 turns off. The inductor current
must—and will—continue to flow, finding a path
through Q2. This current now supports the load, in
addition to replenishing the current removed from COUT
during the switch ON time. The diagram in Figure 4
represents the Switch Open state.
L1
VOUT
VIN
Q2
COUT
Boost Converter Q1 OFF
FIGURE 4:
Switch Q1 OFF.
For steady state operation, the energy that is removed
from COUT during the switch ON time must be replaced
with exactly the same amount of energy during the
switch OFF time. In addition to the charge-time balance
on the output capacitor COUT, the inductor current ramp
during the switch ON time must be exactly equal to the
inductor current ramp during the switch OFF time to
achieve steady state PWM switching. For steady state
operation, the applied volt-time on the inductor must be
balanced or equal in magnitude, and opposite in
direction, for the switch ON and OFF time. This forms
the basis for our first equation:
EQUATION 1:
INDUCTOR VOLT-TIME
BALANCE
V IN  ton =  V OUT – V IN   t off
Using the inductor volt-time balance and replacing the
switch ON time with duty cycle D, and the switch OFF
time with 1-D, the inductor volt-time balance can be
used to derive the switch duty cycle D.
EQUATION 2:
DUTY CYCLE BALANCE
D = V
OUT
–V
IN
V
OUT
VIN
Q1
COUT
Boost Converter/Q1 Closed
FIGURE 3:
DS01311A-page 2
Switch Q1 ON.
 2010 Microchip Technology Inc.
AN1311
Inductor Current Operating Modes
CONTINUOUS INDUCTOR CURRENT MODE
In the previous derivation, there are two inductor
volt-time states.
• State 1: VIN is applied across L1.
• State 2: VOUT-VIN is applied across L1.
For steady state operation, current must be flowing in
L1 at all times.
However, as the boost output current lowers, another
state is entered. In this third state, the inductor current
reaches zero. This adds another term to the volt-time
balance equation.
Figure 5 represents Continuous Inductor Current
mode.
VOUT - VIN
VOUT
VSW
IIN
IL
DISCONTINUOUS INDUCTOR CURRENT
MODE
During Discontinuous Inductor Current mode, the
inductor current reaches zero prior to the end of the
cycle. This operating mode does not impact the
regulation of the boost converter.
Discontinuous mode is entered when the output power
(VOUT * IOUT) is less than the amount of energy stored
in the inductor multiplied by the switching frequency
((1/2*L*ILPK2)*FSW). As the load is reduced, the
inductor current will eventually reach 0A. If the load is
further reduced, the duty cycle must also be reduced to
prevent overcharging the output capacitor or losing
voltage regulation.
To derive the duty cycle equation for Discontinuous
mode, the same procedure (that was used for
Continuous mode) applies. In the Discontinuous
equation, there are three states, versus the two for
Continuous mode.
• State 1: switch is ON, the current is ramping in the
inductor, and the voltage applied is +VIN.
• State 2: switch is OFF, the current is ramping
down, and inductor voltage is -(VOUT-VIN)
• State 3: switch is OFF, the inductor current has
reached zero, and the inductor voltage is zero.
By adding the third state the duty cycle solution
becomes more difficult; but it is solvable, through the
use of two equations.
Since the inductor current ramp up must be equal to the
inductor current ramp down (see Figure 6), the
following relationship can be derived:
VIN
VL
EQUATION 3:
VIN - VOUT
V
ID
OUT
INDUCTOR CURRENT
BALANCE
= V
IN
 D1 + D2 
D2
 ---------------------------
IOUT
D1
1-D1
TS
FIGURE 5:
Waveforms.
Continuous Inductor Current
 2010 Microchip Technology Inc.
DS01311A-page 3
AN1311
Figure 6 represents Discontinuous Inductor Current
mode.
VOUT-VIN
VOUT
VSW
VIN
IIN
IL
VIN
0V
VL
VIN- VOUT
ID
IOUT
D1
D2
TS
D3 D1
D2
D3
EQUATION 4:
1
1
I OUT = -----   ---  I LPK  D2  T S
Ts
2
Substitute VIN/L* TON for ILPK to simplify.
EQUATION 5:
V
1
IN
I OUT = ---   ----------  D1  T S  D2
2
L
The derivation is reduced to two equations and two
unknowns. Solving each equation for D2 and setting
them equal to each other results in the following
solution, after substituting VOUT/R for IOUT.
Solving for VOUT results in two solutions. Disregarding
the imaginary solution, and substituting VOUT and VIN
back into the previous D2 equations, and solving for
D1, results in the following discontinuous duty cycle
equation:
EQUATION 6:
DISCONTINUOUS DUTY
CYCLE
12
2  R
 T s  V OUT  L   V OUT – V IN  
1
LOAD
D1 = ---------------------------------  -------------------------------------------------------------------------------------------------------------------------------------R
T

V
LOAD
s
IN
TS
FIGURE 6:
Discontinuous Inductor
Current Waveforms.
For DC-DC converter analysis, the output energy is
equal to the input energy, assuming efficiency is 100%.
Using this relationship, the following equation can be
written to determine the output current. The output
current is equal to the average inductor current during
the switch off time.
DS01311A-page 4
 2010 Microchip Technology Inc.
AN1311
When the inductor current reaches zero at the same
time the switch turns back on, it is defined as the
boundary between continuous and discontinuous
inductor current. To calculate the load for this boundary
condition, use the energy stored per cycle and convert
it to load current.
Pulse Frequency Modulation (PFM)
The MCP1640/MCP1640C devices can operate in a
third mode, Pulse Frequency Modulation (PFM) mode.
PFM mode is entered when the output current reduces
below a predetermined threshold. In PFM mode, the
inductor peak current is fixed at a value that is higher
than required to keep the output in regulation. This
pumps the output voltage up; pulsing stops when the
output voltage reaches the maximum limit, and the
device enters a low quiescent current state to minimize
the current draw on the battery. Higher output voltage
ripple is a result of the PFM mode. Figure 7 shows PFM
mode waveforms versus Pulse-Width Modulation
(PWM) mode waveforms for 1 mA load current.
The MCP1640B/D devices do not enter PFM mode,
and the peak inductor current continues to reduce with
load while the devices operate in normal Discontinuous
Inductor Current mode. Compared to PFM mode, the
output ripple voltage is lower and the device switches
at a constant frequency of 500 kHz. This is desirable
for applications that have audio or low-frequency signals. The disadvantage of not entering PFM mode is
the lower efficiency. Figure 8 compares PFM/PWM
mode efficiency with PWM-only mode efficiency.
V OUT = 3.3V
100
VIN = 2.5V
90
80
Efficiency (%)
CONTINUOUS VS. DISCONTINUOUS
BOUNDARY
70
VIN = 0.8V
60
VIN = 1.2V
50
40
30
PWM / PFM
20
PWM ONLY
10
0
0.01
0.1
1
10
100
1000
IOUT (mA)
FIGURE 8:
Operating Modes.
Efficiency, PFM and PWM
The P-Channel Synchronous rectifier switch turns off
when the inductor current reaches zero, for all devices
and modes of operation. This prevents current from
flowing backwards from output to input, keeping the
efficiency high. For ultra light loads, pulse skipping
does occur when operating in PWM-only mode. The
peak current in the inductor is low, keeping the ripple
voltage low. Figure 9 graphs the current at which the
MCP1640B/D devices begin to skip pulses versus the
input voltage.
PFM Mode
4.5
4
VOUT = 5.0V
3.5
VIN (V)
3
VOUT = 3.3V
2.5
2
VOUT = 2.0V
1.5
1
0.5
0
0
1
2
3
4
5
6
7
8
9
10
IOUT (mA)
PWM Mode
FIGURE 7:
Operation.
PFM Operation vs. PWM
 2010 Microchip Technology Inc.
FIGURE 9:
Pulse Skipping Threshold
Voltage vs. Load Current.
DS01311A-page 5
AN1311
Peak current mode control compares the peak switch
(or inductor current) with the output of the error
amplifier. As the load demands change, the error
amplifier (with integrated compensation) changes to
set the proper peak current for voltage regulation.
Peak Current Mode Control
The MCP1640/B/C/D family of devices uses peak current mode control. This control method reduces the
order of the power system to one versus two, when
compared to voltage mode control. The device block
diagram is represented in Figure 10.
VOUT
VIN
Internal
Bias
Direction
Control
GND
Gate Drive
and
Shutdown
Control
Logic
Oscillator
PWM /PFM
Logic
SOFT-START
0V
ILIMIT
+
+
-
ISENSE
Slope
Compensation
S
-
EN
.3V
IZERO
+
SW
+
-
1.21V
FB
+
EA
FIGURE 10:
Peak Current Mode Control.
For sudden changes in load, the peak current mode
control provides a fast response. The response is a
function of the inductor value and the output capacitor
value. Since the compensation for the MCP1640/B/C/D
family is integrated, there are limits on the range of
inductance and output capacitance that can be used.
For peak current mode control, applications that operate with over 50% duty cycle, slope compensation is
necessary to maintain stability. Slope compensation is
added to the current sense signal internally to the
device. This also limits the variation in inductance that
can be used. A peak current limit is set by limiting the
height of the sensed switch current to a safe value. The
MCP1640/B/C/D family of devices limits the peak
current to 850 mA typically.
DS01311A-page 6
FIGURE 11:
Inductor Current Waveform,
850 mA Peak Limit.
 2010 Microchip Technology Inc.
AN1311
The range of the boost inductor and minimum output
capacitor are limited. Table 1 provides some guidance
for how much variation can be used. In most cases, a
4.7 µH inductor and 10 µF capacitor are recommended
for boost inductance and output capacitance.
APPLICATIONS AND
CONSIDERATIONS
Input capacitance should be a minimum of 4.7 µF.
Additional capacitance should be added for
applications that are located far from the battery, or
source, and have high source impedance. For low input
voltage and high output current applications, 10 µF is
recommended.
The MCP1640/B/C/D family of devices is capable of
starting with a very low input voltage with a load
applied. The low voltage startup begins with the
P-Channel MOSFET turning on to charge the output
voltage up to the input voltage. Once the output voltage
is charged, the N-Channel begins to switch, pumping
the output voltage up to approximately 1.6 V. At this
voltage, the internal bias switches from the input to the
output. Typically the device can start with 0.65 V
applied to the input. Typical startup waveforms are
shown in Figure 12.
For very low load applications, smaller output
capacitors can be used. The value depends on the
input voltage, output voltage, and output current.
TABLE 1:
LIMITS ON BOOST
INDUCTANCE AND OUTPUT
CAPACITANCE
VOUT
LMIN
LMAX
CMIN
2.0 V
2.2 µH
4.7 µH
10 µF
3.3 V
4.7 µH
10 µH
10 µF
5.0 V
4.7 µH
15 µH
10 µF
Low Voltage Startup
EFFICIENCY AND PERFORMANCE
Converter efficiency is highly dependent on the input
and output voltage, and current conditions. The
dominant loss for the MCP1640/B/C/D family is
resistance, so lower input/output voltage efficiency is
lower in efficiency than higher input/output voltage
applications. Other factors that can impact efficiency
are the losses in the inductor and capacitor, mostly the
resistive losses of the inductor. Larger inductors result
in lower resistance and higher efficiency, the trade-off
being size and cost.
QUIESCENT CURRENT, LEAKAGE CURRENT
AND HOW IT RELATES TO BATTERY LIFE
The MCP1640/B/C/D family of devices operate with
very low quiescent current (IQ). The typical IQ for the
devices, while operating in PFM mode, is 19 µA. For
applications that have a low Sleep mode current, this
can result in substantial average battery current. For
some multi-cell or coin cell applications, a Bypass
mode that uses the integrated P-Channel MOSFET to
connect the input to the output can be used to provide
bias power to the load. When regulated voltage is
needed, the EN input pin is pulled high and the output
is regulated to the desired voltage. In Shutdown mode,
the bypass current consumption is less than 1 µA,
extending battery life. The output true-disconnect
option isolates the input from the output by reversing
the integrated P-Channel MOSFET body diode. In
Shutdown mode, the output voltage is 0V and the
typical IQ is less than 1 µA.
 2010 Microchip Technology Inc.
FIGURE 12:
Low Voltage Startup.
Low Input Voltage High Output Current
Operation
While operating at low input voltage and high output
current, the input current of a MCP1640/B/C/D device
can reach its peak limit. The peak current is typically
limited to 850 mA, but can be as low as 600 mA. The
peak input current can be estimated by calculating the
output power (VOUT * IOUT), dividing the product (output power) by the input voltage, and dividing the quotient by the estimated efficiency. The final result is the
average input current.
High Duty Cycle Operation
While operating at low input voltage and high output
voltage, the duty cycle of MCP1640/B/C/D devices can
approach the maximum limit of 91% typical. For example, when operating at 0.9 V with a 5.0 V output, the
calculated duty cycle ((VOUT-VIN)/VOUT) = 82%. When
taking efficiency into account, the actual duty cycle can
approach 90%. This results in some PWM jitter and
even loss of output voltage regulation. A maximum duty
cycle limit is necessary for any boost converter;
practical limits from 90% to 92% allow for high step up
ratios.
DS01311A-page 7
AN1311
4.7 µF Output Capacitors
Though 10 µF of output capacitance is recommended
for most applications, 4.7 µF ceramic output capacitors
can be used under certain restrictions. Converter
stability and output voltage ripple will be affected by the
reduction of output capacitance.
STABILITY USING 4.7 μF OUTPUT
CAPACITORS
The MCP1640/B/C/D family of devices has peak
current mode control with internal compensation and
adaptive slope compensation to match the inductor
down-slope. For 4.7 µH inductors and 10 µF
capacitors, the devices offer high phase and gain
margin over the entire input voltage, output voltage,
and output current operating range.
Figure 13 shows that the converter 0dB cross-over
frequency is approximately 15 kHz with 60 degrees of
phase margin and 15 dB of gain margin.
When using a 4.7 µF output capacitor, the 0 dB crossover is pushed out to almost 30 kHz, providing a faster
responding system. However, the phase margin is
reduced to less than 40 degrees and the gain margin to
approximately 10 dB. A phase margin of 40 degrees is
considered marginal for stability; as the input voltage
changes, the phase margin will continue to decrease to
the point of instability. An unstable converter results in
a low frequency AC content to the output ripple that can
be in the audible frequency range.
While operating in Discontinuous Inductor Current
mode, the converter stability is changed, and the order
of the system is reduced by one, resulting in an
increase in phase margin. A bode plot of the converter
while operating in Discontinuous mode is shown in
Figure 15. The 0 dB crossover is approximately
28 kHz, the phase margin is approximately 60 degrees
and the gain margin is high—greater than 20 dB. As
shown, the converter is stable while operating in the
Discontinuous mode.
80
60
60
40
40
20
20
0
0
VIN = 1.2V; VOUT = 3.3V,
IOUT = 75 mA, L = 4.7µH,
COUT = 10µF
-20
-20
-40
-40
-60
-60
-80
-80
1000000
10
100
1000
10000
100000
GAIN (dB)
80
PHASE (DEGREES)
GAIN (DB)
100
100
100
80
80
60
60
40
40
20
20
0
0
VIN = 1.2V; VOUT = 3.3V,
IOUT = 50 mA, L = 4.7 µH,
COUT = 4.7µF
-20
-20
-40
-40
-60
-60
-80
10
100
1000
10000
100000
PHASE (DEGREES)
BOOST PEAK CURRENT MODE BODE PLOT
BOOST PEAK CURRENT MODE BODE PLOT
100
-80
1000000
FREQUENCY (HZ)
FREQUENCY (HZ)
FIGURE 13:
Bode Plot 4.7 µH, 10 µF
Output Capacitor Continuous Current Mode.
FIGURE 15:
Bode Plot 4.7 µH, 4.7 µF
Output Capacitor Discontinuous Current Mode.
Figure 14 shows the system bode plot for the same
conditions as Figure 13, with the output capacitor
changed to 4.7 µF.
In summary, to reduce the output capacitor to 4.7 µF,
the converter must be operating in Discontinuous
Inductor Current mode, which limits the maximum output current. Table 2 can be used as a guide:
TABLE 2:
100
80
80
60
60
40
40
20
20
0
0
VIN = 1.2V; VOUT = 3.3V,
IOUT = 75 mA, L = 4.7µH,
COUT = 4.7µF
-20
-20
-40
-40
-60
-60
-80
10
100
1000
10000
100000
PHASE (DEGREES)
GAIN (DB)
BOOST PEAK CURRENT MODE BODE PLOT
100
-80
1000000
MAX IOUT FOR
DISCONTINUOUS MODE
1 Cell Input
VIN = 0.9 V to 1.6 V
2 Cell Input
VIN = 1.8 V to 3.2 V
3.3 V Input
2.0 V
3.3 V
5.0 V
IOUT<
25 mA
IOUT<
35 mA
IOUT<
50 mA
IOUT<
15 mA
IOUT<
80 mA
IOUT<
150 mA
FREQUENCY (HZ)
FIGURE 14:
Bode Plot 4.7 µH, 4.7 µF
Output Capacitor Continuous Current Mode.
DS01311A-page 8
 2010 Microchip Technology Inc.
AN1311
Sub 2V Output Applications
CONCLUSION
The MCP1640/B/C/D family of devices operates from
an internal voltage that selects the maximum voltage
between VIN and VOUT. During startup, the maximum
voltage is VIN, While up and running, the maximum
voltage is VOUT. For a single cell input, 1.8 V output
applications, it is recommended that the inductor is
changed from 4.7 µH to 2.2 µH and the output capacitor is changed to 20 µF. For single cell inputs, the output current range for 1.8 V VOUT applications is limited
to 100 mA for operation down to 0.9 V. Figure 16
represents the device efficiency while operating with a
1.8 V output.
The MCP1640/B/C/D family of devices enables operation from a single cell input, delivers high efficiency, is
small in size, and provides excellent dynamic performance. Like most DC-DC converters, the details of
topology operation can be understood by balancing the
volt-time on the inductor (or charge-time on the capacitor). Integrated compensation (error amplifier and
slope) make stabilizing the DC-DC converter straight
forward while using the standard 4.7 µH inductor and
10 µF output capacitor. Under limited output current
and input voltage range, the inductor and capacitor values can be changed to further reduce solution size,
cost, and operating range.
VOUT = 1.8V, L = 2.2 µH, COUT = 22 µF
100
VIN = 1.6V
90
Efficiency (%)
80
VIN = 1.2V
70
60
VIN = 0.9V
50
40
30
20
10
0
0.01
0.1
1
10
100
IOUT (mA)
FIGURE 16:
1.8V Output Efficiency.
PFM / PWM Threshold Current
(mA)
For 1.8 V output applications, the PFM/PWM current
threshold will vary as a result of lower internal bias voltage and lower internal gate drive voltage. Figure 17
represents the PWM/PFM mode threshold current
plotted versus input voltage.
25
VOUT = 1.8V, L = 2.2 µH, COUT = 22 µF
20
15
PWM Mode
10
5
PFM Mode
0
0.8
0.9
1
1.1
1.2
1.3
1.4
1.5
1.6
Input Voltage (V)
FIGURE 17:
1.8V Output PFM/PWM
Threshold Current.
Due to rising threshold voltages at cold temperatures,
it is recommend that the MCP1640/B/C/D minimum
output voltage is 1.8 V for ambient temperatures
greater than 0°C. For output currents less than 40 mA,
a 3.3 µH inductor and a 10 µF output capacitor can be
used when operating from a single cell alkaline input.
 2010 Microchip Technology Inc.
DS01311A-page 9
AN1311
NOTES:
DS01311A-page 10
 2010 Microchip Technology Inc.
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Trademarks
The Microchip name and logo, the Microchip logo, dsPIC,
KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART,
PIC32 logo, rfPIC and UNI/O are registered trademarks of
Microchip Technology Incorporated in the U.S.A. and other
countries.
FilterLab, Hampshire, HI-TECH C, Linear Active Thermistor,
MXDEV, MXLAB, SEEVAL and The Embedded Control
Solutions Company are registered trademarks of Microchip
Technology Incorporated in the U.S.A.
Analog-for-the-Digital Age, Application Maestro, CodeGuard,
dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN,
ECONOMONITOR, FanSense, HI-TIDE, In-Circuit Serial
Programming, ICSP, Mindi, MiWi, MPASM, MPLAB Certified
logo, MPLIB, MPLINK, mTouch, Octopus, Omniscient Code
Generation, PICC, PICC-18, PICDEM, PICDEM.net, PICkit,
PICtail, REAL ICE, rfLAB, Select Mode, Total Endurance,
TSHARC, UniWinDriver, WiperLock and ZENA are
trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
SQTP is a service mark of Microchip Technology Incorporated
in the U.S.A.
All other trademarks mentioned herein are property of their
respective companies.
© 2010, Microchip Technology Incorporated, Printed in the
U.S.A., All Rights Reserved.
Printed on recycled paper.
ISBN: 978-1-60932-035-5
Microchip received ISO/TS-16949:2002 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, microperipherals, nonvolatile memory and
analog products. In addition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
 2010 Microchip Technology Inc.
DS01311A-page 11
WORLDWIDE SALES AND SERVICE
AMERICAS
ASIA/PACIFIC
ASIA/PACIFIC
EUROPE
Corporate Office
2355 West Chandler Blvd.
Chandler, AZ 85224-6199
Tel: 480-792-7200
Fax: 480-792-7277
Technical Support:
http://support.microchip.com
Web Address:
www.microchip.com
Asia Pacific Office
Suites 3707-14, 37th Floor
Tower 6, The Gateway
Harbour City, Kowloon
Hong Kong
Tel: 852-2401-1200
Fax: 852-2401-3431
India - Bangalore
Tel: 91-80-3090-4444
Fax: 91-80-3090-4123
India - New Delhi
Tel: 91-11-4160-8631
Fax: 91-11-4160-8632
Austria - Wels
Tel: 43-7242-2244-39
Fax: 43-7242-2244-393
Denmark - Copenhagen
Tel: 45-4450-2828
Fax: 45-4485-2829
India - Pune
Tel: 91-20-2566-1512
Fax: 91-20-2566-1513
France - Paris
Tel: 33-1-69-53-63-20
Fax: 33-1-69-30-90-79
Japan - Yokohama
Tel: 81-45-471- 6166
Fax: 81-45-471-6122
Germany - Munich
Tel: 49-89-627-144-0
Fax: 49-89-627-144-44
Atlanta
Duluth, GA
Tel: 678-957-9614
Fax: 678-957-1455
Boston
Westborough, MA
Tel: 774-760-0087
Fax: 774-760-0088
Chicago
Itasca, IL
Tel: 630-285-0071
Fax: 630-285-0075
Cleveland
Independence, OH
Tel: 216-447-0464
Fax: 216-447-0643
Dallas
Addison, TX
Tel: 972-818-7423
Fax: 972-818-2924
Detroit
Farmington Hills, MI
Tel: 248-538-2250
Fax: 248-538-2260
Kokomo
Kokomo, IN
Tel: 765-864-8360
Fax: 765-864-8387
Los Angeles
Mission Viejo, CA
Tel: 949-462-9523
Fax: 949-462-9608
Santa Clara
Santa Clara, CA
Tel: 408-961-6444
Fax: 408-961-6445
Toronto
Mississauga, Ontario,
Canada
Tel: 905-673-0699
Fax: 905-673-6509
Australia - Sydney
Tel: 61-2-9868-6733
Fax: 61-2-9868-6755
China - Beijing
Tel: 86-10-8528-2100
Fax: 86-10-8528-2104
China - Chengdu
Tel: 86-28-8665-5511
Fax: 86-28-8665-7889
Korea - Daegu
Tel: 82-53-744-4301
Fax: 82-53-744-4302
China - Chongqing
Tel: 86-23-8980-9588
Fax: 86-23-8980-9500
Korea - Seoul
Tel: 82-2-554-7200
Fax: 82-2-558-5932 or
82-2-558-5934
China - Hong Kong SAR
Tel: 852-2401-1200
Fax: 852-2401-3431
Malaysia - Kuala Lumpur
Tel: 60-3-6201-9857
Fax: 60-3-6201-9859
China - Nanjing
Tel: 86-25-8473-2460
Fax: 86-25-8473-2470
Malaysia - Penang
Tel: 60-4-227-8870
Fax: 60-4-227-4068
China - Qingdao
Tel: 86-532-8502-7355
Fax: 86-532-8502-7205
Philippines - Manila
Tel: 63-2-634-9065
Fax: 63-2-634-9069
China - Shanghai
Tel: 86-21-5407-5533
Fax: 86-21-5407-5066
Singapore
Tel: 65-6334-8870
Fax: 65-6334-8850
China - Shenyang
Tel: 86-24-2334-2829
Fax: 86-24-2334-2393
Taiwan - Hsin Chu
Tel: 886-3-6578-300
Fax: 886-3-6578-370
China - Shenzhen
Tel: 86-755-8203-2660
Fax: 86-755-8203-1760
Taiwan - Kaohsiung
Tel: 886-7-536-4818
Fax: 886-7-536-4803
China - Wuhan
Tel: 86-27-5980-5300
Fax: 86-27-5980-5118
Taiwan - Taipei
Tel: 886-2-2500-6610
Fax: 886-2-2508-0102
China - Xian
Tel: 86-29-8833-7252
Fax: 86-29-8833-7256
Thailand - Bangkok
Tel: 66-2-694-1351
Fax: 66-2-694-1350
Italy - Milan
Tel: 39-0331-742611
Fax: 39-0331-466781
Netherlands - Drunen
Tel: 31-416-690399
Fax: 31-416-690340
Spain - Madrid
Tel: 34-91-708-08-90
Fax: 34-91-708-08-91
UK - Wokingham
Tel: 44-118-921-5869
Fax: 44-118-921-5820
China - Xiamen
Tel: 86-592-2388138
Fax: 86-592-2388130
China - Zhuhai
Tel: 86-756-3210040
Fax: 86-756-3210049
01/05/10
DS01311A-page 12
 2010 Microchip Technology Inc.
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