AN-1040: RF Power Calibration Improves Performance of Wireless Transmitters (Rev. 0) PDF

AN-1040
APPLICATION NOTE
One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com
RF Power Calibration Improves Performance of Wireless Transmitters
by Eamon Nash
INTRODUCTION
power in this manner results in some power loss in the transmit
path. This directional coupler insertion loss is usually a few
tenths of a decibel.
Measurement and control of RF power is a critical consideration when designing a wireless transmitter. High power RF
amplifiers (PAs) rarely operate in open-loop mode, that is,
when the power to the antenna is not in some way monitored.
External factors such as regulatory requirements on the amount
of power transmitted, network robustness, and the need to coexist with other wireless networks, demand that there be tight
control of transmitted power. In addition to these external
requirements, precise RF power control can result in improved
spectral performance and can save cost and energy in the
transmitter’s power amplifier.
In wireless infrastructure applications where maximum transmitted power typically ranges from 30 dBm to 50 dBm (1 W to
100 W), the signal coming from the directional coupler is still
too strong for the RF detector that will measure it. As a result,
some additional attenuation is required between the coupler
and the RF detector.
Modern rms and non-rms responding RF detectors have a power
detection range of anywhere from 30 dB to 100 dB and provide
a temperature and frequency stable output. In most applications,
the detector output is applied to an analog-to-digital converter
(ADC) to be digitized. Using calibration coefficients stored in
nonvolatile memory (EEPROM), the code from the ADC is
converted into a transmitted power reading. This power reading
is compared to a setpoint power level. If there is a discrepancy
between the setpoint and the measured power, a power
adjustment is made. This power adjustment can be made at any
one of a number of points in the signal chain. The amplitude of
the baseband data driving the radio can be adjusted, a variable
gain amplifier (at IF or RF) can be adjusted, or the gain of the
PA can be changed. In this way, the gain control loop regulates
itself and keeps the transmitted power within desired limits. It is
important to note that the gain control transfer functions of
VVAs and PAs are often quite nonlinear. As a result, the actual
gain change resulting from a given gain adjustment is uncertain.
This reinforces the need for a control loop that provides feedback
on changes made and further guidance for subsequent
iterations.
To regulate its transmitted power, some form of factory calibration
of the PA output power may be necessary. Calibration algorithms vary vastly in terms of their complexity and effectiveness.
This application note describes how a typical RF power control
scheme is implemented and compares the effectiveness and
efficiency of various factory calibration algorithms.
TYPICAL WIRELESS TRANSMITTER WITH
INTEGRATED POWER CONTROL
Figure 1 shows a block diagram of a typical wireless transmitter
that incorporates measurement and control of transmitted power.
Using a directional coupler, a small portion of the signal from
the PA is coupled off and fed to an RF detector. In this case, the
coupler is located close to the antenna, but after the duplexer and
isolator. Their associated power loss is thus factored in during
calibration.
Directional couplers typically have a coupling factor of 20 dB to
30 dB; therefore, the signal coming from the coupler is 20 dB to
30 dB lower than the signal going to the antenna. Coupling off
TO
RECEIVER
PIN
VGA/
VVA
HPA
DUPL
ATTN
DAC
MICROCONTROLLER
OR DSP
ADC
RF
POWER
METER
RF
DETECTOR
08385-001
EEPROM
POUT
Figure 1. Typical RF Power Amplifier with Integrated Transmit Power Control
(An integrated RF power detector provides continuous feedback on the current level of power being transmitted.
An external RF power meter can be used along with the RF power detector to calibrate the transmitter.)
Rev. 0 | Page 1 of 8
AN-1040
Application Note
TABLE OF CONTENTS
Introduction ...................................................................................... 1
Calibrating an RF Power Control Loop..........................................5
Typical Wireless Transmitter with Integrated Power Control .... 1
Field Operation of an RF Power Control Loop .............................6
The Need for Factory Calibration .................................................. 3
Postcalibration Errors .......................................................................7
RF Detector Transfer Function ....................................................... 3
Conclusions ........................................................................................8
Rev. 0 | Page 2 of 8
Application Note
AN-1040
THE NEED FOR FACTORY CALIBRATION
level change, the calibrated on-board RF detector acts like a
built-in power meter with an absolute accuracy that ensures
that the transmitter is always emitting the desired power within
a defined tolerance.
In the typical wireless transmitter system previously described,
almost none of the components provide very good absolute gain
accuracy specifications. Consider the case of a transmit power
error target of ±1 dB. The absolute gain of devices such as PAs,
voltage variable attenuators (VVAs), RF gain blocks, and other
components in the signal chain can vary from device to device
to such an extent that the resulting output power uncertainty is
significantly greater than ±1 dB. In addition, signal chain gain
varies further as the temperature and frequency change. As a
result, it is necessary to continually monitor and control the
power being transmitted.
A factory calibration procedure is described in the Calibrating
an RF Power Control Loop section. First, the characteristics of a
typical RF power detector should be examined. The linearity
and stability over temperature and frequency of the system’s RF
detector strongly influence the complexity of the calibration
routine and the achievable postcalibration accuracy.
RF DETECTOR TRANSFER FUNCTION
Figure 2 shows the transfer function of a log-responding RF
detector (log amp) vs. temperature exaggerated for illustrative
purposes. The log amp transfer function can be modeled using
a simple first-order equation within its linear operating range.
Three curves are shown: output voltage vs. input power at +25°C,
+85°C, and −40°C. At 25°C, the output voltage of the detector
ranges from around 1.8 V at −60 dBm input power to 0.4 V at
0 dBm. The transfer function closely follows an imaginary
straight line, which has been laid over the trace. Although the
transfer function deviates from this straight line at the extremities, note that there are also signs of nonlinearity at power levels
between −10 dBm and −5 dBm.
Output power calibration can be defined as the transfer of the
precision of an external reference into the system being calibrated. A calibration procedure involves disconnecting the
antenna and replacing it with an external measurement reference
such as an RF power meter, as shown in Figure 1. In this way,
the accuracy of a precise external power meter is transferred
into the transmitter’s integrated power detector. The calibration
procedure involves setting one or more power levels, taking
the reading from the power meter and the voltage from the
RF detector, and storing all of this information in nonvolatile
memory (EEPROM). Then, with the power meter removed and
the antenna reconnected, the transmitter is able to precisely
regulate its own power. As parameters such as amplifier gain vs.
temperature, transmit frequency, and desired output power
DETECTOR OPERATING RANGE
2.2
VOUT AT –40°C
VOUT AT +25°C
VOUT AT +85°C
2.0
1.8
1.6
VOUT (V)
1.4
DETECTOR SLOPE = Y/X
X
1.2
Y
1.0
DETECTOR
NONLINEARITY
0.8
0.6
0.4
INTERCEPT
0
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
PIN (dBm)
0
5
10 15 20
08385-002
0.2
Figure 2. Transfer Function (VOUT vs. PIN) of a Log-Responding RF Power Detector with Temperature Drift Exaggerated for Illustrative Purposes
Rev. 0 | Page 3 of 8
AN-1040
Application Note
A quick calculation suggests that this detector has a slope of
approximately −25 mV/dB; that is, a 1 dB change in input
power results in a 25 mV change in output voltage. This slope
is constant over the linear portion of the dynamic range. Thus,
notwithstanding the slightly degraded nonlinearity that was
identified at around −10 dBm, the behavior of the transfer
function at 25°C can be modeled using the following equation:
VOUT = Slope × (PIN − Intercept)
where Intercept is the point at which the extrapolated straight
line fit crosses the x-axis of the plot (see Figure 2).
Therefore, the transfer function of the detector can be modeled
using this simple first-order equation. From a calibration
perspective, this is useful because it allows the transfer function
of the detector to be established by applying and measuring as
few as two different power levels during the calibration
procedure.
Next, consider the behavior of this imaginary detector over
temperature. At an input power of –10 dBm, note that the
output voltage changes by approximately 100 mV from ambient
temperature to either −40°C or +85°C. From the previous calculation of the slope (−25 mV/dB), this equates to a deviation in
measured power of ±4 dB, unacceptable in most practical systems.
In practice, a detector whose transfer function has minimal
drift vs. temperature is needed. This ensures that a calibration
procedure performed at ambient temperature is also valid over
temperature, allowing the transmitter to be factory calibrated at
ambient temperature and avoiding expensive and time-consuming
calibration cycles at hot and cold temperatures.
If the transmitter is frequency-agile and needs to transmit at
multiple frequencies within a defined frequency band, the user
must pay attention to the behavior of the detector vs. frequency.
Ideally, an RF detector whose response does not change significantly within a defined frequency band should be used. This allows
calibration of the transmitter at a single frequency (generally at
mid-band) and ensures that there is little or no loss of accuracy as
the frequency changes.
Table 1 shows the detection ranges and temperature stability of
various rms and non-rms responding detectors from Analog
Devices, Inc.
Table 1. RMS and Non-RMS Responding RF Power Detectors
Device
AD8317
AD8318
AD8319
ADL5513
ADL5519
AD8361
ADL5501
AD8362
AD8363
AD8364
Max Input
Frequency (GHz)
10
8
10
4
10
2.5
6
3.8
6
2.7
Dynamic Range (dB)
55
70
45
80
62
30
30
65
50
60
Temperature
Drift (dB)
±0.5
±0.5
±0.5
±0.5
±0.5
±0.25
±0.1
±1.0
±0.5
±0.5
Package
2 mm × 3 mm 8-lead LFCSP
4 mm × 4 mm 16-lead LFCSP
2 mm × 3 mm 8-lead LFCSP
3 mm × 3 mm 16-lead LFCSP
5 mm × 5 mm 32-lead LFCSP
6-lead SOT-23, 8-lead MSOP
2.1 mm × 2 mm 6-lead SC-70
6.4 mm × 5 mm 16-lead TSSOP
4 mm × 4 mm 16-lead LFCSP
5 mm × 5 mm 32-lead LFCSP
Rev. 0 | Page 4 of 8
Comments
Non-rms log detector
Non-rms log detector
Non-rms log detector
Non-rms log detector
Dual non-rms log detector
Linear in V/V rms detector
Linear in V/V rms detector
RMS log detector
RMS log detector
Dual rms log detector
Application Note
AN-1040
SET RF POWER TO MAX POWER (APPROXIMATELY)
MEASURE CODE FROM RF LOG DETECTOR ADC (CODEHIGH)
USE RF POWER METER TO
MEASURE POWER AT ANTENNA CONNECTOR (PWRHIGH) (UNIT = dBm)
SET RF POWER TO MIN POWER (APPROXIMATELY)
MEASURE CODE FROM RF LOG DETECTOR ADC (CODELOW)
USE RF POWER METER TO
MEASURE POWER AT ANTENNA CONNECTOR (PWRLOW)
SLOPE = (CODEHIGH – CODELOW)/(PWRHIGH – PWRLOW) (UNIT = CODES/dB)
STORE SLOPE AND INTERCEPT IN NONVOLATILE RAM
08385-003
INTERCEPT = PHIGH – (CODEHIGH/SLOPE)
Figure 3. Simple Two-Point Calibration Procedure to Calibrate a Transmitter with an Integrated Log Detector
CALIBRATING AN RF POWER CONTROL LOOP
Figure 3 shows the flowchart that can be used to calibrate a
transmitter similar to the one shown in Figure 1. This simple
and quick two-point calibration is useful where power levels
need to be set only approximately (but must be measured
precisely). For this calibration to be effective, the integrated RF
detector must be stable vs. temperature and frequency and must
have a predictable response that can be modeled using a simple
equation.
Ensure that the operating power range of the transmitter maps
comfortably into the RF detector’s linear operating range. To
begin, remove the antenna and connect the power meter to the
antenna connector. Next, set an output power level close to
maximum power. The power at the antenna connector is
measured by the power meter and is sent to the transmitter’s onboard microcontroller or digital signal processor (DSP). At the
same time, the RF detector ADC is sampled and its reading is
provided to the transmitter’s processor.
Next, reduce the output power of the transmitter to a level that
is close to minimum power and repeat the procedure (measure
power at the antenna connector and the sample RF detector ADC).
With these four readings (low and high power level, low and
high ADC code), the slope and intercept can be calculated (see
Figure 3) and stored in nonvolatile memory.
Rev. 0 | Page 5 of 8
AN-1040
Application Note
DETERMINE DESIRED OUTPUT POWER (PSET)
SET OUTPUT POWER-BASED ON BEST FIRST GUESS
ENSURING THAT (POUT < PSET)
MEASURE CODE FROM RF LOG DETECTOR ADC (CODEOUT)
CALCULATE TRANSMITTED POWER
POUT = INTERCEPT + CODEOUT/SLOPE
IS ABS |PSET – POUT|
≤0.5dB
YES
NO
POUT < PSET
DECREMENT VGA GAIN
BY APPROXIMATELY 0.5dB
08385-004
POUT > PSET
IS POUT > PSET
OR IS POUT < PSET
INCREMENT VGA GAIN
BY APPROXIMATELY 0.5dB
Figure 4. Operation of Transmitter After Calibration
FIELD OPERATION OF AN RF POWER CONTROL
LOOP
Figure 4 shows the flowchart that can be used to precisely set
power in a transmitter after calibration. In this example, the
goal is to have a transmit power error that is less than or equal
to ±0.5 dB. Initially, an output power level is set based on a best
first guess. Next, the detector ADC is sampled. The slope and
intercept are retrieved from memory and the transmitted
output power level is calculated.
If the output power is not within ±0.5 dB of PSET, the output
power is incremented or decremented by approximately 0.5 dB
using a voltage variable attenuator (VVA). The term approximately is used because the VVA may have a nonlinear transfer
function. The transmitted power is again measured and further
power increments are applied until the transmitted power error
is less than ±0.5 dB.
When the power level is within tolerance, it is continually monitored and adjusted if necessary. For example, if the gain of a
component in the signal chain drifts with changing temperature, the loop is activated when the measured power goes
outside its ±0.5 dB setpoint range.
Other variations on this algorithm exist. For example, if it is
desirable to keep the output power as low as possible but still no
more than 0.5 dB from the setpoint, a different approach must
be taken. In this case, the first power setting is at a level that is
below the desired power level (and outside the tolerance). The
loop then measures the power but setpoint increments are
much smaller, for example, +0.1 dB. In this way, the output
power always approaches the setpoint from a value that is less
than the setpoint. As soon as it enters the −0.5 dB band, power
increments stop. This ensures that the actual level is always
below the setpoint level while still being within tolerance.
Rev. 0 | Page 6 of 8
Application Note
AN-1040
2.2
2.5
2.2
2.0
2.0
1.8
1.5
1.8
1.5
1.6
1.0
1.6
1.0
1.4
0.5
1.4
0.5
1.2
0
1.0
–0.5
0.8
–1.0
–1.5
–2.0
2.0
VOUT AT –40°C
VOUT AT +25°C
VOUT AT +85°C
ERROR AT –40°C
ERROR AT +25°C
ERROR AT +85°C
2.5
VOUT AT –40°C
VOUT AT +25°C
VOUT AT +85°C
ERROR AT –40°C
ERROR AT +25°C
ERROR AT +85°C
2.0
VOUT1
0.6
–1.5
0.6
0.4
–2.0
0.4
0.2
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
PIN2
PIN (dBm)
0
5
INTERCEPT
PIN1
0.2
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
PIN1
Figure 7. Two-Point Calibration with Calibration Points Close Together
Provides Improved Accuracy over a Narrow Range
2.2
2.0
2.0
1.8
1.5
1.8
1.5
1.6
1.0
1.6
1.0
1.4
0.5
1.4
0.5
1.2
0
1.2
0
1.0
–0.5
1.0
–0.5
0.8
–1.0
0.8
–1.0
ERROR AT –40°C
ERROR AT +25°C
ERROR AT +85°C
VOUT (V)
2.0
VOUT (V)
5
2.5
VOUT AT –40°C
VOUT AT +25°C
VOUT AT +85°C
ERROR (dB)
2.2
–2.5
0
PIN (dBm)
PIN2
Figure 5. Two-Point Calibration with Calibration Points in Linear Operating
Range of Detector Provides Good Overall Performance
2.5
VOUT AT –40°C
VOUT AT +25°C
VOUT AT +85°C
ERROR AT –40°C
ERROR AT +25°C
ERROR AT +85°C
2.0
–1.5
0.6
–1.5
0.4
–2.0
0.4
–2.0
0.2
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
–2.5
0.2
–65 –60 –55 –50 –45 –40 –35 –30 –25 –20 –15 –10 –5
0.6
ERROR (dB)
–1.0
08385-007
–0.5
0.8
VOUT2
ERROR (dB)
1.0
VOUT (V)
0
08385-005
VOUT1
1.2
ERROR (dB)
VOUT (V)
VOUT2
58dB DYNAMIC RANGE
PIN (dBm)
Figure 6. Moving Calibration Points Apart and into Less Linear Operating
Range Extends Operating Range but at the Cost of Degraded Accuracy
–2.5
0
5
PIN (dBm)
08385-008
5
08385-006
0
Figure 8. Multipoint Calibration Extends Detector Range and Can Improve
Linearity but at the Cost of a More Complex Calibration Procedure
POSTCALIBRATION ERRORS
Figure 5 to Figure 8 show data from the same RF detector but
use a different choice and number of calibration points. Figure 5
shows the detector transfer function at 2.2 GHz for the AD8318,
a wide dynamic range RF log detector that operates up to 8 GHz.
In this case, the detector has been calibrated using a two-point
calibration (at −12 dBm and −52 dBm). When calibration is
complete, the residual measurement error can be plotted. Note
that the error is not zero, even at the ambient temperature at
which calibration was performed. This is because the log amp
does not perfectly follow the ideal VOUT vs. PIN equation (VOUT =
Slope × (PIN − Intercept)), even within its operating region. The
error at the −12 dBm and −52 dBm calibration points is,
however, equal to zero by definition.
Figure 5 also includes error plots for the output voltage at −40°C
and +85°C. These error plots are calculated using the 25°C slope
and intercept calibration coefficients. Unless a temperature-based
calibration routine is implemented, the 25°C calibration coefficients with slight residual temperature drift must be used.
In many applications, it is desirable to have higher accuracy when
the PA is transmitting at its maximum power. This makes sense
from a number of perspectives. First, there may be regulatory
requirements that demand this higher level of accuracy at full or
rated power. However, from a system design perspective, there
is also value in increased accuracy at rated power. Consider a
transmitter that is designed to transmit 45 dBm (approximately
30 W). If calibration can at best provide accuracy of ±2 dB, then
the PA circuitry (power transistors and heat sinks) must be
designed to safely transmit as much as 47 dBm or 50 W. This
constitutes a waste of money and space. Instead, a system where
the postcalibration accuracy is ±0.5 dB can be designed so that
the PA must be overdimensioned only to safely transmit 45.5 dBm
or approximately 36 W.
By changing the points at which calibration is performed, the
achievable accuracy can in some cases be greatly influenced.
Figure 7 shows the same measured data as Figure 5 but using
different calibration points. Notice how the accuracy is very
high (about ±0.25 dB) from −10 dBm to −30 dBm in Figure 7.
Rev. 0 | Page 7 of 8
AN-1040
Application Note
However, accuracy falls off at lower power levels further away
from the calibration points.
Figure 6 shows how calibration points can be moved to increase
dynamic range at the expense of linearity. In this case, the calibration points are −4 dBm and −60 dBm. These points are at the
end of the device’s linear range. Once again, an error of 0 dB at
the calibration points at 25°C can be seen, and the range over
which the AD8318 maintains an error of <±1 dB is extended to
60 dB at 25°C and 58 dB over temperature. The disadvantage of
this approach is that the overall measurement error increases,
especially in this case at the top end of the detector’s range.
Figure 8 shows the postcalibration error using a more elaborate
multipoint algorithm. In this case, multiple output power levels
(separated by 6 dB in this example) are applied to the transmitter and the detector’s output voltage at each power level is
measured. These measurements are used to break the transfer
function down into segments, with each segment having its own
slope and intercept. This algorithm tends to greatly reduce errors
due to detector nonlinearity and leaves temperature drift as the
main source of errors. The disadvantage of this approach is that
the calibration procedure takes longer and more memory is
required to store the multiple slope and intercept calibration
coefficients.
Figure 8 illustrates an interesting difference between the
behavior of the power detector at the low and high ends of its
dynamic range. Although multipoint calibration extends the
high end dynamic range, this range extension is not very useful
because of the increased temperature drift. Notice how the
ambient, hot, and cold traces diverge above −10 dBm. At low
power levels, the result is more useful. Again, the multipoint
calibration helps to extend the low end dynamic range.
However, in this case, the hot and cold traces closely track the
ambient trace, even as it becomes nonlinear. So when this
nonlinearity has been removed using multipoint calibration, this
calibration holds up very well over temperature. This usefully
extends the transfer function of the AD8318 down to −65 dBm.
CONCLUSIONS
In applications where accurate RF power transmission is required,
some form of system calibration is necessary. Modern IC-based
RF power detectors have linear responses and are temperature
and frequency stable. This can significantly simplify system
calibration and can provide a system accuracy of ±0.5 dB or
better. The placement and number of calibration points can have
a significant effect on the achievable postcalibration accuracy.
©2009 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
AN08385-0-12/09(0)
Rev. 0 | Page 8 of 8
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