### AN-937: Designing Amplifiers Circuits: How to Avoid Common Problems (Rev. 0)

```AN-937
APPLICATION NOTE
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Designing Amplifier Circuits: How to Avoid Common Problems
by Charles Kitchin
When compared to assemblies of discrete semiconductors,
modern operational amplifiers (op amps) and instrumentation amplifiers (in-amps) provide great benefits to designers.
Although there are many published articles on circuit
applications, all too often, in the haste to assemble a circuit,
basic issues are overlooked leading to a circuit that does not
function as expected. This application note discusses the most
common design problems and offers practical solutions.
MISSING DC BIAS CURRENT RETURN PATH
One of the most common application problems encountered is
the failure to provide a dc return path for bias current in accoupled op amp or in-amp circuits. In Figure 1 a capacitor is
connected in series with the noninverting (+) input of an op
amp. This ac coupling is an easy way to block dc voltages
associated with the input voltage (VIN). This is especially useful
in high gain applications, where even a small dc voltage at
amplifier input can limit the dynamic range or even result in
output saturation. However, capacitive coupling into a highimpedance input without providing a dc path for current
flowing in the positive input leads to problems.
down toward the negative supply. The bias voltage is amplified
by the closed-loop dc gain of the amplifier.
This process can be lengthy. For example, an amplifier with a
field effect transistor (FET) input, having a 1 pA bias current,
coupled via a 0.1-μF capacitor, has an IC charging rate, I/C, of
10–12/10–7 = 10 μV per sec
or 600 μV per minute. If the gain is 100, the output drifts at
0.06 V per minute. Therefore, a casual lab test, using an accoupled scope, may not detect this problem, and the circuit
may not fail until hours later. It is important to avoid this
problem altogether.
+VS
C1
VIN
OP AMP
VOUT
R3 –VS
R2
DESIGN EQUATIONS
–3dB INPUT BW = 1/(2π R1 C1)
R1 IS TYPICALLY SET EQUAL TO
THE PARALLEL COMBINATION
OF R2 AND R3.
Figure 2. Correct Method for AC- Coupling an Op Amp Input for
Dual-Supply Operation
0.1µF
VIN
Figure 2 shows a simple solution to this common problem. In
this example, a resistor is connected between the op amp input
and ground to provide a path for the input bias current. To
minimize offset voltages caused by input bias currents, which
track one another when using bipolar op amps, R1 is usually set
equal to the parallel combination of R2 and R3.
VOUT
0.1µF
07034-001
R3 –VS
R2
R1
0.1µF
+VS
OP AMP
0.1µF
07034-002
INTRODUCTION
Figure 1. A Nonfunctional AC-Coupled Op Amp Circuit
The input bias current flows through the coupling capacitor,
charging it, until the common-mode voltage rating of the
amplifier’s input circuit is exceeded or the output is driven into
limits. Depending on the polarity of the input bias current, the
capacitor charges up toward the positive supply voltage or
Note, however, that this resistor always introduces some noise
into the circuit, so there is a trade-off between circuit input
impedance, the size of the input coupling capacitor needed, and
the Johnson noise added by the resistor. Typical resistor values
are generally in the range of about 100,000 Ω to 1 MΩ.
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Introduction ...................................................................................... 1
Correctly Providing In-Amp Reference Voltage.......................4
Missing DC Bias Current Return Path .......................................... 1
Preserving PSR When Amplifiers Are Referenced from the
Supply Rail Using Voltage Dividers ............................................5
Supplying Reference Voltages for In-Amps, Op Amps, and
Decoupling Single-Supply Op Amp Circuits ............................6
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A similar problem can affect an in-amp circuit. Figure 3 shows
in-amp circuits that are ac-coupled using two capacitors, without providing an input bias current return path. This problem
is common with in-amp circuits using both dual-power
supplies (Figure 3a) and single-power supplies (Figure 3b).
+VS
Simple solutions for these circuits are shown in Figure 5 and
Figure 6. A high value resistance (RA, RB) is added between each
input and ground. This is a simple and practical solution for
dual-supply in-amp circuits. The resistors provide a discharge
path for input bias currents. In the dual-supply example, both
inputs are referenced to ground. In the single-supply example,
the inputs can be referenced either to ground (VCM tied to
ground) or to a bias voltage, usually one-half the maximum
input voltage range.
+VS
0.1µF
VOUT
IN-AMP
VOUT
IN-AMP
VREF
VREF
+VS
VS/2
0.1µF
a). DUAL SUPPLY
RA
07034-003
–VS
b). SINGLE SUPPLY
RB
Figure 3. Nonfunctional AC-Coupled In-Amp Circuits
0.1µF
The problem can also occur with transformer coupling, as in
Figure 4, if no dc return path to ground is provided in the
transformer’s secondary circuit.
Figure 5. Correct Method for Transformer Input Coupling to an In-Amp
The same principle can be used for transformer-coupled inputs
(Figure 5) unless the transformer secondary winding has a
center tap, which can be grounded or connected to VCM. In
these circuits, there is a small offset voltage error due to
mismatches between the resistors and/or the input bias
currents. To minimize these errors, a third resistor, about 1/10th
their value (but still large compared to the differential source
resistance), can be connected between the two in-amp inputs
(thus bridging both resistors).
0.1µF
VOUT
VREF
07034-004
0.1µF
–VS
Figure 4. A Nonfunctional Transformer-Coupled In-Amp Circuit
+VS
+VS
0.1µF
VIN
RA
VREF
–VS
+VS
IN-AMP
VOUT
IN-AMP
IN-AMP
RB
0.1µF
–VS
07034-006
0.1µF
VIN
0.1µF
VOUT
VIN
RA
RB
VREF
VCM
VOUT
IN-AMP
VREF
NORMALLY
VS/2
VCM CAN BE GROUND OR, FOR MAXIMUM
INPUT DYNAMIC RANGE, SET VCM TO THE
CENTER OF THE MAXIMUM INPUT RANGE.
a). DUAL SUPPLY
b). SINGLE SUPPLY
07034-005
VIN
Figure 6. A High Value Resistor Between Each Input and Ground Supplies the Necessary Bias Current Return Path
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AN-937
For example, a popular in-amp design configuration uses three
op amps connected as shown in Figure 8. The overall signal
gain is
SUPPLYING REFERENCE VOLTAGES FOR IN-AMPS,
Figure 7 shows a single-supply circuit where an in-amp is
driving a single-ended analog-to-digital converter (ADC). The
amplifier reference provides a bias voltage corresponding to
zero differential input, and the ADC reference provides the
scale factor. A simple RC low-pass antialiasing filter is often
used between in-amp output and ADC input to reduce out-ofband noise. Often designers are tempted to use simple
approaches, such as resistance dividers, to supply the in-amp
in-amps.
5V
R2 R4
=
R1 R3
The gain for the reference input (if driven from low impedance)
is unity. However, in this example, the in-amp has its reference
pin tied directly to a simple voltage divider. This unbalances the
symmetry of the subtractor circuit and the division ratio of the
voltage divider. This reduces the in-amp’s common-mode
rejection and its gain accuracy. However, in some cases, R4 is
accessible, thus its resistance value can be reduced by an
amount equal to the resistance looking back into the paralleled
legs of the voltage divider (50 kΩ in this case). In this case, the
circuit behaves as though a low impedance voltage source equal
to one-half the supply voltage is applied to the original value of
R4. In addition, the subtractor accuracy is maintained.
0.1µF
OUTPUT R
C
TYPICAL VALUES FOR RC
LP FILTER
R = 50Ω TO 200Ω
SET C TO PROVIDE DESIRED –3dB
CIRCUIT BANDWIDTH USING
C = 1/(2π R F)
REFERENCE
TYPICALLY
2.5V TO 5V
07034-007
REFERENCE
TYPICALLY 2.5V
This approach can not be used if the in-amp is provided as a
closed single package (an IC). Another consideration is that the
temperature coefficients of the resistors in the voltage divider
should track those of R4 and the other resistors in the
subtractor. Finally, the approach locks out the possibility of
having the reference be adjustable. If, on the other hand, one
attempts to use small resistor values in the voltage divider to
make the added resistance negligible, this increases power
supply current consumption and increases the dissipation of
the circuit. Such brute force is not a good design approach.
Figure 7. An In-Amp Drives an ADC in a Typical Single-Supply Circuit
CORRECTLY PROVIDING IN-AMP REFERENCE
VOLTAGE
A common assumption is that the in-amp reference-input
terminal is at high impedance, since it is an input. Therefore, a
designer may be tempted to connect a high impedance source,
such as a resistive divider, to the reference pin of an in-amp.
This can introduce serious errors with some types of
instrumentation amplifiers (see Figure 8).
Figure 9 shows a better solution, using a low power op amp
buffer between the voltage divider and the in-amp reference
input. This eliminates the impedance matching and temperature tracking problem and allows the reference to be easily
INVERTING
INPUT
3
A1
R1
1
2
R2
R5
2
A3
RG
6
VOUT
3
+VS
R6
6
A2
5
NONINVERTING
INPUT
7
INPUT
SECTION
R3
R4
OUTPUT
SECTION
REFERENCE
INPUT
100kΩ
VOLTAGE
DIVIDER
100kΩ
Figure 8. Incorrect Use of a Simple Voltage Divider to Directly Drive the
Reference Pin of a 3 Op amp Instrumentation Amplifier
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07034-008
IN-AMP
⎞
⎟⎟
⎠
where:
5V
0.1µF
DIFFERENTIAL
INPUT
⎛
R
R6 ⎞ ⎛ R2
⎟⎜
G = ⎜⎜1 + 5 +
RG RG ⎟⎠ ⎜⎝ R1
⎝
AN-937
INVERTING
INPUT
R1
A1
R2
R5
VOUT
A3
RG
+VS
A2
INPUT
SECTION
R3
REFERENCE
INPUT
R4
OUTPUT
SECTION
100kΩ
OP AMP
BUFFER
VOLTAGE
DIVIDER
100kΩ
07034-009
R6
NONINVERTING
INPUT
Figure 9. Driving the Reference Pin of an In-Amp from the Low Impedance Output of an Op Amp
An often-overlooked consideration is that any noise, transients,
or drift of power supply voltage, VS, fed in through the
reference input adds directly to the output, attenuated only by
the divider ratio. Practical solutions include bypassing and
filtering, and even generating the reference voltage with a
off VS.
This consideration is important when designing circuits with
both in-amps and op amps. Power supply rejection (PSR)
techniques isolate an amplifier from power supply hum, noise,
and any transient voltage variations present on the power rails.
This is important because many real world circuits contain,
connect to, or exist in environments that offer less than ideal
supply voltage. In addition, ac signals present on the supply
lines can be fed back into the circuit, amplified, and, under the
right conditions, stimulate a parasitic oscillation.
Modern op amps and in-amps provide substantial low
frequency power supply rejection as part of their design. This is
something that most engineers take for granted. Many modern
op amps and in-amps have PSR specs of 80 dB to over 100 dB,
reducing the effects of power supply variations by a factor of
10,000 to 100,000. Even a modest PSR specification of 40 dB
isolates supply variations from the amplifier by a factor of 100.
Nevertheless, high frequency bypass capacitors (such as those
in Figure 1 through Figure 7) are always desirable and often
essential. In addition, when designers use a simple resistance
divider on the supply rail and an op amp buffer to supply a
reference voltage for an in-amp, any variations in power supply
voltage are passed through this circuitry with little attenuation
and add directly to the in-amp output level. Therefore, unless
low-pass filtering is provided, the normally excellent PSR of the
IC is lost.
In Figure 10, a large capacitor is added to the voltage divider to
filter its output from power supply variations and preserve PSR.
The −3 dB pole of this filter is set by the parallel combination of
R1/R2 and Capacitor C1. The pole should be set approximately
10 times lower than the lowest frequency of concern.
+VS
BRIDGE
SENSOR
+VS
0.1µF
OUTPUT
IN-AMP
REFERENCE
INPUT
+VS
+VS
R1
100kΩ
0.1µF
VS/2
OP AMP
OP1177
CF
100µF
R2
100kΩ
R3
50kΩ
0.01µF
DESIGN EQUATIONS
CF = 1/((2π) 50kΩ × FREQUENCY IN Hz)
COOKBOOK VALUES: 10µF (0.3Hz) TO 100µF (0.03Hz)
R3 = PARALLEL COMBINATION OF R1, R2
CF = 1/(2πR3f), R3 = 50kΩ, f = –30dB FREQUENCY IN Hz
07034-010
PRESERVING PSR WHEN AMPLIFIERS ARE
REFERENCED FROM THE SUPPLY RAIL USING
VOLTAGE DIVIDERS
Figure 10. Decoupling the Reference Circuit to Preserve PSR
The cookbook values shown in Figure 10 provide a −3 dB pole
frequency of approximately 0.03 Hz. The small (0.01 μF)
capacitor across R3 minimizes resistor noise.
The filter takes time to charge up. Using the cookbook values,
the rise time at the reference input is several time constants
(where T = R3Cf = 5 s), or about 10 to 15 seconds.
The circuit in Figure 11 offers a further refinement. In this case,
the op amp buffer operates as an active filter, which allows the
use of much smaller capacitors for the same amount of power
supply decoupling. In addition, the active filter can be designed
to provide a higher Q and thus give a quicker turn-on time.
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AN-937
This circuit was tested with the component values shown in
Figure 11, with 12 V applied, and a 6 V filtered reference
voltage provided to the in-amp. A 1 V p-p sine wave of varying
frequency was used to modulate the 12 V supply, with the inamp gain set to unity. Under these conditions, as frequency was
decreased, no ac signal was visible on an oscilloscope, at VREF,
or at the in-amp output, until approximately 8 Hz. Measured
supply range for this circuit ranged from 4 V to greater than
25 V, with a low level input signal applied to the in-amp.
Circuit turn-on time was approximately two seconds.
DECOUPLING SINGLE-SUPPLY OP AMP CIRCUITS
Single-supply op amp circuits require biasing of the input
common-mode level to handle the positive and negative swings
of ac signals. When this bias is provided from the power supply
rail using voltage dividers, adequate decoupling is required to
preserve PSR.
A common, though incorrect, practice is to use a 100 kΩ/100
kΩ resistor divider with a 0.1 μF bypass capacitor to supply
VS/2 to the noninverting pin of the op amp. Using these values,
power supply decoupling is often inadequate because the pole
frequency is only 32 Hz.
+VS
+VS
BRIDGE
SENSOR
0.1µF
OUTPUT
C1
2µF
REFERENCE
INPUT
+VS
DESIGN EQUATIONS
Q = C1/4C2
2πf = 1/(R C1C2)
R3
100kΩ
0.1µF
VS/2
C2
1µF
OP AMP
OP1177
WHERE R = R3
AND R1 AND R2 = 2R3
FOR C1 = C2 Q = 0.5
FOR C1 = 2C2 Q = 0.707
+VS
R4
200kΩ
R1
200kΩ
R2
200kΩ
OP-AMP BUFFER
DRIVE TO IN-AMP
REFERENCE PIN
WITH BUILT-IN
ACTIVE FILTER
0.01µF
07034-011
IN-AMP
Figure 11. An Op Amp Buffer Connected as an Active Filter Drives the
Reference Pin of an In-Amp
+VS
1
2π (1/2RA) C2
BW2 =
1
2π RIN CIN
BW3 =
1
2π R1 C1
BW4 =
1
RA
100kΩ
*
VOUT
OP1177
VS/2
VIN
1µF
*
RIN
100kΩ VS/2
RB
100kΩ
C2
+VS
COUT
R2
150kΩ
CIN
FOR RA = RB
AND BW1 = 1/10 TH BW2, BW3, AND BW4
R1
TO MINIMIZE INPUT BIAS CURRENT ERRORS,
R2 SHOULD EQUAL RIN + (1/2 RA)
C1
*STAR
GROUND
VOUT = (VS/2) + VIN (1 + (R2/R1))
WHERE XC1 << R1
Figure 12. A Single-Supply Noninverting Amplifier Circuit
Showing Correct Power Suppy Decoupling.
Midband Gain = 1 + R2/R1
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07034-012
BW1 =
0.1µF
AN-937
biases the output to the same voltage. Coupling Capacitor C1
rolls the low frequency gain down toward unity from BW3.
Circuit instability (motor boating) Figure 12 (noninverting)
and Figure 13 (inverting) show circuits to accomplish VS/2
decoupled biasing for best results. In both cases, bias is
provided at the noninverting input, feedback causes the
inverting input to assume the same bias, and unity dc gain
A good rule of thumb when using a 100 kΩ/100 kΩ voltage
divider, as shown in Figure 12, is to use a C2 value of at least
10 μF for a 0.3 Hz −3 dB roll-off. A value of 100 μF (0.03 Hz
pole) should be sufficient for practically all circuits.
+VS
RA
100kΩ
BW2 =
1
2π R1 C1
C2
RB
100kΩ
1
BW3 =
FOR RA = RB
AND XC2 << XC1
C1
+VS
1µF
*
*
VS/2
OP1177
VS/2
R1
VOUT
COUT
R2
50kΩ
*STAR
GROUND
VOUT = (VS/2) + VIN (R2/R1)
WHERE XC1 << R1
TO MINIMIZE INPUT BIAS CURRENT ERRORS,
R2 SHOULD EQUAL 1/2 RA
07034-013
1
BW1 =
2π (1/2RA) C2
0.1µF
Figure 13. Proper Decoupling for a Single-Supply Inverting Amplifier Circuit.
Midband Gain = –R2/R1
Rev. 0 | Page 7 of 8
AN-937
NOTES