AN-940: Low Noise Amplifier Selection Guide for Optimal Noise Performance (Rev.

```AN-940
APPLICATION NOTE
One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com
Low Noise Amplifier Selection Guide for Optimal Noise Performance
by Paul Lee
INTRODUCTION
When evaluating an amplifier’s performance for a low noise
application, both internal and external noise sources must be
considered. This application note briefly discusses the fundamentals of both internal and external noise and identifies the
tradeoffs associated in selecting the optimal amplifier for low
noise design.
EXTERNAL NOISE SOURCES
External noise includes any type of external influences, such
as external components and electrical/electromagnetic interference. Interference is defined as any unwanted signals arriving
as either voltage or current, at any of the amplifier’s terminals
or induced in its associated circuitry. It can appear as spikes,
steps, sine waves, or random noise. Interference can come from
anywhere: machinery, nearby power lines, RF transmitters or
receivers, computers, or even circuitry within the same equipment (that is, digital circuits or switching-type power supplies).
If all interference is eliminated by careful design and/or layout
of the board, there can still be random noise associated with the
amplifier and its circuit components.
Noise from surrounding circuit components must be accounted
for. At temperatures above absolute zero, all resistances act as
noise sources due to thermal movement of charge carriers called
Johnson noise or thermal noise. This noise increases with resistance, temperature, and bandwidth. Voltage noise is shown in
Equation 1.
Vn  4kTBR
(1)
where:
Vn is voltage noise.
k is Boltzmann’s constant (1.38 × 10−23 J/K).
T is the temperature in Kelvin (K).
B is the bandwidth in hertz (Hz).
R is the resistance in ohms (Ω).
Current noise (noise associated with current flow) is shown in
Equation 2
In 
4kTB
R
where:
In is current noise.
k is Boltzmann’s constant (1.38 × 10−23 J/K).
T is the temperature in Kelvin (K).
B is the bandwidth in hertz (Hz).
R is the resistance in ohms (Ω).
Rev. D | Page 1 of 12
(2)
AN-940
Application Note
Introduction ...................................................................................... 1
Popcorn Noise ...............................................................................5
External Noise Sources .................................................................... 1
Summing the Noise Sources ........................................................5
Internal Noise Sources ..................................................................... 3
Noise Gain ......................................................................................6
Input-Referred Voltage Noise ..................................................... 4
Selecting Low Noise Op Amp ..........................................................7
Input-Referred Current Noise .................................................... 4
Conclusion..........................................................................................9
Flicker Noise.................................................................................. 5
References ........................................................................................ 12
Rev. D | Page 2 of 12
Application Note
AN-940
Resistors
INTERNAL NOISE SOURCES
For the purposes of this application note, the resistor noise is
limited to thermal (Johnson) noise. To keep a low level of this
type of noise, resistance values should be as low as possible
because RMS voltage of thermal (Johnson) noise is proportional
to the square root of the resistor value. For example, a 1 kΩ
resistor has a thermal noise of ~4 nV/√Hz at room temperature.
Noise appearing at the amplifier’s output is usually measured as
a voltage. However, it is generated by both voltage and current
sources. All internal sources are generally referred to the input,
that is, treated as uncorrelated or independent random noise
generators in series or in parallel with the inputs of an ideal
noise-free amplifier (see Figure 1). Because these noise sources
are considered random and/or exhibit Gaussian distribution
behavior, it is important to take proper care when summing the
noise sources as discussed in the Summing the Noise Sources
section.
For an in-depth analysis and low noise designs, other types of
resistor noise should be accounted for, such as contact noise
and shot noise. A few practical notes follow and they should
be considered when selecting a resistor.





Choose the largest practical wattage resistors, as the contact
noise is decreased with a larger volume of material.
Choose low noise resistive element material
Resistive elements composed of pure metals and/or
metal alloys in bulk exhibits low noise characteristics.
Such as Vishay Bulk Metal® foil technology resistors
(such as, S102C, Z201)
Wirewound technology resistors composed of metal
alloys have similar noise characteristics as Bulk Metal
foil technology, but are much more inductive.
Metal film technology resistors as thin film are noisier
than Bulk Metal foil or wirewound technology resistors
because of significant noise contributions from occlusions,
surface imperfections, and nonuniform depositions.
Thick film and carbon composition resistors are the
nosiest resistors.
Reactances
Reactances, such as capacitors and inductors, do not generate
noise, but the noise current through reactances develops noise
voltage as well as the associated parasitic.
Practical Tips
If the same noise appears at two or more points in a circuit (that
is, input bias current cancellation circuitry), the two noise sources
are correlated noise sources and a correlation coefficient factor
should be included in the noise analysis. Further analysis of
correlated noise is limited in this application note as typical
correlation noise sources are less than 10% to 15% and they
can usually be disregarded.
Internal amplifier noise falls into four categories:




Input-referred voltage noise and input-referred current noise
are the most common specifications used for amplifier noise
analysis. They are often specified as an input-referred spectral
density function or the rms noise contained in Δf bandwidth
and usually given in terms of nV/√Hz (for voltage noise) or
pA/√Hz (for current noise). The /√Hz is needed because the
noise power adds with (is cumulative over) bandwidth (Hz) or
the voltage and current noise density adds with square root of
the bandwidth (√Hz) (see Equation 1 and Equation 2).
Output noise from a circuit can be reduced by lowering the
total component resistance or by limiting the circuit bandwidth.
Temperature reduction is generally not very helpful unless a
resistor can be made very cold, because noise power is proportional to the absolute temperature,
T(x) in Kelvin = x°C + 273.15°
Input-referred voltage noise
Input-referred current noise
Flicker noise
Popcorn noise
en
–+
+
RS
–
in
in
(3)
All resistors in a circuit generate noise. The effect of generated
noise must always be considered. In practice, only resistors in
the input and feedback paths (typically in high gain configurations) are likely to have an appreciable effect on total circuit
noise. The noise can be considered as coming from either
current sources or voltage sources (whichever is more convenient in a given circuit).
Rev. D | Page 3 of 12
R2
R1
Figure 1. Op Amp Noise Model
07053-001

AN-940
Application Note
INPUT-REFERRED VOLTAGE NOISE
INPUT-REFERRED CURRENT NOISE
Input-referred voltage noise (en) is typically viewed as a noise
voltage source.
Input-referred current noise (in) is typically seen as two noise
current sources pumping currents through the two differential
input terminals.
Voltage noise is the noise specification that is usually emphasized; however, if input impedance levels are high, current noise
is often the limiting factor in system noise performance. It is
analogous to offsets, where the input offset voltage often bears
the blame for output offset, when in reality the bias current
causes the output offset where input impedances are high.
Note the following points about input-referred voltage noise:




Op amp voltage noise can be lower than 1 nV/√Hz for the
highest performance amplifiers.
Although bipolar op amps traditionally have less voltage
noise than FET op amps, they also have substantially
greater current noise.
Bipolar amplifier noise characteristics are dependent on
the quiescent current.
Present day FET op amps are capable of obtaining both low
current noise and voltage noise similar to bipolar amplifier
performance, though not as low as the best bipolar input
amplifiers.
Shot noise (sometimes called Schottky noise) is current noise
due to random distribution of charge carriers in the current flow
through a potential barrier, such as a PN junction. The shot
noise current, in, is obtained from the formula
in  2 I B qB
(4)
where:
IB is the bias current in ampere (A).
q is the electron charge in coulomb (1.6 × 10−19 C).
B is the bandwidth in hertz (Hz).
The current noise of a simple bipolar and JFET op amp is typically
within 1 dB or 2 dB of the shot noise of the input bias current.
This specification is not always listed on data sheets.
Note the following points regarding input-referred noise:




Rev. D | Page 4 of 12
The current noise of typical bipolar transistor op amps,
such as the OP27, is about 400 fA/√Hz, where IB is 10 nA,
and does not vary much with temperature except for bias,
current-compensated amplifiers.
The current noise of JFET input op amps (such as the
AD8610: 5 fA/√Hz at IB = 10 pA) while lower, doubles
for every 20°C chip temperature increase, because JFET
op amp bias currents double for every 10°C increase.
Traditional voltage feedback op amps with balanced inputs
usually have equal (correlated and uncorrelated) current
noise on both their inverting and noninverting inputs.
Many amplifiers, especially those amps with input bias
current cancellation circuits, have considerably larger
correlated than uncorrelated noise components. Overall,
noise can be improved by adding an impedance-balancing
resistor (matching impedances on both positive and
negative input pins).
Application Note
AN-940
FLICKER NOISE
POPCORN NOISE
The noise of op amps is Gaussian with constant spectral density
(white noise), over a wide range of frequencies. As frequency
decreases, the spectral density starts to rise because of the fabrication process, the IC device layout, and the device type at a
rate of about 3 dB per octave for CMOS amplifiers, 3.5 dB to
4.5 dB per octave for bipolar amplifiers, or up to 5 dB per
octave for JFET amplifiers.
Popcorn noise (not specified or advertised) is an abrupt shift in
offset voltage or current lasting for several milliseconds with
amplitude from several microvolts to hundreds of microvolts.
This burst or pop is random. Low temperatures and high source
resistances usually produce the most favorable conditions for
popcorn noise. Although the root cause of popcorn noise is
not absolute, both metallic contamination and internal or
surface defects in the silicon lattice can cause popcorn noise
in ICs. Although considerable work has been done to reduce
the sources of popcorn noise in modern wafer fabrication, it
cannot be eliminated. Further analysis of popcorn noise is
beyond the scope of this application note.
This low frequency noise characteristic is known as flicker
noise or 1/f noise because the noise power spectral density
goes inversely with frequency (1/f). It has a −1 slope on a log
plot. The frequency at which an extrapolated −3 dB per octave
(for a CMOS-type amplifier) spectral density line intersects the
broadband constant spectral density value is known as the 1/f
corner frequency and is a figure of merit for the amplifier (see
Figure 2). Bipolar and JFET amplifiers typically have lower 1/f
corner frequency than CMOS amplifiers.
100
SUMMING THE NOISE SOURCES
If the noise sources are uncorrelated (that is, one noise signal
cannot be transformed into the other), the resulting noise is
not their arithmetic sum, but the square root of the sum of
their squares.
EXTRAPOLATED 1/f
SPECTRAL NOISE DENSITY
1
EXTRAPOLATED
CONSTANT SPECTRAL
NOISE DENSITY
1/f CORNER FREQUENCY
1
10
100
FREQUENCY (Hz)
Note the following:

1k
10k

Figure 2. Spectral Noise Density
Any resistance in the noninverting input has Johnson noise
and converts current noise to a voltage noise.
Johnson noise in feedback resistors can be significant in
high resistance circuits.
Figure 3 visually shows the Equation 5 as the summation of
vectors by using the Pythagorean Theorem.
Vni, TOTAL
Vn (REX)
RS × in
en
07053-006
0.1
0.1
(5)
where:
Vni, TOTAL is the total noise referred-to-input (RTI).
en is input-referred voltage noise.
in is input-referred current noise.
RS is an equivalent source or input resistance to the amplifier.
Vn (REX) is voltage noise from external circuitry.
10
07053-005
VOLTAGE NOISE (nV/√Hz)
Vni , TOTAL  (e n ) 2  (R S  i n ) 2  Vn (R EX ) 2
Figure 3. Vector Summation of Noise Sources
Rev. D | Page 5 of 12
AN-940
Application Note
OP AMP
NOISE MODEL
COMBINED RTI NOISE
(Vni, TOTAL)
en
–+
+
in
RESISTOR
NOISE RS
–
RESISTOR
NOISE
–
in
R1
R2
+
RESISTOR
NOISE
R1
R2
07053-007
–+
Figure 4. Simplifying the Amplifier Noise Circuit
The noises previously discussed can be grouped into referredto-input (RTI) noise of the amplifier circuit. To calculate the
total output noise of the amplifier circuit, the total combined
noise on the input must be multiplied by the amplifier circuit’s
noise gain. Noise gain is the gain of the amplifier’s circuit for
referred-to-input noise and it is typically used to determine the
stability of the amplifier circuit.
In some cases, the noise gain and the signal gain are not equivalent (see Figure 5). Note that the closed-loop bandwidth is
determined by dividing the gain bandwidth product (or unity
gain frequency) by the noise gain of the amplifier circuit.
NOISE
SOURCE
CASE 1:
SIGNAL
SOURCE
To simplify the noise gain calculation, the noise sources in the
simple amplifier circuit in Figure 1 can be reduced to a single
total RTI noise source (Vni, TOTAL), as shown in Figure 4. It is a
common practice to lump the total combined RTI noise to the
noninverting input of the amplifier.
CASE 2:
SIGNAL
SOURCE
Vno, TOTAL  GN  Vni , TOTAL
where:
Vno, TOTAL is the total referred-to-output (RTO) noise.
Vni, TOTAL is the total referred-to-input (RTI) noise
GN  1 
R1
R2
+
–
R2
R1
07053-008
NOISE GAIN
Figure 5. Signal Gain vs. Noise Gain
Case 1: In a noninverting configuration, both the signal gain
and the noise gain are equal to 1 + R1/R2.
Case 2: In an inverting configuration, signal gain is equal to
−(R1/R2), but the noise gain is still equal to 1 + R1/R2.
where:
GN is the noise gain.
R1 is the feedback equivalent impedance.
R2 is the gain setting equivalent impedance.
Rev. D | Page 6 of 12
Application Note
AN-940
SELECTING LOW NOISE OP AMP
An amplifier can be selected where its noise contribution is
negligible compared to the source resistance by using a figure
of merit, RS, OP, of an op amp. It can be calculated by using an
amplifier’s noise specification.
If an op amp is driven with a source resistance, the equivalent
noise input becomes the square root of the sum of the squares of
the amplifier’s voltage noise, the voltage generated by the source
resistance, and the voltage caused by the amplifiers current
noise flowing through the source impedance.
RS , OP 
For very low source resistances, the noise generated by the
source resistance and amplifier current noise contribute
insignificantly to the total. In this case, the noise at the
input is effectively only the voltage noise of the op amp.
en
in
(7)
where:
en is input-referred voltage noise.
in is input-referred current noise.
If the source resistance is high, the Johnson noise of the source
resistance may dominate both the op amp voltage noise and the
voltage due to the current noise. However, note that, because the
Johnson noise only increases with the square root of the resistance, while the noise voltage due to the current noise is directly
proportional to the input impedance, the amplifier’s current noise
always dominates for a high enough value of input impedance.
When an amplifier’s voltage and current noise are high enough,
there may be no value of input resistance for which Johnson
noise dominates.
Figure 6 shows a comparison of the voltage noise density of
a number of high voltage (up to 44 V) op amps from Analog
Devices, Inc., vs. RS, OP at 1 kHz. The diagonal line plots the
Johnson noise associated with resistance.
100
f = 1kHz
JOHNSON NOISE LINE
OF SOURCE RESISTANCE
OP285
OP467
OP271
10
VOLTAGE NOISE (nV/√Hz)
OPx177
OP275
OP213
OPx84
OP270
OP27/OP37
1
0.1
10
100
1k
10k
SOURCE RESISTANCE (Ω)
Figure 6. Analog Devices Op Amp Noise Plot
Rev. D | Page 7 of 12
100k
1M
07053-002
AN-940
Application Note
2.
Similar types of graph can be constructed for a chosen frequency
from the data in the op amp data sheet (see Figure 8). For example,
the AD8599 has an input-referred voltage noise of 1.07 nV/√Hz
and an input-referred current noise of 2.3 pA/√Hz at 1 kHz. The
RS, OP is about ~465 Ω at 1 kHz. In addition, note the following:

4.
The Johnson noise associated with this device is equivalent
to a source resistor of about 69.6 Ωsee Figure 6).
For a source resistance above ~465 Ω, the noise voltage
produced by the amplifier’s current noise exceeds that
contributed by the source resistance; the amplifier’s
current noise becomes the dominant noise source.
Any amplifiers below and to the right of the lines are good low
noise op amps for the design as highlighted in the shade of gray
in Figure 7.
For the example shown in Figure 7, the following devices are
and OP184.
To use the graph (see Figure 7), follow Step 1 through Step 4.
Typically, the source resistances are known (such as sensor
impedances). If the resistances are not known, calculate them
from the surrounding or preceding circuit components.
100
f = 1kHz
JOHNSON NOISE LINE
LOW NOISE BOUNDRY
IDEAL OP AMPS FOR A
LOW NOISE APPLICATION
OP285
OP467
OP271
10
VOLTAGE NOISE (nV/√Hz)
1.
STEP 2
OP275
OPx177
STEP 3
OP213
OPx84
OP270
OP27/OP37
1
STEP 4
0.1
10
100
1k
10k
SOURCE RESISTANCE (Ω)
Figure 7. Selecting Op Amp for Low Noise Design
Rev. D | Page 8 of 12
100k
1M
07053-003

3.
Locate the given source resistance, such as 1 kΩ, on the
Johnson noise line.
Create a horizontal line from the point located in Step 2 to
the right of the plot.
Create a line down and to the left from the point located in
Step 2) by decreasing one decade of voltage noise per one
Application Note
AN-940
CONCLUSION
Consider all potential noise sources when evaluating an
amplifier’s noise performance for low noise design.
The key noise contribution of an op amp is dependent on
source resistance as follows:



RS >> RS, OP; input-referred current noise dominates.
RS = RS, OP; amplifier noise and resistor noise are equal
RS << RS, OP; input-referred voltage noise dominates.
In summary, reduce or eliminate interference signals by



Proper layout techniques to reduce parasitics.
Proper ground techniques, such as isolating digital
and analog ground.
Proper shielding.
For resistive noise sources, use the following rules:





Restrict bandwidth to only what is necessary.
Reduce resistor value where possible.
Use low noise resistors, such as bulk metal foil, wirewound,
and metal film technology resistors.
Reduce the number of resistive noise sources where
possible.
Use Figure 8 and Figure 9 to assist with the selection of
an Analog Devices low noise amplifier using the criteria
described in this application note.
For more information on noise, see the article, “Noise Optimization in Sensor Signal Conditioning Circuit” available at
http://www.analog.com/noiseoptimization.
Rev. D | Page 9 of 12
PART
NUMBER
OP27
OP37
OP270
OP470
OP184/
OP284/
OP484
OP113 /
OP213/
OP413
SSM2135
OP285
OP275
OP467
OP471
OP1177/
OP2177/
OP4177
OP162/
OP262/
OP462
OP07
OP07D
OP97
OP297
OP497
OP777/
OP727/
OP747
Application Note
VSY
(V)
VOS
MAX
(µV)
TCVOS
(µV/°C)
GBP
(MHz)
SLEW
RATE
(V/µs)
ISY/AMP
MAX
(mA)
en @
1kHz
(nV/√Hz)
1kHz
(pA/√Hz)
in @
RS, OP @
1kHz
(Ω)
1/f
CORNER
(Hz)
IB
MAX
(nA)
ISC
(mA)
CMRR
MIN
(dB)
PSRR
MIN
(dB)
NUMBER
OF
AMPS
10 TO 36
9 TO 36
40
120
0.2
0.8
8
10
20
15
10.5
5.7
0.9
1.07
2
2.3
450
465
60
9
900
200
80
52
120
120
120
120
10 TO 36
125
0.7
12
2.7
2.2
1.8
1.2
1500
5
90
25
110
110
10 TO 36
10 TO 36
10 TO 36
50
75
75
0.2
0.2
0.3
10
10
10
2.5
2.5
4
3.4
2.9
3.5
2.8
2.8
2.8
0.3*
0.3*
0.3*
—
—
—
10
10
10
2
2
12
40
40
30
111
114
100
106
120
110
±4.5 TO ±18
8 TO 44
8 TO 44
9 TO 36
1000
100
100
75
0.3
0.3
0.3
0.2
6.5
8
40
5
12
2.8
17
2.4
2.25
5.7
4.7
3.25
2.8
3.2
3.2
3.2
1.2
0.4
0.4
0.6
2333
8000
8000
5333
5
2.7
2.7
5
100
80
75
20
40
30
30
15
110
100
100
106
106
140
140
110
9.6 TO 36
9.6 TO 36
3 TO 36
1000
500
100
2
2
0.2
4.5
20
4.25
2.8
12.5
4
10
10
2
3.2
3.2
3.9
0.0069
0.0069
0.4
463,768
463,768
9750
50
50
10
0.4
0.25
450
40
40
10
80
90
86
90
100
90
2.7 TO 5.5
250
0.4
28
11
4.5
4
—
—
3000
0.01
220
85
88
4 TO 36
150
0.2
3.4
1.2
3
4.7
0.4
11,750
10
600
40
96
100
4 TO 36
2.2 TO 5.5
9 TO 36
10 TO 27
2000
2.5
250
100
—
0.002
1
0.5
3.5
4
9
25
0.9
0.45
22
60
3
1.7
2.5
3.5
5.2
5.6*
6
6
0.5
0.7*
0.9
0.005
10,400
8000
6667
1,200,000
3
NONE
125
1000
750
0.4
350
0.01
30
30
30
65
87
135
80
90
90
130
85
100
9 TO 44
9 TO 36
9 TO 36
1000
500
200
2
3.5
1
9
28
19
22
170
84
2.5
2.5
7.500
6
6
6.1
1.5
0.8
0.0016
4000
7500
3,812,500
2.24
8
250
350
600
5
14
40
45
80
80
106
85
96
106
1
1/
2
1/
2/
4
2
1
1/
2/
4
2
1
1
2/
4
1
1
1/
2/
4
1/
2
1/
2/
4
2
1
2
1/
2
2
4
1
9 TO 36
5 TO 36
1800
60
4
0.2
6.5
1.3
8
0.7
2.75
0.5
6.5
7.9
0.4
0.2
16,250
39,500
5
10
60
2
10
25
95
120
95
120
9 TO 36
400
1
8
20
2.5
8
—
—
100
0.08
70
86
86
2.7 TO 5.5
350
4
50
41
14
8
0.025
320,000
10000
0.01
80
80
76
2.7 TO 5.5
2500
1.8
24
11
1.5
8
—
—
1000
0.001
120
67
63
2.7 TO 5.5
300
1
10
5
1.2
8
0.01
800,000
500
0.001
80
85
80
2.7 TO 6
2000
1.3
10
5
1.05
8
0.05
160,000
3000
0.001
80
70
80
2.7 TO 12
325
1
15
13
0.8
9.5
0.4
23,750
10
600
30
70
60
6 TO 36
75
0.3
0.6
0.3
4
9.6
0.12
80,000
100
4
30
106
94
8 TO 36
8 TO 36
2.7 TO 5.5
150
130
500
0.5
0.5
1.5
0.6
0.6
24
0.2
0.2
12
1.3
1.3
2
10
10
10
0.074
0.074
0.05
135,135
135,135
200,000
8
8
1000
1
1
0.001
30
30
150
120
120
80
115
115
70
2.7 TO 12
1100
2
8
2.9
1.2
10
0.4
25,000
80
300
70
63
60
5 TO 16
2500
3
4
3.5
1.55
10
0.1
100,000
1000
0.001
140
90
98
4 TO 36
125
0.5
0.56
0.48
0.250
11
0.15
73,333
20
200
40
125
125
5 TO 16
160
4
4
3.5
1.55
12
0.1
120,000
1000
0.001
140
90
95
4 TO 40
75
0.3
0.9
0.2
0.38
14
0.02*
1,166,667
200
0.15
10
110
110
3 TO 36
100
0.3
0.7
0.2
0.35
15
0.13
115,384
20
11
30
110
120
*REFER TO DEVICE DATA SHEET FOR SPECIFICATION CONDITIONS.
Figure 8. Analog Devices Low Input Voltage Noise Amplifier Selection Table
Rev. D | Page 10 of 12
4
1/
2/
4
1/
2/
4
1/
2
1/
2(SD)/
4
1/
2/
4
1(SD)/
2(SD)/
4(SD)
1/
2/
4
1
1
1
1/
2/
4
1/
2
1/
2/
4
2/
4
1/
2/
4
1/
2/
4
1/
2/
4
07053-004
AN-940
AN-940
ISY/AMP
MAX
(mA)
en @
1kHz
(nV/√Hz)
1kHz
(fA/√Hz)
in @
RS, OP @
1kHz
(Ω)
1/f
CORNER
(Hz)
IB
MAX
(pA)
3
0.7
35
0.22
159,090,909
100
5
0.850
16
0.5
35,000,000
-
3.5
3
0.290
27.5
0.5
57,000,000
2
1.9
3*
0.900
16
0.8
200
1
19
84
7.500
6.1
500
5
1
1.8
0.2
30
10 TO 26
100
0.5
25
60
3.500
8 TO 36
2500
4
1.4
3.3
0.220
9.6 TO 36
1000
2
4.5
2.8
10
3.2
6.9
463,768
50
0.4
40
80
90
1
9.6 TO 36
500
2
20
12.5
10
3.2
6.9
463,768
50
0.25
40
90
100
1
9 TO 36
250
5
4
20
2.8
18
10
1,800,000
200
25
25
86
86
1
2.7 TO 6
300
1
10
5
1.2
8
10
800,000
500
1
80
85
80
1/
2/
4
OP282/
OP482
9 TO 36
3000
10
4
9
0.250
36
10
3,600,000
40
100
10
70
110
2/
4
2/
4
1/
2/
4
VSY
(V)
VOS
MAX
(µV)
TCVOS
(µV/°C)
GBP
(MHz)
SLEW
RATE
(V/µs)
10 TO 36
10 TO 26
500
10
5
750
2.5
5
5 TO 26
750
2.5
5 TO 36
1000
8 TO 36
9 TO 36
PART
NUMBER
IOUT
(mA)
CMRR
MIN
(dB)
PSRR
MIN
(dB)
NUMBER
OF
AMPS
0.06
20
90
90
1
1
15*
76
80
1/
2/
4
250
1
12*
90
90
1/
2/
4
20,000,000
90
10
15*
74*
70
1/
2/
4
1.6
3,812,500
250
5
45
106
106
1
1.8
16,666,666
700
10
15
82
86
1
6
5
1,200,000
1000
10
45
90
100
1/
2
36
5
7,200,000
30
50
20
73
74
2/
4
9 TO 36
1000
10
3.5
9
0.250
36
10
3,600,000
40
20
10
70
92
8 TO 36
1700
2
5
20
1.650
16
10
1,600,000
100
40
28
80
82
OP97/
OP297/
OP497
4 TO 40
75*
0.3*
0.9*
0.2*
0.38*
14*
20*
1,166,667*
200*
150
10
110*
110*
1/
2/
4
2.7 TO 5.5
350
4
50
41
14
8
25
320,000
10,000
10
80
80
76
1/
2
2.7 TO 6
500
1.5
24
12
1.3
10
50
200,000
1000
1
150
80
70
1/
2/
4
2.7 TO 6
2000
1.3
10
5
1.05
8
50
160,000
3000
1
80
70
80
1(SD)/
2(SD)/
4(SD)
5 TO 6
160
4
4
3.5
1.55
12
100
120,000
1000
1
140
90
95
1/
2/
4
OP07
6 TO 36
75
0.3
0.6
0.3
4
9.6
120
80,000
100
4000
30
106
94
1
5 TO 36
125
0.5
0.56
0.48
0.250
11
150
73,333
20
200
40
125
125
2/
4
*REFER TO DEVICE DATA SHEET FOR SPECIFICATION CONDITIONS.
Figure 9. Analog Devices Low Input Current Noise Amplifier Selection Table
Rev. D | Page 11 of 12
07053-104
Application Note
AN-940
Application Note
REFERENCES
Analog Devices, Inc., AN-280 Application Note Mixed Signal Circuit Techniques.
Barrow, J., and Paul Brokaw. 1989. “Grounding for Low- and High-Frequency Circuits.” Analog Dialogue. Analog Devices, Inc. (23-3).
Bennett, W. R. 1960. Electrical Noise. New York: McGraw-Hill.
Bowers, Derek F. 1989. “Minimizing Noise in Analog Bipolar Circuit Design.” IEEE Press.
Brockman, Don and Arnold Williams. AN-214 Application Note Ground Rules for High-Speed Circuits. Analog Devices, Inc.
Brokaw, Paul. 2000. AN-202 Application Note An IC Amplifier User’s Guide to Decoupling, Grounding, and Making Things Go Right for a
Change. Analog Devices, Inc. (February).
Brokaw, Paul and Jeff Barrow. AN-345 Application Note Grounding for Low- and High-Frequency Circuits. Analog Devices, Inc.
Bryant, James Bryant and Lew Counts. 1990. “Op Amp Issues–Noise ” Analog Dialogue. Analog Devices Inc. (24–2).
Freeman, J. J. 1958. Principles of Noise. New York: John Wiley & Sons, Inc.
Gupta, Madhu S., ed., 1977. Electrical Noise: Fundamentals & Sources. New York: IEEE Press. Collection of classical reprints.
Hernik, Yuval and Belman, Michael. Linearity and Noise Capabilities of Ultra High Precision Bulk Metal® Foil Resistors. Vishay Intertechnology,
Inc. (February 2010).
Johnson, J. B. 1928. “Thermal Agitation of Electricity in Conductors” (Physical Review 32): 97–109.
Motchenbacher, C. D., and J. A. Connelly. 1993. Low-Noise Electronic Design. New York: John Wiley & Sons, Inc.
Nyquist, H. 1928. “Thermal Agitation of Electric Charge in Conductors” (Physical Review 32): 110–113.
Rice, S.O. 1944. “Math Analysis for Random Noise” Bell System Technical Journal (July): 282–332.
Rich, Alan. 1982. “Understanding Interference-Type Noise.” Analog Dialogue. Analog Devices Inc., (16–3).
Rich, Alan. 1983. “Shielding and Guarding.” Analog Dialogue. Analog Devices Inc. (17–1).
Ryan, Al and Tim Scranton. 1984. “DC Amplifier Noise Revisited.” Analog Dialogue. Analog Devices, Inc., (18–1).
Schottky, W. 1926. “Small-Shot Effect and Flicker Effect.” (Phys. Rev. 28): 74–103.
Van Der Ziel, A. 1954. Noise. Englewood Cliffs, NJ: Prentice-Hall, Inc.
Vishay Intertechnology, Inc. AN0003 Application Note Audio Noise Reduction Through the Use of Bulk Metal® Foil Resistors-“Hear the
Difference”.