AP6502

AP6502
340kHz 18V 2A SYNCHRONOUS DC/DC BUCK CONVERTER
Description
Pin Assignments
( Top View )
The AP6502 is a 340kHz switching frequency external compensated
synchronous DC/DC buck converter. It has integrated low RDSON
BS
1
8
SS
IN
2
7
EN
SW
3
6
COMP
GND
4
5
FB
The AP6502 enables continues load current of up to 2A with
efficiency as high as 95%.
The AP6502 features current mode control operation, which enables
SO-8EP
fast transient response times and easy loop stabilization.
The AP6502 simplifies board layout and reduces space requirements
with its high level of integration and minimal need for external
components, making it ideal for distributed power architectures.
The AP6502 is available in a standard Green SO-8 and SO-8EP
package with exposed PAD for improved thermal performance and is
RoHS compliant.
Figure 1 Package Pin Out
Features
•
VIN 4.7V to 18V
•
•
2A Continuous Output Current, 3A Peak
VOUT Adjustable from 0.925V to 16V
•
Gaming Consoles
•
340kHz Switching Frequency
•
Flat Screen TV Sets and Monitors
•
Programmable Soft-Start
•
Set Top Boxes
•
Enable Pin
•
Distributed power systems
•
Protection
•
Home Audio
ƒ
OCP
•
Consumer Electronics
ƒ
Thermal Shutdown
•
Network Systems
Applications
•
Totally Lead-Free & Fully RoHS Compliant (Notes 1 & 2)
•
FPGA, DSP and ASIC Supplies
•
Halogen and Antimony Free. “Green” Device (Note 3)
•
Green Electronics
Notes:
1. No purposely added lead. Fully EU Directive 2002/95/EC (RoHS) & 2011/65/EU (RoHS 2) compliant.
2. See http://www.diodes.com for more information about Diodes Incorporated’s definitions of Halogen- and Antimony-free, "Green" and Lead-free.
3. Halogen- and Antimony-free "Green” products are defined as those which contain <900ppm bromine, <900ppm chlorine (<1500ppm total Br + Cl)
and <1000ppm antimony compounds.
Typical Application Circuit
100
90
VIN = 5V
EFFICIENCY (%)
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high and low side MOSFETs.
VIN = 12V
80
70
60
50
VOUT = 3.3V
L = 10µH
40
0
0.4
0.8
1.2
1.6
LOAD CURRENT (A)
Efficiency vs. Load Current
AP6502
Document Number: DS35423 Rev. 9 - 2
2
Figure 2 Typical Application Circuit
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AP6502
Pin
Number
Pin
Name
1
BS
2
IN
3
SW
4
GND
5
FB
6
COMP
7
EN
8
SS
EP
EP
Function
High-Side Gate Drive Boost Input. BS supplies the drive for the high-side N-Channel MOSFET switch. Connect
a 0.01µF or greater capacitor from SW to BS to power the high side switch.
Power Input. IN supplies the power to the IC, as well as the step-down converter switches. Drive IN with a 4.7V
to 18V power source. Bypass IN to GND with a suitably large capacitor to eliminate noise on the input to the IC.
See Input Capacitor.
Power Switching Output. SW is the switching node that supplies power to the output. Connect the output LC
filter from SW to the output load. Note that a capacitor is required from SW to BS to power the high-side switch.
Ground (Connect the exposed pad to Pin 4).
Feedback Input. FB senses the output voltage and regulates it. Drive FB with a resistive voltage divider
connected to it from the output voltage. The feedback threshold is 0.925V. See Setting the Output Voltage.
Compensation Node. COMP is used to compensate the regulation control loop. Connect a series RC network
from COMP to GND. In some cases, an additional capacitor from COMP to GND is required. See
Compensation Components.
Enable Input. EN is a digital input that turns the regulator on or off. Drive EN high to turn on the regulator; low
to turn it off. Attach to IN with a 100kΩ pull up resistor for automatic startup.
Soft-Start Control Input. SS controls the soft-start period. Connect a capacitor from SS to GND to set the softstart period. A 0.1µF capacitor sets the soft-start period to 15ms. To disable the soft-start feature, leave SS
floating.
EP exposed thermal pad connect to Pin 4 GND, not applicable in the SO-8 package.
Functional Block Diagram
+
OVP
RAMP
1.1V
E
+
5
100/340 KHz
CLK
-
Logic
0.3 V
8
1
BS
3
-
+
0.925 V
IN
100mΩ
+
SS
2
-
FB
OSCILLATOR
CURRENT
SENSE
AMPLIFIER
+
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Pin Descriptions
+
6uA
ERROR
AMPLIFIER
CURRENT
COMPARATOR
100mΩ
4
COMP
SW
GND
6
+
2.5V
EN OK
-
disable
LOCKOUT
COMPARATOR
IN < 4.10V
IN
EN
+
7
0.9V
-
SHUTDOWN
COMPARATOR
INTERNAL
REGULATORS
5V
Figure 3 Functional Block Diagram
AP6502
Document Number: DS35423 Rev. 9 - 2
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AP6502
Absolute Maximum Ratings (Note 4) (@TA = +25°C, unless otherwise specified.)
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Symbol
Parameter
VIN
Supply Voltage
VSW
Switch Node Voltage
Rating
-0.3 to +20
V
-1.0 to VIN +0.3
V
VBS
Bootstrap Voltage
VSW -0.3 to VSW +6
V
VFB
Feedback Voltage
-0.3 to +6
V
VEN
Enable/UVLO Voltage
-0.3 to +6
V
-0.3 to +6
V
TST
Storage Temperature
-65 to +150
°C
TJ
Junction Temperature
+150
°C
+260
°C
3
250
kV
V
VCOMP
Comp Voltage
Lead Temperature
TL
ESD Susceptibility (Note 5)
HBM
Human Body Model
MM
Machine Model
Notes:
Unit
4. Stresses greater than the 'Absolute Maximum Ratings' specified above may cause permanent damage to the device. These are stress ratings only;
functional operation of the device at these or any other conditions exceeding those indicated in this specification is not implied. Device reliability
may be affected by exposure to absolute maximum rating conditions for extended periods of time.
5. Semiconductor devices are ESD sensitive and may be damaged by exposure to ESD events. Suitable ESD precautions should be taken when
handling and transporting these devices.
Thermal Resistance (Note 6) (@TA = +25°C, unless otherwise specified.)
Symbol
Note:
Parameter
θJA
Junction to Ambient
θJC
Junction to Case
Rating
74
126
16
28
SO-8EP
SO-8
SO-8EP
SO-8
Unit
°C/W
6. Test condition: SO-8:
Device mounted on 1"x1" FR-4 substrate PCB, 2oz copper, with minimum recommended pad layout.
SO-8EP: Device mounted on 1" x 1" FR-4 substrate PC board, 2oz copper, with minimum recommended pad on top layer and
thermal vias to bottom layer ground plane.
Recommended Operating Conditions (Note 7) (@TA = +25°C, unless otherwise specified.)
Symbol
Note:
Min
Max
VIN
Supply Voltage
Parameter
4.7
18
Unit
V
TA
Operating Ambient Temperature Range
-40
+85
°C
7. The device function is not guaranteed outside of the recommended operating conditions.
AP6502
Document Number: DS35423 Rev. 9 - 2
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AP6502
Electrical Characteristics (VIN = 12V, @TA = +25°C, unless otherwise specified.)
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Symbol
Typ
Max
IIN
Shutdown Supply Current
Parameter
VEN = 0V
Test Conditions
Min
0.3
3.0
Unit
µA
IIN
Supply Current (Quiescent)
VEN = 2.0V, VFB = 1.0V
0.6
1.5
mA
RDS(ON)1
High-Side Switch On-Resistance (Note 8)
130
mΩ
RDS(ON)2
Low-Side Switch On-Resistance (Note 8)
130
mΩ
ILIMIT
HS Current Limit
Minimum duty cycle
4.4
A
ILIMIT
LS Current Limit
From Drain to Source
0.9
A
High-Side Switch Leakage Current
VEN = 0V, VSW = 0V,
VSW = 12V
0
AVEA
Error Amplifier Voltage Gain
(Note 8)
GEA
Error Amplifier Transconductance
GCS
COMP to Current Sense
Transconductance
FSW
Oscillator Frequency
ΔIC = ±10µA
VFB = 0.75V
FFB
Fold-back Frequency
VFB = 0V
DMAX
Maximum Duty Cycle
VFB = 800mV
TON
Minimum On Time
VFB
Feedback Voltage
TA = -40°C to +85°C
300
900
0.7
EN Lockout Threshold Voltage
2.2
EN Lockout Hysteresis
INUVVth
INUVHYS
TSD
Note:
V/V
1000
µA/V
2.8
A/V
340
380
fSW
%
130
ns
925
950
mV
0.8
0.9
V
2.5
2.7
V
220
3.80
VIN Under Voltage Threshold Rising
4.05
250
VIN Under Voltage Threshold Hysteresis
kHz
90
1.1
EN Rising Threshold
μA
800
0.30
Feedback Overvoltage Threshold
VEN_Rising
10
V
mV
4.40
V
mV
Soft-Start Current
VSS = 0V
6
μA
Soft-Start Period
CSS = 0.1µF
15
ms
160
°C
Thermal Shutdown (Note 8)
8. Guaranteed by design
AP6502
Document Number: DS35423 Rev. 9 - 2
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AP6502
Typical Performance Graphs (VIN = 12V, @TA = +25°C, unless otherwise specified.)
SHUTDOWN SUPPLY CURRENT (µA)
QUIESCENT SUPPLY CURRENT (mA)
0.074
0.58
0.56
0.54
0.52
0.5
0.48
0
0.064
0.054
0.044
0.034
0.024
0.014
0.004
5
10
15
20
INPUT VOLTAGE (V)
Quiescent Supply Current vs. Input Voltage
0
6.2
5
10
15
20
INPUT VOLTAGE (V)
Shutdown Supply Current vs. Input Voltage
3.33
3.329
6
5.8
OUTPUT VOLTAGE (V)
CURRENT LIMIT (A)
3.328
5.6
5.4
5.2
5
4.8
-60
3.327
VIN = 12V
3.326
3.325
3.324
3.323
3.322
3.321
-40
-20 0
20 40 60 80
TEMPERATURE (C)
Current Limit vs. Temperature
3.32
100
4
10
15
INPUT VOLTAGE (V)
Line Regulation
20
375
0.92
OSCILLATOR FREQUENCY (Khz)
0.918
FEEDBACK VOLTAGE (V)
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0.6
0.916
0.914
0.912
0.91
0.908
0.906
0.904
0.902
0.9
-60 -40
-20 0
20 40 60 80
TEMPERATURE (°C)
Feedback Voltage vs. Temperature
AP6502
Document Number: DS35423 Rev. 9 - 2
100
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370
365
360
355
350
-60 -40
-20 0
20 40 60 80 100
TEMPERATURE (°C)
Oscillator Frequency vs. Temperature
January 2013
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AP6502
90
85
85
80
80
75
75
EFFICIENCY (%)
EFFICIENCY (%)
90
70
VIN = 12V
65
60
55
50
40
VIN = 5V
VIN = 12V
70
65
60
55
50
VOUT = 1.2V
L = 3.3µH
45
0
0.4
0.8
1.2
1.6
LOAD CURRENT (A)
Efficiency vs. Load Current
VOUT = 1.8V
L = 3.3µH
45
2
40
0
0.4
0.8
1.2
1.6
LOAD CURRENT (A)
Efficiency vs. Load Current
2
100
90
EFFICIENCY (%)
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Typical Performance Graphs (cont.) (VIN = 12V, VOUT = 3.3V, (@TA = +25°C, unless otherwise specified.)
80
70
60
VIN = 12V
VOUT = 5V
L = 10µH
50
40
0
0.4
0.8
1.2
1.6
LOAD CURRENT (A)
Efficiency vs. Load Current
AP6502
Document Number: DS35423 Rev. 9 - 2
2
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AP6502
Typical Performance Characteristics
(VIN = 12V, VOUT = 3.3V, L = 10µH, C1 = 22µF, C2 = 47µF, @TA = +25°C, unless otherwise specified.)
Steady State Test 2A
Startup Through Enable_no load
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Steady State Test no load
Time -2µs/div
Startup Through Enable 2A
Time -2ms/div
Load Transient Test 1.0A to 2.0A
Time -100µs/div
AP6502
Document Number: DS35423 Rev. 9 - 2
Time -2µs/div
Shutdown Through Enable_no load
Time -10ms/div
Short Circuit Test
Time -20µs/div
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Time -10ms/div
Shutdown Through Enable 2A
Time -5ms/div
Short Circuit Recovery
Time -20µs/div
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AP6502
Applications Information
Theory of Operation
The AP6502 is a 2A current mode control, synchronous buck regulator with built in power MOSFETs. Current mode control assures excellent
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line and load regulation and a wide loop bandwidth for fast response to load transients. Figure 3 depicts the functional block diagram of
AP6502.
The operation of one switching cycle can be explained as follows. At the beginning of each cycle, HS (high-side) MOSFET is off. The error
amplifier (EA) output voltage is higher than the current sense amplifier output, and the current comparator’s output is low. The rising edge of
the 340kHz oscillator clock signal sets the RS Flip-Flop. Its output turns on HS MOSFET. The current sense amplifier is reset for every
switching cycle.
When the HS MOSFET is on, inductor current starts to increase. The current sense amplifier senses and amplifies the inductor current. Since
the current mode control is subject to sub-harmonic oscillations that peak at half the switching frequency, ramp slope compensation is utilized.
This will help to stabilize the power supply. This ramp compensation is summed to the current sense amplifier output and compared to the
error amplifier output by the PWM comparator. When the sum of the current sense amplifier output and the slope compensation signal
exceeds the EA output voltage, the RS Flip-Flop is reset and HS MOSFET is turned off.
For one whole cycle, if the sum of the current sense amplifier output and the slope compensation signal does not exceed the EA output, then
the falling edge of the oscillator clock resets the Flip-Flop. The output of the error amplifier increases when feedback voltage (VFB) is lower
than the reference voltage of 0.925V. This also increases the inductor current as it is proportional to the EA voltage.
If in one cycle the current in the power MOSFET does not reach the COMP set current value, the power MOSFET will be forced to turn off.
When the HS MOSFET turns off, the synchronous LS MOSFET turns on until the next clock cycle begins. There is a “dead time” between the
HS turn off and LS turn on that prevents the switches from “shooting through” from the input supply to ground.
The voltage loop is compensated through an internal transconductance amplifier and can be adjusted through the external compensation
components.
Enable
Above the ‘EN Rising Threshold’, the internal regulator is turned on and the quiescent current can be measured above this threshold. The
enable (EN) input allows the user to control turning on or off the regulator. To enable the AP6502, EN must be pulled above the ‘EN Lockout
Threshold Voltage’ and to disable the AP6502, EN must be pulled below ‘EN Lockout Threshold Voltage - EN Lockout Hysteresis’
(2.2V-0.22V =1.98V).
External Soft Start
Soft start is traditionally implemented to prevent the excess inrush current.
This in turn prevents the converter output voltage from
overshooting when it reaches regulation. The AP6502 has an internal current source with a soft start capacitor to ramp the reference voltage
from 0V to 0.925V. The soft start current is 6uA. The soft start sequence is reset when there is a Thermal Shutdown, Under Voltage Lockout
(UVLO) or when the part is disabled using the EN pin.
External Soft Start can be calculated from the formula below:
ISS = C *
DV
DT
Where;
ISS = Soft Start Current
C = External Capacitor
DV = change in feedback voltage from 0V to maximum voltage
DT = Soft Start Time
Current Limit Protection
In order to reduce the total power dissipation and to protect the application, AP6502 has cycle-by-cycle current limiting implementation. The
voltage drop across the internal high-side MOSFET is sensed and compared with the internally set current limit threshold. This voltage drop is
sensed at about 30ns after the HS turns on. When the peak inductor current exceeds the set current limit threshold, current limit protection is
activated. During this time the feedback voltage (VFB) drops down. When the voltage at the FB pin reaches 0.3V, the internal oscillator shifts
the frequency from the normal operating frequency of 340kHz to a fold-back frequency of 102kHz. The current limit is reduced to 70% of
nominal current limit when the part is operating at 102kHz. This low fold-back frequency prevents runaway current.
AP6502
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AP6502
Applications Information (cont.)
Under Voltage Lockout (UVLO)
Under Voltage Lockout is implemented to prevent the IC from insufficient input voltages. The AP6502 has a UVLO comparator that monitors
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the input voltage and the internal bandgap reference. If the input voltage falls below 4.0V, the AP6502 will latch an under voltage fault. In this
event the output will be pulled low and power has to be re-cycled to reset the UVLO fault.
Over Voltage Protection
When the AP6502 FB pin exceeds 20% of the nominal regulation voltage of 0.925V, the over voltage comparator is tripped and the COMP
pin and the SS pin are discharged to GND, forcing the high-side switch off.
Thermal Shutdown
The AP6502 has on-chip thermal protection that prevents damage to the IC when the die temperature exceeds safe margins. It implements a
thermal sensing to monitor the operating junction temperature of the IC. Once the die temperature rises to approximately +160°C, the thermal
protection feature gets activated. The internal thermal sense circuitry turns the IC off thus preventing the power switch from damage.
A hysteresis in the thermal sense circuit allows the device to cool down to approximately +120°C before the IC is enabled again through soft
start. This thermal hysteresis feature prevents undesirable oscillations of the thermal protection circuit.
Setting the Output Voltage
The output voltage can be adjusted from 0.925V to 16V using an external resistor divider. Table 1 shows a list of resistor selection for
common output voltages. Resistor R1 is selected based on a design tradeoff between efficiency and output voltage accuracy. For high
values of R1 there is less current consumption in the feedback network. However the trade off is output voltage accuracy due to the bias
current in the error amplifier. R1 can be determined by the following equation:
⎛V
⎞
R1 = R 2 ⋅ ⎜⎜ OUT − 1⎟⎟
⎝ 0.925
⎠
Figure 4 Feedback Divider Network
When output voltage is low, network as shown in Figure 4 is recommended.
Table 1 – Resistor Selection for Common Output Voltages
R1 (kΩ)
R2 (kΩ)
VOUT (V)
5
45.3
10
3.3
26.1
10
2.5
16.9
10
1.8
9.53
10
1.2
3
10
Compensation Components
The AP6502 has an external COMP pin through which system stability and transient response can be controlled. COMP pin is the output of
the internal trans-conductance error amplifier. A series capacitor-resistor combination sets a pole-zero combination to control the
characteristics of the control system. The DC gain of the voltage feedback loop is given by:
A VDC = RLOAD × GCS × A VEA ×
VFB
VOUT
Where VFB is the feedback voltage (0.925V), RLOAD is the load resistor value, GCS is the current sense trans-conductance and AVEA is the
error amplifier voltage gain.
AP6502
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AP6502
Applications Information (cont.)
Compensation Components (cont.)
The control loop transfer function incorporates two poles one is due to the compensation capacitor (C3) and the output resistor of error
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amplifier, and the other is due to the output capacitor and the load resistor. These poles are located at:
fP1 =
fP2 =
GEA
2π × C3 × A VEA
1
2π × C2 × RLOAD
Where GEA is the error amplifier trans-conductance.
One zero is present due to the compensation capacitor (C3) and the compensation resistor (R3). This zero is located at:
fZ1 =
1
2π × C3 × R3
The goal of compensation design is to shape the converter transfer function to get a desired loop gain. The system crossover frequency
where the feedback loop has the unity gain is crucial.
A rule of thumb is to set the crossover frequency to below one-tenth of the switching frequency. Use the following procedure to optimize the
compensation components:
1. Choose the compensation resistor (R3) to set the desired crossover frequency. Determine the R3 value by the following equation:
R3 =
2π × C2 × fc VOUT 2π × C2 × 0.1× fs VOUT
×
<
×
VFB
GEA ×GCS
VFB
GEA × GCS
Where fC is the crossover frequency, which is typically less than one tenth of the switching frequency.
2. Choose the compensation capacitor (C3) to achieve the desired phase margin set the compensation zero, fZ1, to below one fourth of the
crossover frequency to provide sufficient phase margin. Determine the C3 value by the following equation:
C3 >
2
π × R3 × fc
Where R3 is the compensation resistor value.
VOUT
(V)
1.2
1.8
2.5
3.3
5
12
CIN/C1
(µF)
22
22
22
22
22
22
COUT/C2
(µF)
47
47
47
47
47
47
RC/R3
(kΩ)
3.24
6.8
6.8
6.8
6.8
6.8
CC/C3
(nF)
6.8
6.8
6.8
6.8
6.8
6.8
L1
(µH)
3.3
3.3
10
10
10
15
Table 2 – Recommended Component Selection
Inductor
Calculating the inductor value is a critical factor in designing a buck converter. For most designs, the following equation can be used to
calculate the inductor value;
L=
VOUT ⋅ (VIN − VOUT )
VIN ⋅ ΔIL ⋅ fSW
Where ΔIL is the inductor ripple current.
And fSW is the buck converter switching frequency.
Choose the inductor ripple current to be 30% of the maximum load current. The maximum inductor peak current is calculated from:
IL(MAX) = ILOAD +
AP6502
Document Number: DS35423 Rev. 9 - 2
ΔIL
2
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Applications Information (cont.)
Inductor (cont.)
Peak current determines the required saturation current rating, which influences the size of the inductor. Saturating the inductor decreases
the converter efficiency while increasing the temperatures of the inductor and the internal MOSFETs. Hence choosing an inductor with
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appropriate saturation current rating is important.
A 1µH to 10µH inductor with a DC current rating of at least 25% percent higher than the maximum load current is recommended for most
applications.
For highest efficiency, the inductor’s DC resistance should be less than 200mΩ. Use a larger inductance for improved efficiency under light
load conditions.
Input Capacitor
The input capacitor reduces the surge current drawn from the input supply and the switching noise from the device. The input capacitor has
to sustain the ripple current produced during the on time on the upper MOSFET. It must hence have a low ESR to minimize the losses.
The RMS current rating of the input capacitor is a critical parameter that must be higher than the RMS input current. As a rule of thumb,
select an input capacitor which has RMs rating that is greater than half of the maximum load current.
Due to large dI/dt through the input capacitors, electrolytic or ceramics should be used. If a tantalum must be used, it must be surge
protected. Otherwise, capacitor failure could occur. For most applications, a 4.7µF ceramic capacitor is sufficient.
Output Capacitor
The output capacitor keeps the output voltage ripple small, ensures feedback loop stability and reduces the overshoot of the output voltage.
The output capacitor is a basic component for the fast response of the power supply. In fact, during load transient, for the first few
microseconds it supplies the current to the load. The converter recognizes the load transient and sets the duty cycle to maximum, but the
current slope is limited by the inductor value.
Maximum capacitance required can be calculated from the following equation:
ESR of the output capacitor dominates the output voltage ripple. The amount of ripple can be calculated from the equation below:
Vout capacitor = ΔI inductor * ESR
An output capacitor with ample capacitance and low ESR is the best option. For most applications, a 22µF ceramic capacitor will be
sufficient.
ΔIinductor 2
)
2
Co =
2
(Δ V + Vout ) − Vout 2
L(Iout +
Where ΔV is the maximum output voltage overshoot.
AP6502
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AP6502
Applications Information (cont.)
PC Board Layout
This is a high switching frequency converter. Hence attention must be paid to the switching currents interference in the layout. Switching
current from one power device to another can generate voltage transients across the impedances of the interconnecting bond wires and
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circuit traces. These interconnecting impedances should be minimized by using wide, short printed circuit traces.
AP6502SP-13 is exposed at the bottom of the package and must be soldered directly to a well designed thermal pad on the PCB. This will
help to increase the power dissipation. This is not applicable for the AP6502S-13.
External Bootstrap Diode
It is recommended that an external bootstrap diode be added when the input voltage is no greater than 5V or the 5V rail is available in the
system. This helps to improve the efficiency of the regulator. This solution is also applicable for D > 65%. The bootstrap diode can be a low
cost one such as BAT54 or a schottky that has a low Vf.
Figure 7—External Bootstrap
Compensation Components
Recommended Diodes:
Part Number
Voltage/Current
Rating
Vendor
B130
30V, 1A
Diodes Inc
SK13
30V, 1A
Diodes Inc
AP6502
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AP6502
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Ordering Information
Device
Package
Code
Packaging
SP
S
SO-8EP
SO-8
AP6502SP-13
AP6502S-13
Quantity
2500/Tape & Reel
2500/Tape & Reel
13” Tape and Reel
Part Number Suffix
-13
-13
Marking Information
Package Outline Dimensions (All dimensions in mm.)
(1) SO-8EP
Exposed Pad
8
5
E1
1
H
4
F
b
Bottom View
9° (All sides)
N
7°
A
e
D
A1
AP6502
Document Number: DS35423 Rev. 9 - 2
E
45°
Q
4° ± 3°
E0
C
Gauge Plane
Seating Plane
L
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SO-8EP (SOP-8L-EP)
Dim Min Max Typ
A 1.40 1.50 1.45
A1 0.00 0.13
b 0.30 0.50 0.40
C 0.15 0.25 0.20
D 4.85 4.95 4.90
E 3.80 3.90 3.85
E0 3.85 3.95 3.90
E1 5.90 6.10 6.00
e
1.27
F 2.75 3.35 3.05
H 2.11 2.71 2.41
L 0.62 0.82 0.72
N
0.35
Q 0.60 0.70 0.65
All Dimensions in mm
January 2013
© Diodes Incorporated
AP6502
Package Outline Dimensions (cont.) (All dimensions in mm.)
0.254
(2) SO-8
NEW PRODUCT
E1 E
A1
L
SO-8
Dim
Min
Max
A
1.75
A1
0.10
0.20
A2
1.30
1.50
A3
0.15
0.25
b
0.3
0.5
D
4.85
4.95
E
5.90
6.10
E1
3.85
3.95
e
1.27 Typ
h
0.35
L
0.62
0.82
θ
0°
8°
All Dimensions in mm
Gauge Plane
Seating Plane
Detail ‘A’
h
7°~9°
45°
Detail ‘A’
A2 A A3
b
e
D
Suggested Pad Layout (All dimensions in mm.)
(1) SO-8EP
X2
Dimensions
C
X
X1
X2
Y
Y1
Y2
Y1
Y2
X1
Value
(in mm)
1.270
0.802
3.502
4.612
1.505
2.613
6.500
Y
C
X
(2) SO-8
X
Dimensions
X
Y
C1
C2
C1
Value (in mm)
0.60
1.55
5.4
1.27
C2
Y
AP6502
Document Number: DS35423 Rev. 9 - 2
15 of 15
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January 2013
© Diodes Incorporated
AP6502
IMPORTANT NOTICE
NEW PRODUCT
DIODES INCORPORATED MAKES NO WARRANTY OF ANY KIND, EXPRESS OR IMPLIED, WITH REGARDS TO THIS DOCUMENT,
INCLUDING, BUT NOT LIMITED TO, THE IMPLIED WARRANTIES OF MERCHANTABILITY AND FITNESS FOR A PARTICULAR
PURPOSE (AND THEIR EQUIVALENTS UNDER THE LAWS OF ANY JURISDICTION).
Diodes Incorporated and its subsidiaries reserve the right to make modifications, enhancements, improvements, corrections or other changes
without further notice to this document and any product described herein. Diodes Incorporated does not assume any liability arising out of the
application or use of this document or any product described herein; neither does Diodes Incorporated convey any license under its patent or
trademark rights, nor the rights of others. Any Customer or user of this document or products described herein in such applications shall
assume all risks of such use and will agree to hold Diodes Incorporated and all the companies whose products are represented on Diodes
Incorporated website, harmless against all damages.
Diodes Incorporated does not warrant or accept any liability whatsoever in respect of any products purchased through unauthorized sales
channel.
Should Customers purchase or use Diodes Incorporated products for any unintended or unauthorized application, Customers shall indemnify
and hold Diodes Incorporated and its representatives harmless against all claims, damages, expenses, and attorney fees arising out of,
directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized application.
Products described herein may be covered by one or more United States, international or foreign patents pending. Product names and
markings noted herein may also be covered by one or more United States, international or foreign trademarks.
This document is written in English but may be translated into multiple languages for reference. Only the English version of this document is
the final and determinative format released by Diodes Incorporated.
LIFE SUPPORT
Diodes Incorporated products are specifically not authorized for use as critical components in life support devices or systems without the
express written approval of the Chief Executive Officer of Diodes Incorporated. As used herein:
A. Life support devices or systems are devices or systems which:
1. are intended to implant into the body, or
2. support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the
labeling can be reasonably expected to result in significant injury to the user.
B. A critical component is any component in a life support device or system whose failure to perform can be reasonably expected to cause
the
failure of the life support device or to affect its safety or effectiveness.
Customers represent that they have all necessary expertise in the safety and regulatory ramifications of their life support devices or systems,
and acknowledge and agree that they are solely responsible for all legal, regulatory and safety-related requirements concerning their products
and any use of Diodes Incorporated products in such safety-critical, life support devices or systems, notwithstanding any devices- or systemsrelated information or support that may be provided by Diodes Incorporated. Further, Customers must fully indemnify Diodes Incorporated and
its representatives against any damages arising out of the use of Diodes Incorporated products in such safety-critical, life support devices or
systems.
Copyright © 2013, Diodes Incorporated
www.diodes.com
AP6502
Document Number: DS35423 Rev. 9 - 2
16 of 15
www.diodes.com
January 2013
© Diodes Incorporated
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