MIC5190 DATA SHEET (11/05/2015) DOWNLOAD

MIC5190
Micrel
MIC5190
Ultra High-Speed, High-Current Active Filter/LDO Controller
General Description
Features
The MIC5190 is an ultra high-speed linear regulator. It uses
an external N-Channel FET as its power device.
The MIC5190 offers ultra high-speed to cope with the fast
load demands of microprocessor cores, ASICs, and other
high-speed devices. Signal bandwidths of greater than 500kHz
can be achieved with a minimum amount of capacitance
while at the same time keeping the output voltage clean,
regardless of load demand. A powerful output driver delivers
large MOSFETs into their linear regions, achieving ultra-low
dropout voltage.
1.25VIN±10% can be turned into 0.9V ±1% without the use of
a large amount of capacitance.
MIC5190 (0.5V reference) is optimized for output voltages of
below 1.0V.
The MIC5190 is offered in 10-lead 3mm × 3mm MLF™ and
10-lead MSOP-10 packages and has an operating junction
temperature range of –40°C to +125°C.
All support documentation can be found on Micrel’s web
site at www.micrel.com.
• Input voltage range:
VIN = 0.9V to 5.5V
• +1.0% initial output tolerance
• Dropout down to 25mV@10A
• Filters out switching frequency noise on input
• Very high large signal bandwidth >500kHz
• PSRR >40dB at 500kHz
• Adjustable output voltage down to 0.5V
• Stable with any output capacitor
• Excellent line and load regulation specifications
• Logic controlled shutdown
• Current limit protection
• 3mm × 3mm 10-lead MLF™ and MSOP-10 packages
• Available –40°C to +125°C junction temperature
Applications
• Distributed power supplies
• ASIC power supplies
• DSP, µP, and µC power supplies
Typical Application
VCC = 12V
C1
0.01µF
VIN =1.2V
VOUT = 0.9V@7A
IR3716S
MIC5190
IS
OUT
VIN
VCC1
FB
VCC2
PGND
EN
C3
0.01µF
R1
100Ω
R2
125Ω
C2
10µF
SGND
COMP
R3
12.5kΩ
GND
GND
MicroLeadFrame and MLF are trademarks of Amkor Technology, Inc.
PowerPAK is a trademark of Siliconix, Inc.
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
December 2005
1
M9999-120105
MIC5190
Micrel
Ordering Information
FB
Output Output
Part Number
Voltage Current Voltage
Standard
Pb-Free
MIC5190BML MIC5190YML
0.5V
ADJ
ADJ
MIC5190BMM MIC5190YMM
0.5V
ADJ
ADJ
Junction Temp. Range
Package
–40°C to +125°C
–40°C to +125°C
10-pin MLF™
MSOP-10
Pin Configuration
VIN 1
FB 2
SGND 3
VCC1 4
COMP 5
10 IS
VIN 1
9 PGND
FB 2
8 OUT
SGND 3
7 VCC2
VCC1 4
6 EN
COMP 5
MLF™-10 (ML)
10 IS
9 PGND
8 OUT
7 VCC2
6 EN
MSOP-10 (MM)
Pin Description
Pin Number
Pin Name
1
VIN
Input voltage (Current Sense +).
2
FB
Feedback input to error amplifier.
3
SGND
Signal ground.
4
VCC1
Supply to the internal voltage regulator.
5
COMP
Error amplifier output for external compensation.
6
EN
7
VCC2
Power to output driver.
8
OUT
Output drive to gate of power MOSFET.
9
PGND
Power ground.
10
IS
Current sense.
December 2005
Pin Function
Enable (Input): CMOS-compatible.
Logic high = Enable, Logic low = Shutdown. Do not float pin.
2
M9999-120105
MIC5190
Micrel
Absolute Maximum Ratings(1)
Operating Ratings(3)
Supply Voltage (VIN) .................................................. +6.0V
Enable Input Voltage (VEN) ......................................... +14V
VCC1, VCC2 ............................................................... +14V
Junction Temperature (TJ) ................ –40°C ≤ TJ ≤ +125°C
ESD ......................................................................... Note 2
Supply Voltage (VIN) ................................... +0.9V to +5.5V
Enable Input Voltage (VEN) ................................. 0V to VCC
VCC1,VCC2 ............................................... +4.5V to +13.2V
Junction Temperature (TJ) ................ –40°C ≤ TJ ≤ +125°C
Package Thermal Resistance
MLF™ (θJA)(4) ..................................................... 60°C/W
MSOP (θJA) (5) .............................................................. 200°C/W
Electrical Characteristics(6)
TA = 25°C with VIN = 1.2V, VCC = 12V, VOUT = 0.5V; bold values indicate –40°C < TJ < +125°C; unless otherwise specified.
Parameter
Condition
Output Voltage Accuracy
At 25°C
Over temperature range
Output Voltage Line Regulation
Min
VIN = 1.2V to 5.5V
Feedback Voltage
Typ
Max
Units
–1
+1
%
–2
+2
%
–0.1
0.005
+0.1
%/V
0.495
0.5
0.505
V
0.02
0.5
%
Output Voltage Load Regulation
IL = 10mA to 1A
VCC Pin Current (VCC1 + VCC2)
Enable = 0V
40
VCC Pin Current (VCC1 + VCC2)
Enable = 5V
15
20
mA
VIN Pin Current
Current from VIN
10
15
µA
13
30
µA
50
70
mV
25
100
µs
FB Bias Current
Current Limit Threshold
35
Start-up Time
VEN = VIN
Enable Input Threshold
Regulator enable
0.8
Regulator shutdown
Enable Pin Input Current
0.6
0.5
Enable Hysteresis
µA
V
0.2
V
100
mV
VIL < 0.2V (Regulator shutdown)
100
nA
VIH > 0.8V (Regulator enabled)
100
nA
Notes:
1. Exceeding the absolute maximum ratings may damage the device.
2. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
3. The device is not guaranteed to function outside its operating ratings.
4. Per JESD 51-5 (1S2P Direct Attach Method).
5. Per JESD 51-3 (1S0P).
6. Specification for packaged product only.
December 2005
3
M9999-120105
MIC5190
Micrel
Typical Characteristics
0.501
0.5
0.499
0.498
0.499
0.498
12.5
11.5
0.495
9.5
20 40 60 80 100 120
Temp (C)
10.5
0.497
0.496
8.5
0.497
0.496
0.495
-40 -20 0
0.501
0.5
7.5
0.496
0.495
0 1 2 3 4 5 6 7 8 9 10
Output Current (A)
0.504
0.503
0.502
6.5
0.498
0.497
VOUT vs. VCC Voltage
0.505
5.5
0.5
0.499
VOUT vs. Temperature
Vout (V)
0.503
0.502
0.501
Vout (V)
Output Voltage (V)
0.505
0.504
0.503
0.502
4.5
Load Regulation
0.505
0.504
VCC Current
vs. VCCVoltage
0.8
0.5
10
8
6
Feedback Current
vs. Temperature
13.5
40
VCC (V)
13.5
12.5
11.5
10.5
9.5
8.5
7.5
6.5
Enable Time
vs. VCC Voltage
5.5
4.5
50
45
40
35
30
25
20
15
10
5
0
0
-40 -20 0 20 40 60 80 100 120
Temperature ( °C)
45
12.5
VCC Voltage(V)
5
50
11.5
13.5
12.5
11.5
10.5
9.5
8.5
7.5
6.5
4.5
5.5
10
10
55
10.5
11
15
60
9.5
12
20
8.5
13
65
CURRENT LIMIT (mA)
Feedback Current (µA)
14
Current Limit Threshold
vs. Vcc Voltage
7.5
25
9
4
2
0
-40 -20 0 20 40 60 80 100 120
Temperature (°C)
6.5
13.5
12.5
11.5
9.5
10.5
8.5
7.5
6.5
VCC Voltage (V)
CC
15
5.5
4.5
13.5
12.5
11.5
9.5
10.5
8.5
7.5
6.5
5.5
0.3
5.5
0.4
Feedback Current
Voltage
vs. V
Feedback Current (µA)
Input Current ( µA)
ENTH (V)
0.6
0.2
Input Current
vs. Temperature
20
18
16
14
12
0.7
VCC Voltage (V)
Enable Time (µsec)
Enable Threshold
vs. VCC Voltage
4.5
20
18
16
14
12
10
8
6
4
2
0
4.5
Input Current (mA)
Vcc (V)
VCC (V) Voltage
December 2005
4
M9999-120105
MIC5190
Micrel
Functional Characteristics
Disable Transient
ENABLE
(1V/div)
ENABLE
(1V/div)
OUTPUT
(500mV/div)
OUTPUT
(500mV/div)
Enable Transient
TIME (100µs/div)
TIME (10µs/div)
10A Load Transient
December 2005
LOAD CURRENT OUTPUT
(5A/div)
(10mV/div)
LOAD CURRENT
(5A/div)
OUTPUT
(10mV/div)
INPUT
(100mV/div)
INPUT
(100mV/div)
Transient Response
TIME (100µs/div)
5
TIME (100µs/div)
M9999-120105
MIC5190
Micrel
Functional Diagram
VCC1
INTERNAL
VOLTAGE
REGULATOR
50mV
VIN
IS
CURRENT LIMIT
AMPLIFIER
VCC2
EN
ENABLE
OUTPUT
CONTROL
AND
LEVEL
SHIFT
OUT
PGND
FB
0.5V
ERROR
AMPLIFIER
SGND
COMP
Figure 1. MIC5190 Block Diagram
Enable
The MIC5190 comes with an active-high enable pin that
allows the regulator to be disabled. Forcing the enable pin low
disables the regulator and sends it into a low off-modecurrent state. Forcing the enable pin high enables the output
voltage. The enable pin cannot be left floating; a floating
enable pin may cause an indeterminate state on the output.
FB
The feedback pin is used to sense the output voltage for
regulation. The feedback pin is compared to an internal 0.5V
reference and the output adjusts the gate voltage accordingly
to maintain regulation. Since the feedback biasing current is
typically 13µA, smaller feedback resistors should be used to
minimize output voltage error.
COMP
COMP is the external compensation pin. This allows complete control over the loop to allow stability for any type of
output capacitor, load currents and output voltage. A detailed
explanation of how to compensate the MIC5190 is in the
“Designing with the MIC5190” section.
SGND, PGND
SGND is the internal signal ground which provides an isolated ground path from the high current output driver. The
signal ground provides the grounding for noise sensitive
circuits such as the current limit comparator, error amplifier
and the internal reference voltage.
PGND is the power ground and is the grounding path for the
output driver.
Functional Description
VIN
The VIN pin is connected to the N-Channel drain. VIN is the
input power being supplied to the output. This pin is also used
to power the internal current limit comparator and compare
the ISENSE voltage for current limit. The voltage range is
from 0.9V min to 5.5V max.
ISENSE
The ISENSE pin is the other input to the current limit comparator. The output current is limited when the ISENSE pin's
voltage is 50mV less than the VIN pin. In cases where there
is a current limited source and there isn’t a need for current
limit, this pin can be tied directly to VIN. Its operating voltage
range, like the VIN pin, is 0.9V min to 5.5V max.
VCC1, VCC2
VCC1 supplies the error amplifier and internal reference,
while VCC2 supplies the output gate drive. For this reason,
ensure these pins have good input capacitor bypassing for
better performance. The operating range is from 4.5V to
13.2V and both VCC pins should be tied together. Ensure that
the voltage supplied is greater than a gate-source threshold
above the output voltage for the N-Channel MOSFET selected.
Output
The output drives the external N-Channel MOSFET and is
powered from VCC. The output can sink and source over
150mA of current to drive either an N-Channel MOSFET or an
external NPN transistor. The output drive also has short
circuit current protection.
December 2005
6
M9999-120105
MIC5190
Micrel
Applications Information
∆V =
Designing with the MIC5190
Anatomy of a transient response
Output voltage variation will depend on two factors: loop
bandwidth and output capacitance. The output capacitance
will determine how far the voltage will fall over a given time.
With more capacitance, the drop in voltage will fall at a
decreased rate. This is the reason that more capacitance
provides a better transient response for the same given
bandwidth.
Load Current
The measure of a regulator is how accurately and effectively
it can maintain a set output voltage, regardless of the load's
power demands. One measure of regulator response is the
load step. The load step gauges how the regulator responds
to a change in load current. Figure 2 is a look at the transient
response to a load step.
Output Voltage
AC-Coupled
1
∫ idt
C
∆V ↓=
∆V = L
di
V=
The time it takes for the regulator to respond is directly
proportional to its bandwidth gain. Higher bandwidth control
loops respond quicker causing a reduced drop on the supply
for the same amount of capacitance.
1
∫ idt
C
dt
1
BW
Output voltage vs. time
during recovery is
directly proportional to
gain vs. frequency.
1
∫ idt ↓
C
Final recovery back to the regulated voltage is the final phase
of transient response and the most important factors are gain
and time. Higher gain at higher frequency will get the output
voltage closer to its regulation point quicker. The final settling
point will be determined by the load regulation, which is
proportional to DC (0Hz) gain and the associated loss terms.
∆V ↓=
Time
Figure 2. Typical Transient Response
At the start of a circuit's power demand, the output voltage is
regulated to its set point, while the load current runs at a
constant rate. For many different reasons, a load may ask for
more current without warning. When this happens, the regulator needs some time to determine the output voltage drop.
This is determined by the speed of the control loop. So, until
enough time has elapsed, the control loop is oblivious to the
voltage change. The output capacitor must bear the burden
of maintaining the output voltage.
There are other factors that contribute to large signal transient response, such as source impedance, phase margin,
and PSRR. For example, if the input voltage drops due to
source impedance during a load transient, this will contribute
to the output voltage deviation by filtering through to the
output reduced by the loops PSRR at the frequency of the
voltage transient. It is straightforward: good input capacitance reduces the source impedance at high frequencies.
Having between 35° and 45° of phase margin will help speed
up the recovery time. This is caused by the initial overshoot
in response to the loop sensing a low voltage.
Compensation
The MIC5190 has the ability to externally control gain and
bandwidth. This allows the MIC5190 design to be individually
tailored for different applications.
In designing the MIC5190, it is important to maintain adequate phase margin. This is generally achieved by having
the gain cross the 0dB point with a single pole 20dB/decade
roll-off. The compensation pin is configured as Figure 3
demonstrates.
di
dt
Since this is a sudden change in voltage, the capacitor will try
to maintain voltage by discharging current to the output. The
first voltage drop is due to the output capacitor's ESL (equivalent series inductance). The ESL will resist a sudden change
in current from the capacitor and drop the voltage quickly. The
amount of voltage drop during this time will be proportional to
the output capacitor's ESL and the speed at which the load
steps. Slower load current transients will reduce this effect.
∆V = L
di
dt ↑
Placing multiple small capacitors with low ESL in parallel can
help reduce the total ESL and reduce voltage droop during
high speed transients. For high speed transients, the greatest
voltage deviation will generally be caused by output capacitor
ESL and parasitic inductance.
∆V ↓= L
Internal
Error Amplifier
3.42MΩ
Driver
20pF
di
dt
After the current has overcome the effects of the ESL, the
output voltage will begin to drop proportionally to time and
inversely proportional to output capacitance.
∆V ↓= L ↓
December 2005
1
∫ idt
↑C
External
Comp
Figure 3. Internal Compensation
7
M9999-120105
MIC5190
Micrel
This places a pole at 2.3 kHz at 80dB and calculates as
follows.
1
2π × 3.42MΩ × 20pF
FP = 2.32kHz
100
FP =
225
The Dominant Pole
1
Fp =
2 × 3 .42 M × Ccomp
80
180
135
180
60
135
40
90
20
45
0
0
-20
0.01
1
10
100
1000
Frequency (KHz)
90
R LOAD × COUT Pole
20
45
0
0
-45
0.01
0.1
1
10
100
1000
Frequency (KHz)
10000 100000
Figure 6. External Compensation
Frequency Response
It is recommended that the gain bandwidth should be designed to be less than 1 MHz. This is because most capacitors lose capacitance at high frequency and becoming resistive or inductive. This can be difficult to compensate for and
can create high frequency ringing or worse, oscillations.
By increasing the amount of output capacitance, transient
response can be improved in multiple ways. First, the rate of
voltage drop vs. time is decreased. Also, by increasing the
output capacitor, the pole formed by the load and the output
capacitor decreases in frequency. This allows for the increasing of the compensation resistor, creating a higher mid-band
gain.
10000 100000
Figure 4. Internal Compensation
Frequency Response
There is single pole roll off. For most applications, an output
capacitor is required. The output capacitor and load resistance create another pole. This causes a two-pole system
and can potentially cause design instability with inadequate
phase margin. External compensation is required. By providing a dominant pole and zero–allowing the output capacitor
and load to provide the final pole–a net single pole roll off is
created, with the zero canceling the dominant pole. Figure 5
demonstrates placing an external capacitor (CCOMP) and
resistor (RCOMP) for the external pole-zero combination.
Where the dominant pole can be calculated as follows:
100
225
80
180
Gain (dB)
60
Internal
Error Amplifier
3.42MΩ
2 × Rcomp × Ccomp
-20
-45
0.1
Fz =
40
1
Driver
Increasing COUT reduces
the load resistance and
output capacitor pole
allowing for an increase
in mid-band gain.
40
135
90
20
45
0
0
Phase (Deg)
80
Phase (Deg)
225
Gain (dB)
100
Gain (dB)
External Zero
Phase (Deg)
60
20pF
-20
0.01
External
Comp
Figure 5. External Compensation
1
2π × 3.42MΩ × CCOMP
And the zero can be calculated as follows:
1
2π × RCOMP × CCOMP
This allows for high DC gain, and high bandwidth with the
output capacitor and the load providing the final pole.
December 2005
10
100
1000
Frequency (KHz)
10000 100000
This will have the effect of both decreasing the voltage drop
as well as returning closer and faster to the regulated voltage
during the recovery time.
MOSFET Selection
The typical pass element for the MIC5190 is an N-Channel
MOSFET. There are multiple considerations when choosing
a MOSFET. These include:
• VIN to VOUT differential
• Output current
• Case size/thermal characteristics
• Gate capacitance (CISS<10nF)
• Gate to source threshold
CCOMP
FZ =
1
Figure 7. Increasing Output Capacitance
RCOMP
FP =
-45
0.1
8
M9999-120105
MIC5190
Micrel
The VIN(min) to VOUT ratio and current will determine the
maximum RDSON required. For example, for a 1.8V (±5%) to
1.5V conversion at 5A of load current, dropout voltage can be
calculated as follows (using VIN(min)):
RDSON
(V
=
RDSON
(1.71V − 1.5V)
=
RDSON
IN
− VOUT
)
θ JA
TSSOP-8
<950mW
TSSOP-8
<1W
<1.1W
<1.125W
TO-220/TO-263
(D2Pack)
<1.4W
>1.4W
Table 1. Power Dissipation and
Package Recommendation
In our example, our power dissipation is greater than
1.4W, so we’ll choose a TO-263 (D2Pack) N-Channel
MOSFET. θJA is calculated as follows.
θJA = θJC + θCS + θSA
Where θJC is the junction-to-case resistance, θCS is the
case-to-sink resistance and the θSA is the sink-to-ambient
air resistance.
In the D2 package we’ve selected, the θJC is 2°C/W. The
θCS, assuming we are using the PCB as the heat sink, can
be approximated to 0.2°C/W. This allows us to calculate
the minimum θSA:
θSA= θJA– θCS – θJC
θSA= 31°C/W – 0.2°C/W – 2°C/W
θSA= 28.8°C/W
Referring to Application Hint 17, Designing PCB Heat
Sinks, the minimum amount of copper area for a D2Pack
at 28.8°C/W is 2750mm2 (or 0.426in2 ). The solid line
denotes convection heating only (2 oz. copper) and the
dotted line shows thermal resistance with 250LFM airflow. The copper area can be significantly reduced by
increasing airflow or by adding external heat sinks.
Now that we know the amount of power we will be dissipating,
we will need to know the maximum ambient air temperature.
For our case we’re going to assume a maximum of 65°C
ambient temperature. Different MOSFETs have different
maximum operating junction temperatures. Most MOSFETs
are rated to 150°C, while others are rated as high as 175°C.
In this case, we’re going to limit our maximum junction
temperature to 125°C. The MIC5190 has no internal thermal
protection for the MOSFET so it is important that the design
provides margin for the maximum junction temperature. Our
design will maintain better than 125°C junction temperature
with 1.95W of power dissipation at an ambient temperature of
65°C. Our thermal resistance calculates as follows:
θ JA
<850mW
PowerPAK™ SO-8 D-Pack
PD = (VIN – VOUT) × IOUT
PD = (1.89V – 1.5V) × 5A
PD = 1.95W
( ) (
TSOP-6
SO-8
Running the N-Channel in dropout will seriously affect transient response and PSRR (power supply ripple rejection). For
this reason, we want to select a MOSFET that has lower than
42mΩ for our example application.
Size is another important consideration. Most importantly,
the design must be able to handle the amount of power being
dissipated.
The amount of power dissipated can be calculated as follows
(using VIN(max)):
TJ max − TJ ambient
Power Dissipation
PowerPAK™1212-8
IOUT
5A
= 42mΩ
θ JA =
Package
PC Board Heat Sink
Thermal Resistance vs. Area
)
PD
125°C − 65°C
=
1.95W
= 31°C / W
So our package must have a thermal resistance less than
31°C /W. Table 1. shows a good approximation of power
dissipation and package recommendation.
Figure 8. PC Board Heat Sink
Another important characteristic is the amount of gate
capacitance. Large gate capacitance can reduce transient performance by reducing the ability of the MIC5190
to slew the gate. It is recommended that the MOSFET
used has an input capacitance <10nF (CISS).
December 2005
9
M9999-120105
MIC5190
Micrel
Deviations on the input voltage will be reduced by the
MIC5190’s PSRR, but nonetheless appear on the output.
There really is no minimum input capacitance, but it is
recommended that the input capacitance be equal to or
greater than the output capacitance for best performance.
Output Capacitor
The MIC5190 is stable with any type or value of output
capacitor (even without any output capacitor!). This allows
the output capacitor to select which parameters of the regulator are important. In cases where transient response is the
most important, low ESR and low ESL ceramic capacitors are
recommended. Also, the more capacitance on the output, the
better the transient response.
The gate-source threshold specified in most MOSFET data
sheets refers to the minimum voltage needed to fully enhance
the MOSFET. Although for the most part, the MOSFET will be
operating in the linear region and the VGS (gate-source
voltage) will be less than the fully enhanced VGS, it is
recommended the VCC voltage has 2V over the minimum
VGS and output voltage. This is due to the saturation voltage
of the MIC5190 output driver.
VCC1,2 ≥ 2V + VGS + VOUT
For our example, with a 1.5V output voltage, our MOSFET is
fully enhanced at 4.5VGS, and so our VCC voltage should be
greater or equal to 8V.
Input Capacitor
Good input bypassing is important for improved performance. Low ESR and low ESL input capacitors reduce both
the drain of the N-Channel MOSFET, as well as the source
impedance to the MIC5190. When a load transient on the
output occurs, the load step will also appear on the input.
VIN
J1
+VIN
330µF
16V
10µF
10µF
10µF
10Ω
100k
22µF
U1 MIC2198-BML
1µF
25V
J2
EN
CSH
6
2
HSD
EN/UVLO
VSW
IRF7821
11
L1
CSH
VOUT
1.8µH
CDEP134-1R8MC-H
10k
VOUT
1VOUT @10A
0.1µF
10Ω
VOUT
VIN
12
4
CSH
BST
10µF
10
100pF
10Ω
MIC5190
5
8
VOUT
VOUT
3
10k
1
330µF
Tantalum
D2
1N5819HW
OUT
VCC1
VIN
VCC2
ISENSE
7
GND
560pF
11.5k
D1
SD103BWS
VDD
GND
9
100Ω
IRF7821
LSD
FB
COMP
10µF
2.2µF
10V
1µF
FB
COMP
10nF
8.06k
100Ω
12.4k
Figure 9. Post Regulator
December 2005
10
M9999-120105
MIC5190
Micrel
Active Filter
Another application for the MIC5190 is as an active filter on
the output of a switching regulator. This improves the power
supply in several ways.
First, using the MIC5190 as a filter on the output can significantly reduce high frequency noise. Switching power supplies tends to create noise at the switching frequency in the
form of a triangular voltage ripple. High frequency noise is
also created by the high-speed switching transitions. A lot of
time, effort , and money are thrown into the design of
switching regulators to minimize these effects as much as
possible. Figure 9 shows the MIC5190 as a post regulator.
Feedback Resistors
VOUT
IR3716S
MIC5190
R1
FB
R2
COUT
GND
INPUT RIPPLE
(100mV/div)
Figure 10. Adjustable Output
The feedback resistors adjust the output to the desired
voltage and can be calculated as follows:
OUTPUT
(10mV/div)
 R1
VOUT = VREF 1 +

 R2 
VREF is equal to 0.5V for the MIC5190. The minimum output
voltage (R1=0) is 0.5V. For output voltages greater than 1V,
use the MIC5191.
The resistor tolerance adds error to the output voltage. These
errors are accumulative for both R1 and R2. For example, our
resistors selected have a ±1% tolerance. This will contribute
to a ±2% additional error on the output voltage.
The feedback resistors must also be small enough to allow
enough current to the feedback node. Large feedback resistors will contribute to output voltage error.
VOUT = 1V
ILOAD = 10A
TIME (1µs/div)
Figure 11. Ripple Reduction
Figure 11 shows the amount of ripple reduction for a 500 kHz
switching regulator. The fundamental switching frequency is
reduced from greater than 100mV to less than 10mV.
INPUT
(100mV/div)
VERROR = R1× IFB
VERROR = 1kΩ × 12µA
VERROR = 12mV
OUTPUT
(10mV/div)
For our example application, this will cause an increase in
output voltage of 12mV. For the percentage increase,
VERROR
× 100
VOUT
12mV
× 100
VERROR % =
1.5V
VERROR % = 0.8%
LOAD CURRENT
(5A/div)
VERROR % =
By reducing R1 to 100Ω, the error contribution by the feedback resistors and feedback current is reduced to less than
0.1%. This is the reason R1 should not be greater than 100Ω.
Figure 12. 10A Load Transient
The transient response also contributes to the overall AC
output voltage deviation. Figure 12 shows a 1A to 10A load
transient. The top trace is the output of the switching regulator
(same circuit as Figure10). The output voltage undershoots
by 100mV. Just by their topology, linear regulators have the
ability to respond at much higher speeds than a switching
regulator. Linear regulators do not have the limitation or
restrictions of switching regulators which must reduce their
bandwidth to less than their switching frequency.
Applying the MIC5190
Linear Regulator
The primary purpose of the MIC5190 is as a linear regulator,
which enables an input supply voltage to drop down through
the resistance of the pass element to a regulated output
voltage.
December 2005
TIME (100µs/div)
11
M9999-120105
MIC5190
Micrel
lower voltages these parasitic values can easily bump the
output voltage out of a usable tolerance.
Using the MIC5190 as a filter for a switching regulator
reduces output noise due to ripple and high frequency switching noise. It also reduces undershoot (Figure 12) and overshoot (Figure 13) due to load transients with decreased
capacitance.
Circuit Board
Load
INPUT
(100mV/div)
Load
LOAD CURRENT OUTPUT
(5A/div)
(10mV/div)
Load
Switching
Power
Supply
Long Traces
Load
Figure 14. Board Layout
TIME (100µs/div)
Figure 13. Transient Response
But by placing multiple small MIC5190 circuits close to each
load, the parasitic trace elements caused by distance to the
power supply are almost completely negated. By adjusting
the switching supply voltage to 1.2V, for our example, the
MIC5190 will provide accurate 1V output, efficiently and with
very little noise.
Due to the high DC gain (80dB) of the MIC5190, it also adds
increased output accuracy and extremely high load regulation.
Distributed Power Supply
As technology advances and processes move to smaller and
smaller geometries, voltage requirements go down and current requirements go up. This creates unique challenges
when trying to supply power to multiple devices on a board.
When there is one load to power, the difficulties are not quite
as complex; trying to distribute power to multiple loads from
one supply is much more problematic.
If a large circuit board has multiple small-geometry ASICs, it
will require the powering of multiple loads with its one power
source. Assuming that the ASICs are dispersed throughout
the board and that the core voltage requires a regulated 1V,
Figure 14 shows the long traces from the power supply to the
ASIC loads. Not only do we have to contend with the tolerance of the supply (line regulation, load regulation, output
accuracy, and temperature tolerances), but the trace lengths
create additional issues with resistance and inductance. With
Circuit Board
Load
MIC5190
MIC5190
Load
Load
MIC5190
Switching
Power
Supply
MIC5190
Load
Figure 15. Improved Distributed Supplies
December 2005
12
M9999-120105
MIC5190
Micrel
3.15 (0.122)
2.85 (0.114)
DIMENSIONS:
MM (INCH)
4.90 BSC (0.193)
3.10 (0.122)
2.90 (0.114)
1.10 (0.043)
0.94 (0.037)
0.30 (0.012)
0.15 (0.006)
0.15 (0.006)
0.05 (0.002)
0.50 BSC (0.020)
0.26 (0.010)
0.10 (0.004)
6° MAX
0° MIN
0.70 (0.028)
0.40 (0.016)
10-Pin MS0P (MM)
10-Lead MLF™ (ML)
MICREL, INC. 1849 FORTUNE DRIVE SAN JOSE, CA 95131
TEL
+ 1 (408) 944-0800
FAX
USA
+ 1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2004 Micrel, Incorporated.
December 2005
13
M9999-120105