A Simple 12 Vout, 22 W, Off-line Forward Converter Using ON Semiconductor's NCP1027/1028 Monolithic Switcher

AND8489/D
A Simple 12 Vout, 22 W,
Off-line Forward Converter
Using ON Semiconductor's
NCP1027/1028 Monolithic
Switcher
http://onsemi.com
APPLICATION NOTE
Prepared by: Frank Cathell
ON Semiconductor
Introduction
Circuit Operation
Most power supplies with less than 100 W output utilize
a flyback switching topology due to the simplicity and low
cost of the circuit implementation. As with any power
topology there are trade−offs which usually involve circuit
simplicity versus performance and cost. In the case of the
forward converter topology, the trade−off usually involves
the addition of a freewheeling diode and output choke. It
should be noted, however, that this “addition”, depending on
the source and cost of these extra components, can actually
be a “wash” due to the increased output capacitance
generally required for a flyback topology. Flybacks usually
require multiple low impedance output capacitors and even
a small inductor in a pi−filter to minimize output ripple.
Since the forward converter utilizes a choke−capacitor
output filter, the output capacitor requirements are
minimized due to the output choke being the main filtering
element. In addition, the peak−to−average switching
currents in the forward converter can be almost half of that
of an equivalent flyback design which lowers EMI
generation and allows the use of a lower current rated
MOSFET and output rectifier in the overall converter
design.
The low power forward converter in this application note
is intended for use in white goods, E−meters, and low power
communications equipment where low EMI generation and
high efficiency are required. This particular example
provides 12 V at up to 2 A peak output current with an
efficiency of greater than 80% for typical loads over the
entire universal ac input range (90 to 265 Vac).
The ON Semiconductor NCP1027/1028 series of
monolithic switcher is implemented as a single switch
forward converter using a resistor−capacitor−diode (RCD)
reset scheme which allows for a very simple transformer
design (no reset winding) and a maximum duty cycle of up
to 80% at minimum input line. An off−the−shelf slug type
inductor is utilized for the output choke and minimal output
capacitance is required for less than 100 mV of output
ripple. The schematic of the forward converter circuit is
shown in Figure 1. A conducted EMI input filter is
comprised of C1, C2, C3, C10, L1 and L2. L1 and C10
comprise the common mode filter while the remaining
components form the differential mode filter. R1 serves as
an optional inrush limiter during initial power supply
turn−on when C3 and C4 are discharged. The resistor should
be a wire wound or a similar construction that can tolerate
the high joule surge rating that will be present at initial
supply turn−on. Metal film resistors are not recommended
due to eventual transient stress fatigue.
The primary control chip can be either the NCP1028 or
NCP1027. The 1027 version has a built in OVP sensor on the
VCC pin that detects if the chip’s operating VCC becomes
excessive and will latch the controller off under such a
condition. The 1028 does not have this feature and was used
in this design because the VCC for the chip is derived by a
simple peak detector circuit composed of D7, R8 and C13
which is driven by the auxiliary winding on T1. R8 limits the
maximum current to the chip while C9 is the main VCC filter
capacitor. Since the peak value of the “raw” VCC voltage on
C13 varies with input line, an output OVP function is not
usable here, hence, use of the NCP1028. The NCP1027
could be used if indirect output OVP sensing were desired
by changing the auxiliary peak detector circuit to a forward
converter type rectifier with L/C filter network (integrator)
by adding a freewheeling diode after D7 and inserting L4, a
15 mH to 25 mH choke (RF type choke) between the
cathode of D7 and C13. The voltage on C13 would then be
regulated against line changes and would provide a
representative dc analog of the main output voltage. R8
General Specifications
Topology: Single Switch Forward Converter
Input: 90 to 270 Vac (universal input)
Output: 12 V at 2 A max. (22 W continuous)
Output Ripple: less than 100 mV peak−to−peak at full load
Combined Load/Line Regulation: ±2%
Efficiency: 80% minimum from half to full load
Conducted EMI Compliance: EN55022 Level B (average)
Over−temperature and overcurrent protection
© Semiconductor Components Industries, LLC, 2011
April, 2011 − Rev. 2
1
Publication Order Number:
AND8488/D
C1
C2
100nF
”x”
L1
3.9 mH
R2 1M
0.5W
C11
1 nF
R11
2M
R12
2M
C3
2
R9
82K
R10
28K
L2
470 uH
10 uF,
400Vdc
D1−D4
1N4007
C4
1
7
8
4
5
2
U1
NCP1028
(100 kHz)
3
47uF
400V
D5
1N4937
R4
47K,
1/2W 2
C13
2.2uF,
100V
D7
10 uF
25V
+ C9
R8
3.9K,
1/4W
3
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7
9
R13
47
D6B
D6A
MBRS2H100
C6
150uH
L3
3
4
opto
U2
2
1
47
R6
R7
470
Z1
MMSZ5241B
R5
Vtrim
10 ohm
C12 MBRS2H100 1000uF
25V
470pF
C10
1nF
”Y1”
1 nF
C8
T1
0.1
C7
22 Watt, 12 Volt Output NCP1028
Based Forward Converter (R4)
10 4
5
MMSD4148A R14
C5
2.2nF
2 kV
R3
10
1
+
_
12V @ 1.8A
(2A peak)
J2
would then have to be adjusted to provide the proper current
into U1’s VCC pin such that the OVP trip level could be
properly calibrated. The schematic of the VCC/OVP sense
configuration using the NCP1027 is shown in Figure 2. For
Figure 1. Forward Converter Schematic
1. L1 is Coilcraft E3491−AL common mode EMI inductor (3.9 mH)
2. L2 is Coilcraft RFB0810−471L or similar (470 uH, 600 mA)
3. L3 is Wurth #7447709151 (150 uH, 2.5 A)
4. R9 sets slope compensation.
5. R8 value dependent on Vout and Vcc winding turns.
6. Z1 zener sets Vout: Vout = Vz + 0.85V; R5 is optional voltage trim resistor
7. R10 sets brownout level.
8. R1 is optional inrush limiter.
9. U1 recommended with Aavid #580100W00000G clip−on DIP8 heatsink.
10. Crossed schematic lines are not connected.
NOTES:
1.5 A,
250 Vac
90 − 265Vac
100nF
Input
”x”
F1
4.3, 3W
R1
+
J1
AND8489/D
more information on the internal functioning of OVP
sensing in the NCP1027 please see the device data sheet at
http://www.onsemi.com/pub_link/Collateral/NCP1027−D.
PDF.
AND8489/D
D7A
MMSD4148A
are typically less than ±2%. Capacitor C8 helps to filter noise
on the feedback pin and stabilize the overall feedback loop.
Overcurrent and over−temperature protection is inherent
in the NCP1028/1027 monolithic controllers. The current
limit level is set by the MOSFET’s peak current sensing
level of the current mode control circuitry. This will occur
at approximately 2.3 A of output current for this design and
will result in a “hiccup” type of start−stop overcurrent
limiting.
4
5
D7B
L4
25mH
NCP1027
(100 kHz)
U1
2
7
5
+
3
C13
4
8
1
10uF,
50V
Magnetics Design
As with most switching power supplies, the key to
effective performance is the design of the power
transformer. In this application it was desirable to have a
small, yet efficient transformer similar to what a similar
flyback topology would require, but without the reset
winding that single−switch forward converters usually
employ. For this low of power level, a
resistor−capacitor−diode (RCD) type of snubber reset
scheme was chosen for its simplicity and for the fact that it
will allow the converter duty ratio to exceed 50% which
helps with transformer core utilization. Of course there is a
price to pay and that is allowing U1’s internal MOSFET
drain voltage to swing up high enough for full core reset, but
not sufficiently high as to damage the device under high line
transient conditions. A small E25/13/7 type ferrite core (also
known as an EF25) was chosen for minimal space. Using the
standard transformer design equation to calculate the
required primary turns at minimum dc bulk voltage yields:
ǒVpx10 8Ǔ
R8
Vcc
+ C9
10 uF
25V
Figure 2. NCP1027 Implementation with Primary VCC
OVP Sensing
The network of R10, R11, R12, and C11 provides for
brown−out sensing of the rectified mains by monitoring the
level of the dc bulk voltage. The level is set so that the chip
shuts down when the mains is at approximately 75 Vac. The
trip level threshold can be easily adjusted with R10.
Since the NCP1028/1027 controllers utilizes current
mode control, and the duty ratio of the converter can exceed
50%, it is necessary to provide slope compensation to the
internal current sensing to avoid sub−harmonic instabilities
if and when the duty ratio exceeds 50%. This compensation
is provided by R9 and is adjustable depending on the level
of reflected inductor magnetizing current seen by the
controller’s current sense circuit.
The output rectifier/filter stage is a conventional forward
converter rectifier/freewheel diode implementation
consisting of dual diode D6, output choke L3, and main
output capacitor C6. Snubber network C12 and R13
attenuates noise and switching spikes on D6 while capacitor
C7 provides additional high frequency output noise
filtering.
Output voltage sensing and feedback is accomplished via
zener diode Z1 and optocoupler U2 and the associated
circuitry. When the output voltage exceeds the zener voltage
(plus Vf of the optocoupler photo diode) the opto turns on
and establishes control of the feedback pin (pin 4) of U1. R7
is necessary to provide a minimum zener current to avoid
poor regulation effects that could occur if the zener were
operated near the “knee” of its associated transfer curve.
Despite the simplicity, this sensing scheme allows for
adequate line and load regulation and a reasonable unity gain
bandwidth for sufficient 120 Hz output ripple attenuation.
The output voltage can be adjusted by changing the value of
Z1. For 12 V output, a 11 V zener (MMSZ5241B) is used for
Z1, and if necessary, resistor R5 can be adjusted until the
desired output voltage is reached. Temperature and
set−point variation due to zener and optocoupler tolerances
Np +
+
(f
B max
Ae)
ǒ120 V
ǒ100 kHz
+ 100
(eq. 1)
10 8Ǔ
2300 gauss
0.52 cm 2Ǔ
Where Vp is minimum bulk voltage, f is the switching
frequency, Bmax is a reasonable maximum flux density in the
ferrite core, and Ae is the cross sectional area of the core.
Checking the available core bobbin width (~0.5 inches or
12.7 mm) indicates that approximately 25 turns of #28
AWG magnet wire will occupy one full winding layer
comfortably, so four layers will be required for the full
primary. To minimize leakage inductance, we can wind half
of the primary (50 turns over two layers) first, then the
secondary and auxiliary VCC windings, and finally the
second half of the primary. By “sandwiching” the secondary
between the primary halves, the primary−to−secondary
leakage will be minimized. The 12 V secondary turns are
given by:
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3
AND8489/D
Ns + Np
+ 100
100 kHz (8 ms); and dI is the choke ripple current which is
twice the chosen critical choke dc current level (0.6 A).
(V s ) Dvf)
(Vp
D max)
(eq. 2)
ǒ12 V ) 1 VǓ
(100
L+
0.8)
ǒ
+
ƪ(15120) 1)ƫ
V D7
Ǔ
V bulk min
120 V
5
* 12 V
ƫ ǒ Ǔ
8 ms
0.6 A
+ 160 mH
(eq. 4)
>> choose 180 mH, a common value.
A 180 mH, 3 A rated off−the−shelf ferrite inductor from
Coilcraft was chosen for the output inductor.
Now we can calculate the various MOSFET/T1 primary
current components at minimum line:
♦ Reflected primary dc load current: 2 A/5 = 400 mA
(Iout/turns ratio)
♦ Reflected choke magnetizing current: 0.6 A/5 =
120 mA
♦ Transformer magnetizing current: This involves
knowing the transformer’s primary L:
+ 16
Where Vs is the main secondary voltage; Dvf is the
forward drop of diode D6, and Dmax is the maximum duty
ratio of the converter. From wire tables it appears that up to
22 turns of #26 AWG magnet wire (adequate of the average
secondary current) will fit across one layer of the bobbin. In
order to allow for tolerances in the chip’s duty ratio and
efficiency issues, 20 turns were selected for a margin
allowance and to allow end−cuffing of the secondary
winding with Mylar tape for safety insulation issues.
18 turns would have probably also been an acceptable
compromise if more margin were desired.
Since the VCC will be derived from peak charging of C13,
and the VCC in U1 is clamped at approximately 9 V with an
internal zener, the VCC winding turns were ratioed from the
primary to yield approximately 15 V when the line bulk is
at the minimum level (120 Vdc).
Ns(V CC) + V aux )
ƪǒ Ǔ
Lp + core AL value
Np 2
10−3 + 1500
10 −3 + 15, 000 mH or 15 mH
(100)
2
(eq. 5)
This will typically be slightly less due to micro gaps between
the core halves, so choose 12 mH.
Where AL for E25/13/7 core of 3C90 material is 1500 per
manufacturer core specs.
100
So dI = (Vp x dt)/L yields: (120 V x 8 mS)/12,000 mH =
0.080 A or 80 mA.
The peak MOSFET/T1 primary current is then:
(eq. 3)
100
400 mA ) 120 mA ) 80 mA + 600 mA
+ 13.3>>14 turns
(eq. 6)
This value is well within the rated 720 mA minimum peak
current limit level of the NCP1028. Assuming a nominal
duty ratio (D) of 0.55 at 120 Vac input, then the rms primary
current will be the square root of D times the peak current =
0.74 x 600 = 0.44 A. The original selection of #28 AWG
primary wire is sufficient to handle this current with
acceptable temperature rise.
Likewise the T1 secondary rms current for 120 Vac input
and 1.8 A output can also be calculated: 0.74 x 1.8 A =
1.33 A which can also be handled with an acceptable
temperature rise using the #26 AWG wire selected for the
12 V secondary.
Figure 3 shows the final design of the forward converter
transformer T1.
The components for the RCD reset snubber (R4, C5, D5)
were selected based on U1’s internal MOSFET drain voltage
waveforms during worst case line and load conditions and
the primary inductance of T1. Since C5 will resonate with
T1’s primary inductance after MOSFET turn−off, the value
of C5 should be selected such that the resonant frequency of
this L/C network is less than half of the inverter switching
frequency. In this case T1 has a primary inductance of about
12 mH, so with C5 at 2.2 nF, this would result in a resonant
frequency of about 31 kHz. Keeping this resonant frequency
low will minimize resonant voltage peaking of the drain
waveform at turn−off. R4 discharges C5 during each
Assuming a max chip VCC current drain of 1.5 mA, the
approximate value of R8, the VCC current limiting resistor,
can be calculated: (15 V – 9 V)/0.0015 A = 4.0 kW >> use
3.9k.
One other consideration that should be checked is the peak
inverter MOSFET current to make sure it is within the
proper current limit ratings of the NCP1028. Since the
circuit is operating in continuous conduction mode with
respect to the output choke, the peak primary current will be
the sum of three components; the peak load current, the
choke magnetizing current (both reflected through the
transformer turns ratio) and the primary magnetizing current
of the transformer. The magnetizing current represents
stored energy in both magnetic elements and should ideally
be minimized. Unfortunately this would mean a high as
possible inductance for both which is not practical. This
leads us to the output choke design. Assuming a maximum
peak output current of 2 A, the minimum dc current, or
critical current in the choke is typically chosen to be 10% to
20% of the max load current. For this case we choose 15%
which is 0.3 A. This current is the point at light output load
where the choke current just becomes discontinuous. Using
the V = L x dI/dt relationship we can rearrange to find L: L
= V x (dt/dI) where V is the voltage across the choke at low
line (calculated using the transformer turns ratio of 5:1 and
Vout); dt is the maximum on−time of the MOSFET at
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4
AND8489/D
addition, the quasi−resonant effect of the snubber capacitor
C5 interacting with T1’s primary inductance helps to shape
the drain waveform so as to minimize leakage inductance
ringing and spikes, thus reducing EMI issues. This is clearly
demonstrated in the conducted EMI profile (Level B) shown
in Figure 8. This was taken with a load of 1.8 A (22 W).
The output voltage ripple for a 1.75 A load is shown in
Figure 6, with a scale of 100 mV per division vertical.
The efficiency versus load curves are shown in Figure 7
for both 120 and 230 Vac input. Note that the supply
efficiency is at or above 80% from half to full load.
switching cycle so the capacitor can control the voltage level
across T1’s primary so as to adequately allow the proper
volt−second reset conditions for the transformer primary.
Waveforms of the drain voltage and associated primary
current are shown in Figures 4 and 5 for 120 Vac and
230 Vac inputs, respectively, with an output current of
1.75 A. The peak voltages are well within the voltage rating
of the NCP1028 MOSFET drain, and are, in fact, lower than
what would exist if a conventional reset winding were used
on the transformer. The voltage margins were also
completely adequate at a high line of 265 Vac input. In
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5
AND8489/D
MAGNETICS DESIGN DATA SHEET
Project / Customer: ON Semiconductor − 24 watt, 12 vout NCP1028 Fwd Conv
Part Description: 24 watt NCP1027 resonant reset forward conv. xfmr (Rev 3)
Schematic ID: T1
Core Type: EF25 (E25/13/7); 3C90 material or similar
Core Gap: No gap
Inductance: (Primary) 12 mH minimum
Bobbin Type: 10 pin horizontal mount for EF25
Windings (in order):
Winding # / type
Turns / Material / Gauge / Insulation Data
Primary A (1 − 2)
50T of #28HN over 2 layers (25 TPL). Insulate for
1 kV to next winding. Self leads to pins.
Vcc (4 − 5)
14 turns of #28 HN over 1 layer, close wound
and centered in window. Self leads to pins.
Insulate to 3 kV to next winding
12V Secondary (9 − 7)
20 turns of #26 triple insulated wire over one layer.
Self leads to pins.
Primary B (2 − 3)
Same as Primary A. Insulate with tape and self−
leads to pins.
Hipot: 3 kV from primaries & Vcc to secondary for 1 minute.
Lead Breakout / Pinout
Schematic
1
(Top View of Bobbin)
Pri A
2
Pri B
3
4
10 9 8 7 6
9
0.20” pin
separation
(8 places)
12V sec
0.80”
7
1 2 3 4 5
Vcc
5
Figure 3. Forward Converter Transformer Design
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6
AND8489/D
Figure 4. MOSFET Drain Voltage and Current at
120 Vac (1.75 A Load)
Figure 5. MOSFET Drain Voltage and Current at
230 Vac (1.75 A Load)
90
Efficiency @
120 Vac (%)
EFFICIENCY (%)
85
80
Efficiency @
230 Vac (%)
75
70
65
60
Efficiency
vs. Loading
0
0.5
1
1.5
2
LOAD (A)
Figure 6. Output Ripple
Figure 7. Efficiency versus Load Curves
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7
2.5
AND8489/D
dBuV
NCP1028 FWD
12 V @ 1.8 A
80
70
60
EN 55022; Class B Conducted, Quasi−Peak
EN 55022; Class B Conducted, Average
50
40
220V Neutral Average
30
120V Neutral Average
20
10
0
−10
−20
1
10
(Start = 0.15, Stop = 30.00) MHz
1/5/2011 10:19:31 AM
Figure 8. Average Conducted EMI Profile at 120 Vac (blue) and 220 Vac (red) 22 W Output
(Note: lower dashed line is 6 dB margin level for Class B)
BILL OF MATERIALS FOR 12 VOUT, 22 W, NCP1027/1028 FORWARD CONVERTER
Designator
Footprint
Manufacturer
Manufacturer Part
Number
Substitution
Allowed
2 A, 100 V
SMB
ON Semiconductor
MBRS2H100T3G
No
Diode − 60 Hz,
1 A, 800 V
SMA
ON Semiconductor
MRA4007
No
1
Diode − fast
recov
1 A, 600 V
axial lead
ON Semiconductor
1N4937
No
D7
1
Signal diode
100 mA,
100 V
SOD−123
ON Semiconductor
MMSD4148A
No
Z1
1
Zener diode
12 V, 500
mA
SOD−123
ON Semiconductor
MMSZ5241B
No
U2
1
Optocoupler
CTR >/ =
0.5
DIP4 SMD
Vishay or NEC
SFH6156A−4 or
PS2561L−1
Yes
U1
1
Monolithic
Controller
100 kHz
DIP8
ON Semiconductor
NCP1027 or NCP1028
No
C1, C2
2
”X” cap, box
type
100 nF, X2
LS = 15 mm
Rifa, Wima
TBD
C10
1
”Y1” cap, disc
type
1 nF, Y1
LS = 7.5 mm
Rifa, Wima
TBD
C5
1
Ceramic cap,
disc
2.2 nF, 2 kV
5%
LS = 7.5 mm
Rifa, Wima
TBD
C8, C11
2
Ceramic cap,
monolythic
1 nF, 50 V
10%
1206
AVX, Murata
TBD
C7
1
Ceramic cap,
monolythic
100 nF, 50 V
10%
1206
AVX, Murata
TBD
C12
1
Ceramic cap,
monolythic
470 pF,
200 V
5%
1206
AVX, Murata
TBD
C3
1
Electrolytic cap
10 mF,
400 Vdc
10%
LS = 5 mm,
D = 12.5 mm
UCC, Panasonic
TBD
C4
1
Electrolytic cap
47 mF, 400 V
10%
LS = 7.5 mm,
D = 16 mm
UCC, Panasonic
TBD
C9
1
Electrolytic cap
10 mF,
25 Vdc
10%
LS = 2.5 mm,
D = 6.3 mm
UCC, Panasonic
TBD
Qty
Description
Value
D6A,
D6B
2
Schottky diode
D1, 2, 3,
4
4
D5
Tolerance
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8
AND8489/D
BILL OF MATERIALS FOR 12 VOUT, 22 W, NCP1027/1028 FORWARD CONVERTER
Designator
Qty
Description
Value
Tolerance
Footprint
Manufacturer
Manufacturer Part
Number
C13
1
Electrolytic cap
2.2 mF,
100 V
10%
LS = 2.5 mm,
D = 6.3 mm
UCC, Panasonic
TBD
C6
1
Electrolytic cap
1000 mF,
25 V
10%
LS = 5 mm,
D = 12.5 mm
UCC, Panasonic
TBD
R1
1
Resistor, 3W,
Wire wound
4.4 W, 3 W
10%
LS = 7.5 mm,
D = 7 mm
Ohmite, Dale
TBD
R2
1
Resistor, 1/2W,
metal film
1 Meg, 1/2W
10%
Axial lead;
LS=12.5mm
Ohmite, Dale
TBD
R4
1
Resistor, 1/2W
metal film
47k, 1/2W
10%
Axial lead;
LS=12.5mm
Ohmite, Dale
TBD
R3, R14
2
Resistor, 1/4W
SMD
10 W
5%
SMD 1206
AVX, Vishay, Dale
TBD
R6, R13
2
Resistor, 1/4W
SMD
47 W
5%
SMD 1206
AVX, Vishay, Dale
TBD
R5
1
Resistor, 1/4W
SMD
TBD (10 W)
5%
SMD 1206
AVX, Vishay, Dale
TBD
R7
1
Resistor, 1/4W
SMD
470 W
5%
SMD 1206
AVX, Vishay, Dale
TBD
R11,
R12
2
Resistor, 1/4W
SMD
2 MW
5%
SMD 1206
AVX, Vishay, Dale
TBD
R10
1
Resistor, 1/4W
SMD
28k
5%
SMD 1206
AVX, Vishay, Dale
TBD
R9
1
Resistor, 1/4W
SMD
82k
5%
SMD 1206
AVX, Vishay, Dale
TBD
R8
1
Resistor, 1/4W
SMD
3.9k
5%
SMD 1206
AVX, Vishay, Dale
TBD
F1
1
Fuse, TR−5 style
1.5 A
TR−5, LS =
5 mm
Minifuse
L3
1
Heatsink for U1
DIP8 clip−on
Aavid
Aavid 580100W00000G
1
Inductor (output
choke)
150 mH,
2.5 A
5%
1210 SMD (12 x
12 mm)
Wurth
7447709151
(alternate)
180 mH,
2.2 A
5%
Axial lead choke
(1.1”)
Coilcraft
PCH−45X−184LT
L2
1
Inductor (EMI
choke)
470 mH,
600 mA
See Wurth
Drawing
Wurth Magnetics
744772471
L1
1
EMI Inductor
3.9 mH,
See Coilcraft
Drawing
Coilcraft
E3491−AL
T1
1
Transformer
E25/13/7
core
See Mag
Drawing
Wurth Magnetics
750312228
J1, J2
2
Screw Terminal
LS = 0.2”
DigiKey
# 281−1435−ND
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9
Substitution
Allowed
AND8489/D
References:
NCP1028 Data Sheet: http://www.onsemi.com/pub_link/Collateral/NCP1028−D.PDF
NCP1027/NCP1028 Application Notes, Design Notes and Reference Designs:
1. http://www.onsemi.com/PowerSolutions/supportDoc.do?type=AppNotes&rpn=NCP1028
2. http://www.onsemi.com/PowerSolutions/supportDoc.do?type=AppNotes&rpn=NCP1027
3. http://www.onsemi.com/PowerSolutions/supportDoc.do?type=Reference Designs&rpn=NCP1027
4. http://www.onsemi.com/PowerSolutions/supportDoc.do?type=Design Notes&rpn=NCP1027
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