The One-Transistor Forward Converter

AND8039/D
The One−Transistor
Forward Converter
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APPLICATION NOTE
Introduction
modifications, it could be made work as a 110 VAC off−line
power supply.
The one−transistor forward converter is the most
elementary form of transformer−isolated buck converter. It
is typically used in off−line applications in the
100 W − 300 W region. This application note illustrates the
approach one would take to design a high DC input voltage,
one−transistor forward converter. With additional
Description of Operation
A simplified schematic of a one−transistor forward
converter can be seen in Figure 1.
LO
D1
+VIN
+VOUT
+
RESET
WINDING
D2
ISW
+
COUT
GND
CIN
CONTROL
+
VSW
−
GND
Figure 1. Simplified Schematic of a One Transistor Forward Converter
rectifiers are not conducting. Any winding can provide the
reset function, but the higher the voltage on the winding, the
quicker the core will reset. Typically, this is the primary
winding or a separate reset winding of equal turns to the
primary. Current from the reset winding can then be returned
to the input capacitor and reused during the next cycle of
operation.
The typical switch voltage and current can be seen in
Figure 2. When the power switch is ON, the switch sees the
output filter inductor’s current reflected by through the
transformer. The amplitude of the primary current is the
output rectifier current times turns ratio of the transformer
(N1/N2) plus a small amount of magnetization current.
During the power switch OFF time, the switch voltage “flys”
up to about twice the input voltage. During this time, the
reset winding begins to output magnetization current back
to the input capacitor.
One can see a transformer has been placed between the
input voltage and a buck converter output stage. The power
switch (SW) is used to create a rectangular voltage
waveform whose amplitude is the input voltage and its duty
cycle is the controllable variable. The transformer provides
both a step−up or down function and a safety dielectric
isolation between the input line and the output load.
The major restriction of this topology is the maximum
duty cycle must be about 50%. Whenever a core is driven in
a unidirectional fashion, that is, current only being driven
from one direction into the primary, the core must be reset.
Magnetization energy which serves only to reorient the
magnetic domains within the core must be emptied, or else
the core will “walk−up” to saturation after a few cycles. To
do this, one needs to reset the core. Resetting is done by
drawing current from a winding during the period when the
transformer is unloaded, that is, when the power switch and
© Semiconductor Components Industries, LLC, 2005
August, 2005 − Rev. 1
1
Publication Order Number:
AND8039/D
AND8039/D
SWITCH
VOLTAGE
Philips EFD family which yields a very trim, low profile
appearance, but can cost slightly more for the basic
core−bobbin sets. Selecting an approximate core size is done
by appreciating that first the core must have a sufficient core
crossectional area to contain the needed flux density to
transport the power from the primary to the secondary
winding(s). Secondly, there must be enough winding area to
contain the required turns of the needed wire gauges.
Thirdly, for off−line transformers, the core family must have
the ability to meet the minimum creepage and clearance
dimensions of the safety agencies after the transformer is
finished. To begin, one would use an equation like equation
1 which is an artificial quantity derived from the product of
the core crossectional area (Ac) times the winding area(Wa).
RESET
VIN
MAGNETIZATION
CURRENT
SWITCH
CURRENT
Figure 2. Power Switch Waveforms
The output rectification and filter section works
identically to the buck converter. The voltage waveform of
secondary looks like an inverted primary winding waveform
except the zero voltage point is the input voltage point on the
primary waveform. The waveform goes positive when the
power switch is conducting. The output rectifier also
conducts during this time. This presents a unipolar, PWM
rectangular voltage signal to the inductor, just as found in a
typical buck converter. The catch diode then operates when
the power switch and the output rectifier are OFF.
Continuous current is then maintained through the output
filter inductor.
WaAc [ 0.7 (Pout Wd(pri)
108)ńfB max (USA) (eq. 1A)
where: Wd(pri) is the average wire diameter needed to
carry the primary current in cm.
Bmax is the maximum operating flux density in
Gauss (Webers/cm2)
In the MKS system (Europe and the rest of the world)
WaAc [ 0.7 (Pout Wd(pri))ńfB max
(eq. 1B)
where: Wd(pri) is the average wire diameter needed to
carry the primary current in meters (m).
Bmax is the maximum operating flux density in
Teslas (Webers/m2)
The result is in cm4 (eq. 1A) or m4 (eq. 1B). The core
manufacturers usually provide the WaAc for each core size.
The core size can then be chosen and should be as large or
larger than this result. For off−line applications, of which
this is not, one should increase the result by about 20 percent
to accommodate the added insulating tape needed for an
IEC−qualified transformer. Also, a core and bobbin set must
be used that has sufficient creepage (distance over a surface)
and clearance (distance through air) dimensions. For
110 VAC − 220 VAC applications, this is 3.2 mm between
phases, and 8.0 mm between the input and output circuits.
This may be difficult determining the off−line−suitability of
a core and bobbin from its data sheet.
In one−transistor forward converters, the operating flux
density (Bmax) dictates how much magnetization energy,
which is not used, must be released by the core prior to the
next power switch conduction cycle. This is a point of
tradeoff, if Bmax is set too low, then there will be many turns
on the transformer, thus making the transformer larger than
it needs to be. Setting Bmax too high, makes the transformer
smaller, but increases the losses related to the core reset
function. A good point of compromise is to set Bmax at about
25% of Bsat at 100 kHz. This level should be reduced by a
factor of 0.04 per 100 kHz above this frequency. One can
then calculate the turns by:
Design of the One−Transistor Forward Converter
Please refer to the schematic in Figure 5 when Component
designations are mentioned.
Design Specifications:
Input Voltage Range: +140 − +200 VDC
Output Voltage: +28 VDC
Output Current: 0.5 A−4.0 A
Max. Output Ripple Voltage: 30 mV
Predesign Estimates:
Output Power:
Pout(max) = (Vout)(Iout(max)) = 112 W
Peak Input Current:
Ipk ≈ 2.8 Pout/Vin(min) = 2.24 A
Average Input Currents:
Iav(low) = Pout/eff(Vin(max)) = 0.66 A
Iav(hi) = Pout/eff(Vin(min)) = 0.94 A
Design of the Transformer
One begins with the transformer for every switching
power supply design. All of the needed parameters are now
known and it serves as the backbone for the remainder of the
design.
One must first select a core family that will house the
transformer. This is done first by reviewing various core
styles and their attributes. The most common off−line core
is the E−E core, for which there are several variations. The
standard E−E core is based upon the old 50 Hz − 60 Hz
lamination core styles, which are very adequate for most
applications. There are some low−profile styles such as the
Npri [ (Vin(nom)
108)ń4fB max Ac (US)
(webers/cm2)
where: Bmax is in Gauss
Ac is the core crossectional area in cm2
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(eq. 2A)
AND8039/D
low input voltage. Peak rectifying the auxiliary winding in
the forward conduction mode, yields a winding with
3.5 turns. Lets round up to 4 and add a series resistor (about
100 W) and a 18 V zener diode across the auxiliary voltage
filter capacitor to limit the maximum voltage. This will
protect the gate of the power MOSFET.
In this example, an EFD25 core will be used. The primary
turns were calculated to be 41 turns of a #24 AWG. The reset
winding will be 41T of #28 AWG. The secondary is 21 turns
of 2 stands of #22 AWG. The auxiliary winding will be 4
turns of #28 AWG. The primary and reset windings will be
wound first onto the bobbin. Next the auxiliary winding is
wound on top of these windings. Three layers of mylar tape
are applied to provide some degree of dielectric isolation
(not quite IEC), then the secondary winding will be applied
last. A last layer of tape is added to provide some protection
to the outer winding.
A cautious note must be now conveyed, this design
example is a non−isolated, high−voltage input power supply.
It is for example only and cannot be built for sale because it
does not meet the IEC (UL CSA or other) specifications for
dielectric isolation and for creepage (the distance along a
surface). To make this an off−line one transistor forward
converter, the input rectifier bridge, EMI filter, an
opto−isolated feedback circuit, an opto−isolated feedback
circuit and the transformer would have to be built to IEC
specifications.
In the MKS system (Europe and elsewhere)
Npri [ (Vin(nom))ń4fB max Ac
(eq. 2B)
where: Bmax is in Teslas (webers/m2)
Ac is the core crossectional area in m2
This should be viewed as a nominal−minimum turns−count
since adding more turns lowers the operating flux density,
which may be counter−intuitive the average electric−based
engineer.
The reset winding is identical in turns to the primary
winding and usually about 3−4 wire gauges smaller than that
of the primary winding. It is phased oppositely from the
primary so that it can discharge the magnetization energy
when the power switch is off.
The secondary turns needed for this application is found
by realizing that the secondary voltage must provide an
output waveform that will have a volt−time average that will
create the proper output voltage when presented to the L−C
filter. In other words, (DCmaxVout(min)) plus the forward
voltage drop of the output rectifier must be greater than the
DC output voltage. This can be done by:
N sec [ 1.1 Npri (Vout ) Vfwd)ńVin(min) DC max (eq. 3)
where: DCmax is the maximum duty cycle of the system
(<0.5)
Vfwd is the nominal forward voltage drop of the
rectifier.
The 1.1 factor provides a 10% margin in the supply’s low
voltage dropout point and also provides margin for other
variations in the circuit. This secondary should be the main
output which would then serve as the reference winding for
all of the other secondary windings. One should round the
result up to the next integer turn.
When determining any additional secondary winding, one
must account for each of the forward voltage drops of their
respective rectifiers. This can be done by:
Selection of the Power Semiconductors
Power Switch
In one−transistor forward converters, the power switch
will see twice the maximum input voltage plus any spikes
caused winding leakage inductance, and rectifier forward
and reverse characteristics. So the minimum VDSS rating for
the power MOSFET is about:
VDSS(min) + 2 (Vin(max)) ) Vclamp(est) + 450 V
Nsec(n) [ Nsec(1) (Vout(n) ) Vfwd(n))ń(Vsec(1)
) Vfwd(1))
The minimum drain current rating should be greater than
just slightly less than slightly less than the maximum peak
current. This is 2.24 A.
Another major consideration, especially for surface
mount components, is the heat generated by the device. The
RDS(ON) and the drive circuit have the greatest influence on
this. By over−rating the drain current, some reduction in heat
can be realized. This lessens the amount of PCB area needed
to keep the junction temperature of the MOSFET at a
reasonable temperature (about +40− +60°C). A reasonable
estimation of the maximum RDS(on) assuming a heatsink
area of twice the minimum footprint area is:
(eq. 4)
The accuracy of each of the output voltages must now be
considered. Some variation can be gotten by changing the
output rectifier technology, otherwise the turns can be
adjusted by raising the reference secondary winding by a
turn and adjusting the other windings. This is an iterative
process done until the output voltages are within an
acceptable tolerance and all of the windings are integer
turns.
This design example only has one output voltage. The
auxiliary winding which provides power to the control IC,
need not be regulated or accurate. It needs to only exceed the
low voltage inhibit limit of the UC3845 which is 8.0 V at the
RDS(on)(max) + 3.3 (DT)ń(Iin(av))2(q (jA))
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(eq. 6)
AND8039/D
This results in a maximum RDS(ON) of 3.5 W. So a summary
of the MOSFET ratings are:
VDSS > 450 V
ID > 2.24 A
RDS(on) < 3.5 W
To further reduce the heat, an MTB8N50E was chosen.
Output Rectifier
The output rectifier will also be a surface mount D2PAK.
This efficiently couples the heat to the copper pad on the
PCB.
The maximum reverse voltage is:
T1
N1
D3
Vr(min) + Vin(max)(N sec ńNpri) u102 V
The Peak output current is:
Iout(pk) + 2.8 Iout(max) or 11.2 A
The selected rectifier is the MURB1620CT.
Design of the Output Filter Section
As in all forward−mode converters, the output is
converted back to DC by the use of an L−C filter. A
two−stage filter is going to be used which is a much more
efficient output filter than a single stage filter. The
abbreviated schematic is shown in Figure 3.
L1
L2
+VOUT
+
N2
+
C9
C10
+
C11
GND
Figure 3. Schematic of the Two−Stage Output Filter
supply. So the common method of calculating the value of
the output filter capacitance is by the ripple−reduction
function. Assuming a very benign load (resistor) and so that
only the ripple is considered, one then calculates:
Below the voltage feedback crossover frequency
(fxo ≈ about 8.0 kHz) all of the output capacitors appear to
be essentially in parallel (i.e., C9, C10 and C11). The first
stage inductor should be calculated such that it does not
enter the discontinuous−mode at light load. The
second−stage filter has its corner frequency at about 22 kHz
and provides an additional 15 dB − 20 dB of ripple
attenuation with little additional phase lag and no
additional output capacitance.
The first stage inductor should be sized to allow 20% of
the AC ripple current through to the capacitor. This is a little
more than is typically allowed, but the existence of the
second filter provides a more pronounced effect, thus
allowing the first filter to be smaller.
Lo [ (Vsec(min) * Vout) toff(min)ń1.4 Iout(min)
Co + Iout(max) (1 * DC max)ńVripple
(eq. 5)
where: Vripple is the desired p−p ripple voltage on the
output.
This results in a total output capacitance of 533 uF. If one
allocates about one−third of this value to the first−stage filter
and two−thirds to the output, and rounding−up to the next
standard value, one gets C9, C10 and C11 as 220 mF,
50 VDC or Nichicon Part number EVR2E470MPA which
has a 430 mArms ripple current rating.
The second−stage filter inductance is determined by
setting its pole above the crossover frequency of the closed
feedback loop so that it will not contribute significant
additional phase shift, but will further reduce the ripple
voltage. If we set the output filter’s filter pole at no more than
25 percent of the switching frequency and at least three times
the filter pole of the first−stage filter, then the nominal corner
frequency of the second−stage filter is around
20 kHz − 25 kHz. The second−stage filter inductor can then
be found by:
(eq. 4)
where: Vsec(min) is 1.1 Vin(min)(Ns/Npri)
The resulting minimum inductance is 88 uH. Lets round this
up to 100 mH which will give us a more standard value
off−the−shelf inductor and extend the minimum current
capabilities of the supply. Now one must choose an inductor
whose core can be driven with 4+ A on its winding without
the fear of core saturation. Coiltronics P/N CTX100−2−52.
Next the output filter capacitor is calculated. In
forward−mode converters, the roles of the output capacitor
are transient hold−up voltage and output ripple reduction.
The output filter inductor greatly reduces the RMS ripple
current to the output capacitor(s) thus relaxing their ratings
somewhat. The transient load holdup function is typically
shared with other filter capacitors outside of the power
Lo(2) + (2pfp)2ń(C10 ) C11)
(eq. 6)
Setting the second−stage filter pole at 22 kHz, the resulting
second−stage inductor value is 0.1 mH. This can easily be
done as an air−core inductor or a spiral PCB inductor, which
is what I will do.
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AND8039/D
Design of the Primary Current Sensing Network
Design of the Voltage Feedback and Compensation
The UC3845, current−mode control IC is being used. Its
current sensing input has a maximum trip voltage of 1.0 V
when the current−mode circuit is just starting−up. To
minimize the losses associated with the current sensing
resistance, one should use about a trip voltage of between
0.3 V and 0.4 V. This results in a current sensing resistor of:
Design of the Resistor Divider
The UC3845 has a 2.5 volt reference. One should set the
value of the top resistor of the resistor divider (R11) between
2.0 k to 15 k W. This then makes the other values in the
compensation network reasonable values. This can be done
by selecting the sense current, that is the current allowed to
flow through the resistor divider. As an estimate one can first
calculate:
Rsc (R8) t VtripńIpk(max) + 0.3ń2.24 A + 0.13 W
make this value 0.1 W for a convenient off−the−shelf value.
A spike filter should be placed between the current
sensing resistor (R8) and the IC. The time constant of this
R−C filter, if set too long, will enter a pulse−skipping mode
at light loads. If its time constant is made too short, then
some spikes may still enter the current comparator and
produce erratic pulse widths. A time constant of 300 nS is a
good time.
One must first select one of the values. By making the R
larger, one can provide some series protection between the
power switch and the input pin of the IC. I will assign a value
of 1.0 K to R7. The capacitor then becomes:
Isense + (28 V * 2.5 V)ń7.0 Kohms + 3.65 mA
Using that sense current the lower resistor (R5) then
becomes:
R5 + 2.5 Vń3.65 mA + 684 ohms
* closest resistance 680 ohms
The upper resistor is then:
R11 + (28.0 V * 2.5 V)ń3.65 mA) + 6986 ohms
or 6.98 kohms 1%
Design of the Feedback Loop Compensation
This is a current−mode controlled, forward converter
where only a 1−pole, 1−zero method of compensation is
required (2 poles if the op amp compensation is considered).
This provides maximum of +90 degrees phase boost, which
helps in avoiding unstable operation.
Determining the Control−to−Output Characteristic
The gain at DC for this topology is:
C7 + 300 nSń1.0 kW + 300 pF
Design of the Bootstrap Start−Up Circuit
The purpose of this circuit is to initially start the control
circuit up from a turned−off state. The control circuit then
would draw its power directly from the transformer. The
most efficient circuit cuts off its start−up current after the
power supply has begun steady−state operation. This
reduces an unnecessary loss.
The circuit seen in the schematic (Figure 5) is essentially
a current−limited, high−voltage, linear regulator. When the
auxiliary power supply from the transformer is less than
10 V, the startup circuit is operational. When the auxiliary
supply exceeds 10 V, it cuts off its collector current, which
is about 1.0 mA. A 10 mF or greater capacitor (C2) must be
placed on the auxiliary bus to store enough energy to
actually start the supply, since the IC will draw about 10 mA
in the operate mode.
R1 = (Vin(min) − Vz)/1.0 mA = (140−12)/1.0 mA
= 128 K Make 120 K
R2 = (Vin(min) − Vz)/2.0 mA
= 64 K Make 62 K
The zener diode (Z1) is a 500 mW 12 V, 1N5242
The selection of high voltage bipolar small signal
transistors is limited. An MPSW42 works nicely for Q1. The
purpose of D1 is to avoid stressing the base−emitter junction
in the reverse direction, if the auxiliary voltage goes far
above the +12 V base voltage. The typical reverse
breakdown voltage (V(BR)EBO) is between 3.0 V − 6.0 V. A
1N4148 is going to be used for D1.
ADC + ƪ(Vin * Vout)2ńVinVeƫ (Nsec ńNpri)
+ 13.5
GDC + 20 Log (ADC)
+ 22.6 dB
The output filter pole is:
ffp + 1ń(2pRLCo)
+ 4.3 Hz (light load (0.5 A))
+ 34.5 Hz (rated load (4.0 A))
where: RL is the equivalent resistance of the load
(Vout/Iout)
Co is the net value of the output capacitance
(C9+C10+C11)
The ESR zero of the net output capacitance is:
fz(esr) + 1ń(2pResrCo)
+ 1ń(2p(50 m ohms) (660 mF))
+ 4822 Hz
where: Resr is all of the ESR resistances in parallel.
Calculating the Compensation Elements
Locating the compensating breakpoints:
fez + ffp(light load) + 4.3 Hz
fep + fp(esr) + 4.8 kHz
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AND8039/D
The crossover frequency will be set at about 8.0 kHz. To
accomplish this, one assumes that the eventual closed loop
bode gain response of the system will be −20 dB/decade
continuous slope. Then one can calculate the amount of
mid−band gain that the error amplifier must provide to
“push−up” or “lower” the gain function so that the crossover
frequency is set at 8.0 kHz. This is done by:
converting this value to absolute gain for later use:
Axo + 10[Gxoń20] + )8.13
Now one can begin to calculate the actual error amplifier
feedback component values.
C5 + 1ń2pfxoAxoR11 + 360 pF
R4 + AxoR11 + 56 K ohms
Gxo + 20Log(fp(esr)ńfxo) ) Gdc + 18.2 dB
C6 + 1ń2pfezR4 + 0.56 mF
+80
+60
+40
Control−to−Output
Closed Loop
GAIN (dB)
+20
Light Load
0
1.0
10
100
1.0K
10K
100K
−20
−40
−60
−80
Control−to−Output
0
Rated Load
Light Load
PHASE (°)
−90
−180
Error Amp
Rated Load
−270
−360
Figure 4. Compensation Bode Plots for the Example
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1M
AND8039/D
TP5
R13 C12
L1
10
+VIN
R1
10
+VOUT
1, 2
R2
RESET
WINDING
+
D3
9
C1
D1
+
C2
8
D5
Z2
+
C9
C10
+
C11
GND
6
Vaux
R12
GND
+
4, 5
Vaux
Z1
L2
T1
D4
TP2
VCC
REF
C3
8
R3
OSC
7
7
U1
UC3845
R9
3
R10
C8
4
C4
TP6
Q2
TP7
TP3
1
2
5
C5 VFB
COMP
C6
TP4
R8
C7
R11
R8A
R4
R5
TP1
Figure 5. 112 Watt, One−Transistor Forward Converter
Conclusion
design for the “real world” one should also include:
dielectric isolation from the input to output, an input
rectification and filter section and some additional methods
of protection.
This application note illustrated the design steps needed
to complete a one−transistor forward converter. This
demonstration unit is only for instruction and to complete a
BILL OF MATERIAL
Designator
Part Number
Manufacturer
Ratings
Description
C1
UVR2E470MPA
Nichicon
250 V
47 mF. Electrolytic
C2*
UMA1E100MDA
Nichicon
25 V
10 mF, Tantalum
C3
MR055C105JAA
AVX
50 V
0.1 mF, Ceramic
C4
MR055C102JAA
AVX
50 V
1000 pF, Ceramic
C5
MR055C361JAA
AVX
50 V
360 pF, Ceramic
C6
MR055C564JAA
AVX
50 V
0.56 mF, Ceramic
C7
MR055C301JAA
AVX
50 V
300 pF, Ceramic
C8*
68Q101MDAAA
AVX
500 V
100 pF Ceramic
C9
UVR1H221MPA
Nichicon
50 V
220 mF, Electrolytic
C10
UVR1H221MPA
Nichicon
50 V
220 mF, Electrolytic
C11*
UVR1H221MPA
Nichicon
50 V
220 mF, Electrolytic
*Snubber components − values to be assigned at prototyping
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7
AND8039/D
BILL OF MATERIAL
Designator
Part Number
Manufacturer
Ratings
C12
68Q101MDAAA
AVX
500 V
Description
D1*
1N4148
ON Semiconductor
200 V, 0.1 A
Signal Diode
D2
1N4148
ON Semiconductor
200 V, 0.1 A
Signal Diode
D3*
MURB1620CT
ON Semiconductor
200 V, 16 A
Dual Ultrafast Rectifier
D4
1N4148
ON Semiconductor
200 V, 0.1 A
Signal Diode
J1*
570−500
Deltron
Banana Socket−Black
J2
570−500
Deltron
Banana Socket−Red
J3*
570−500
Deltron
Banana Socket−Black
J4
570−500
Deltron
Banana Socket−Red
L1
CTX100−5−52
CoilTronics
100 uH, 6 A
Inductor
Q1
MPSW42
ON Semiconductor
300 V, 0.1 A
Small Signal Bipolar
HV Power MOSFET
100 pF Ceramic
Q2
MTB8N50E
ON Semiconductor
500 V, 8 A
R1*
OK1245R52
Ohmite
120 K
Resistor, 1/4 W
R2*
OK6235R52
Ohmite
62 K
Resistor, 1/4 W
R3
OK1535R52
Ohmite
15 KΩ
Resistor, 1/4 W
R4
OK5635R52
Ohmite
56 KΩ
Resistor, 1/4 W
R5
OK6815R52
Ohmite
680 Ω
Resistor, 1/4 W
R6
OK1015R52
Ohmite
100 Ω
Resistor, 1/4 W
R7
OK1015R52
Ohmite
100 Ω
Resistor, 1/4W
R8*
RWR100
Ohmite
0.1 Ω
Resistor, Wirewound
R9
OK1560R52
Ohmite
56 Ω
Resistor, 1/4 W
R10
OK2005R52
Ohmite
20 Ω
Resistor, 1/4 W
R11*
MK6981F
Ohmite
6.98 KΩ
R12
OK1015R52
Ohmite
100 Ω
Resistor, 1/4 W
R13
OK1015R52
Ohmite
100 Ω
Resistor, 1/4 W
T1*
N34356
Cramer Magnetics
Transformer−Custom
U1
UC3845BN
ON Semiconductor
Controller IC
Z1
1N5242B
ON Semiconductor
12 V, 500 mW
Zener Diode
Z2
1N5248B
ON Semiconductor
18 V, 500 mW
Zener Diode
Resistor, 1/4 W, 1%
*Snubber components − values to be assigned at prototyping
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
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