High Safety GreenLine¿ PWM Controller for (Multi) Synchronized Applications

MC44605
High Safety, Latched Mode,
GreenLinet PWM Controller
for (Multi) Synchronized
Applications
The MC44605 is a high performance current mode controller that is
specifically designed for off−line converters. This circuit has several
distinguishing features that make it particularly suitable for
multisynchronized monitor applications.
The MC44605 synchronization arrangement enables operation from
16 kHz up to 130 kHz. This product was optimized to operate with
universal mains voltage, i.e., from 80 V to 280 V, and its high current
totem pole output makes it ideally suited for driving a power MOSFET.
The MC44605 protections enable a well−controlled and safe power
management. Four major faults while detected, activate the analogic
counter of a disabling block designed to perform a latched circuit
output inhibition.
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MARKING
DIAGRAM
PDIP−16
P SUFFIX
CASE 648
A
WL
YY
WW
G
• This is a Pb−Free Device*
Current Mode Controller
Current Mode Operation up to 250 kHz Output Switching Frequency
Inherent Feed Forward Compensation
Latching PWM for Cycle−by−Cycle Current Limiting
Oscillator with Precise Frequency Control
Externally Programmable Reference Current
Secondary or Primary Sensing (Availability of Error Amplifier Output)
Synchronization Facility
High Current Totem Pole Output
Vcc Undervoltage Lockout with Hysteresis
Low Output dV/dT for Low EMI Radiations
Low Startup and Operating Current
•
•
•
•
Soft−Start Feature
Demagnetization (Zero Current Detection) Protection
Overvoltage Protection Facility against Open Loop
EHT Overvoltage Protection (E.H.T.OVP): Detection of too High
Synchronization Pulses
Winding Short Circuit Detection (W.S.C.D.)
Limitation of the Maximum Input Power (M.P.L.): Calculation of
Input Power for Overload Protection
Overheating Detection (O.H.D.): to Prevent the Power Switch from
an Excessive Heating
Safety/Protection Features
•
•
•
Latched Disabling Mode
MC44605P
AWLYYWWG
1
1
Features
•
•
•
•
•
•
•
•
•
•
•
16
= Assembly Location
= Wafer Lot
= Year
= Work Week
= Pb−Free Package
PIN CONNECTIONS
VCC
1
16 Rref
VC
2
15 WSCD* Program
Output
3
14 Voltage Feedback Input
GND
4
13 Error Amp Output
Max Power Limitation
5
12 Disabling Block (Cext)
Overheating
Detection
Current Sense Input
6
11 Soft-Start Input
7
10 Osc Capacitor (CT)
Demagnetization
Detection Input
8
9 Sync and
EHTOVP Input
(Top View)
*Winding Short Circuit Detection
ORDERING INFORMATION
Device
Package
Shipping
MC44605PG
PDIP−16
(Pb−Free)
25 Units/Rail
• When one of the following faults is detected: EHT overvoltage,
•
Winding Short Circuit (WSCD), a too high input power (M.P.L.),
power switch overheating (O.H.D.), an analogic counter is activated
If the counter is activated for a time that is long enough, the circuit
gets definitively disabled. The latch can only be reset by making
decrease the Vcc down to about 3.0 V, i.e., practically by unplugging
or turning off the SMPS.
*For additional information on our Pb−Free strategy and soldering details, please download the ON Semiconductor Soldering and Mounting
Techniques Reference Manual, SOLDERRM/D.
© Semiconductor Components Industries, LLC, 2009
July, 2009 − Rev. 6
1
Publication Order Number:
MC44605/D
MC44605
Block Diagram
R ref
V CC
16
1
i ref
V ref
Demagnetization
Detection Input 8
Demagnetization
Management
V demag out
V DT
Output
C T 10
Synchronization
and EHTOVP 9
Input
2 VC
Vcs
−
4 Gnd
V ref
Thermal
Shutdown
V shift
Sf
dis MPL
V CC
enable
I ref
dis OHD
V cs 2
MPL
block
O.H.D.
block
5
6
Soft−Start
E/A Output 13
7
*W.S.C.D. = Winding Short Circuit Detection
UVLO2
15
dis OHD
Disout
Error
AMP
V shift Level
Programmation
Vcs
dis MPL
V CC
Over Voltage
Management
V WSCD
I sense
Voltage
Feedback 14
Input
+
Current
Sense
3 Output
Buffer
Set
Q
PWM
Latch
Reset
W.S.C.D*
Comparator
Disabling
Block
V ref
Disout
VS
I sense
E.H.T.OVP
Block
18 V
UVLO1
UVLO2
I ref
Sf
C ext 12
Supply
Initialization
Block
Reference
Block
Oscillator
V CC
V cc enable
11
Current Maximum
Over
Soft−Start
Heating
Input
Power
Sense
Input Limitation Detection
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2
UVLO1
V CC enable
MC44605
WSCD
Programmation
MC44605
MAXIMUM RATINGS
Rating
Pin #
Symbol
Value
Unit
(ICC + IZ)
40
mA
VC
VCC
18
V
IO(Source)
IO(Sink)
−750
750
W
5.0
J
Soft−Start
VSS
−0.3 to 2.2 V
V
Current Sense, Voltage Feedback, E/A Output, CT, Rref, MPL, OHD, Cext, WSCD
Vin
−0.3 to 5.5 V
Total Power Supply and Zener Current
Output Supply Voltage with Respect to Ground
2
1
Output Current
Source
Sink
3
Output Energy (Capacitive Load per Cycle)
E.H.T.OVP, Sync Input Current
mA
V
mA
Source
9
6
Isync (Source)
IEHT (Source)
−4.0
Sink
9
6
Isync (Sink)
IEHT (Sink)
10
Idemag−ib (Source)
Idemag−ib (Sink)
−4.0
10
IE/A (Sink)
20
mA
PD
RJA
0.6
100
W
°C/W
Operating Junction Temperature
TJ
150
°C
Operating Ambient Temperature
TA
−25 to +85
°C
Demagnetization Detection Input Current
Source
Sink
8
Error Amplifier Output Sink Current
13
Power Dissipation and Thermal Characteristics
Maximum Power Dissipation at TA = 85°C
Thermal Resistance, Junction−to−Air
mA
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
ELECTRICAL CHARACTERISTICS (VCC and VC = 12 V, Rref = 10 k, CT = 2.2 nF, for typical values TA = 25°C, for
min/max values TA = −25° to +85°C unless otherwise noted.) (Note 1)
Pin #
Characteristic
Symbol
Min
Typ
Max
VOL
−
−
−
−
1.0
1.4
1.5
2.0
1.2
2.0
2.0
2.7
−
−
−
−
0.1
0.1
1.0
1.0
1.0
Unit
OUTPUT SECTION (Note 2)
Output Voltage (Note 3)
Low Level Drop Voltage
High Level Drop Voltage
3
(ISink = 100 mA)
(ISink = 500 mA)
(ISource = 200 mA)
(ISource = 500 mA)
VOH
Output Voltage During Initialization Phase
VCC − 0 to 1.0 V, ISink = 10 A
VCC − 1.0 to 5.0 V, ISink = 100 A
VCC − 5.0 to 13 V, ISink = 1.0 A
3
VOL
V
V
Output Voltage Rising Edge Slew−Rate (CL = 1.0 nF, TJ = 25°C)
dVo/dT
−
300
−
V/s
Output Voltage Falling Edge Slew−Rate (CL = 1.0 nF, TJ = 25°C)
dVo/dT
−
−300
−
V/s
1. Adjust VCC above the startup threshold before setting to 12 V. Low duty cycle pulse techniques are used during test to maintain junction
temperature as close to ambient as possible.
2. No output signal when the Error Amplifier output is in Low State, i.e., when for instance, VFB = 2.7 V.
3. VC must be greater than 5.0 V.
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MC44605
ELECTRICAL CHARACTERISTICS (VCC and VC = 12 V, Rref = 10 k, CT = 2.2 nF, for typical values TA = 25°C, for
min/max values TA = −25° to +85°C unless otherwise noted.) (Note 4)
Characteristic
Pin #
Symbol
Min
Typ
Max
Unit
Voltage Feedback Input (VE/A out = 2.5 V)
14
VFB
2.4
2.5
2.6
V
Input Bias Current (VFB = 2.5 V)
14
IFB−ib
−2.0
−0.6
−
A
AVOL
65
70
−
ERROR AMPLIFIER SECTION
Open Loop Voltage Gain (VE/A out = 2.0 V to 4.0 V)
Unity Gain Bandwidth
TJ = 25°C
TA = −25° to +85°C
BW
Voltage Feedback Input Line Regulation (VCC = 10 V to 15 V)
VFBline−reg
Output Current
Sink (VE/A out = 1.5 V, VFB = 2.7 V)
TA = −25° to +85°C
Source (VE/A out = 5.0 V, VFB = 2.3 V)
TA = −25° to +85°C
13
Output Voltage Swing
High State (IE/A out (source) = 0.5 mA, VFB = 2.3 V)
Low State (IE/A out (sink) = 0.33 mA, VFB = 2.7 V)
13
ISink
−
−
−
−
−
5.5
−10
−
10
mV
mA
2.0
12
−
−2.0
−
−0.2
VOH
VOL
5.5
−
6.5
1.0
7.5
1.1
ISource
dB
MHz
V
CURRENT SENSE SECTION
Maximum Current Sense Input Threshold
(VFeedback (pin14) = 2.3 V and VSoft−Start (pin11) = 1.2 V)
7
Vcs−th
0.96
1.0
1.04
V
Input Bias Current
7
Ics−ib
−10
−2.0
−
A
tPLH(In/Out)
−
120
200
ns
FOSC
16
−
20
kHz
Frequency Change with Voltage (VCC = 10 V to 15 V)
FOSC/V
−
0.05
−
%/V
Frequency Change with Temperature (TA = −25° to +85°C)
FOSC/T
−
0.05
−
%/°C
Ratio Charge Current/Reference Current (TA = −25° to +85°C)
Icharge/Iref
0.39
−
0.48
−
D
72
75
78
%
Vsyncth
−250
−200
−150
mV
NEG−SYNC
−0.65
−0.5
−0.34
V
Propagation Delay (Current Sense Input to Output at VTH of MOS
transistor = 3.0 V)
OSCILLATOR AND SYNCHRONIZATION SECTION
Frequency (TA = −25° to +85°C)
Free Mode Oscillator Ratio = Idischarge/(Idischarge + Icharge)
Synchronization Input Threshold Voltage
9
Negative Clamp Level (Isyncth−in = 2.0 mA)
UNDERVOLTAGE LOCKOUT SECTION
Startup Threshold
1
Vstup−th
13.6
14.5
15.4
V
Disable Voltage After Threshold Turn−On (UVLO 1)
(TA = −25° to +85°C)
1
Vdisable1
8.3
−
9.6
V
Disable Voltage After Threshold Turn−On (UVLO 2)
(TA = −25° to +85°C)
1
Vdisable2
7.0
7.5
8.0
V
4. Adjust VCC above the startup threshold before setting to 12 V. Low duty cycle pulse techniques are used during test to maintain junction
temperature as close to ambient as possible.
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MC44605
ELECTRICAL CHARACTERISTICS (VCC and VC = 12 V, Rref = 10 k, CT = 2.2 nF, for typical values TA = 25°C, for
min/max values TA = −25° to +85°C unless otherwise noted.) (Note 5)
Characteristic
Pin #
Symbol
Min
Typ
Max
Unit
Reference Output Voltage (VCC = 10 V to 15 V)
16
Vref
2.4
2.5
2.6
V
Reference Current Range (Iref = Vref/Rref, R = 5.0 k to 25 k)
16
Iref
−500
−
−100
A
Vref
−40
−
40
mV
Vdemag−th
tPLH(In/Out)
Idemag−lb
50
−
−0.5
65
0.5
−
80
−
−
mV
s
A
Minimum Off−Time when the pin 8 is grounded
TDEM−GND
1.5
3.0
4.5
s
Negative Clamp Level (Idemag = −2.0 mA)
CLVL−neg
−0.50
−0.38
−0.25
V
Positive Clamp Level (Idemag = +2.0 mA)
CLVL−pos
0.50
0.72
0.85
V
Ratio Charge Current/Iref (TA = −25° to +85°C)
Iss−ch/Iref
0.37
−
0.43
−
Discharge Current (Vsoft−start = 1.0 V)
Idischarge
1.5
5.0
−
mA
Clamp Level
VSS−CLVL
2.2
2.4
2.6
V
REFERENCE SECTION
Reference Voltage Over Iref Range
DEMAGNETIZATION DETECTION SECTION (Note 6)
8
Demagnetization Detect Input
Demagnetization Comparator Threshold (Vpin9 Decreasing)
Propagation Delay (Input to Output, Low to High)
Input Bias Current (Vdemag = 65 mV)
SOFT−START SECTION (Note 7)
Circuit Inhibition Threshold (Note 8)
VCS Soft−Start Clamp Level (Rsoft−start = 5 k)
VSSinhi
30
−
150
mV
VCSsoft−start
0.45
0.5
0.55
V
OVERVOLTAGE SECTION
Propagation Delay (VCC > 18.1 V to Vout Low)
TPHL(In/Out)
1.0
−
4.0
s
Protection Level on VCC (TA = −25° to +85°C)
VCC prot
15.9
−
18.1
V
NEG−SYN
C
−0.65
−0.5
−0.35
V
Vref
7.0
7.4
7.8
V
IEHTOVP
−5.0
−
0
A
Vshift
70
100
120
mV
ΓMPL
0.185
0.240
0.295
V−1
VMPL−th
2.4
2.5
2.6
V
ΓOHD
1.15
1.50
1.85
V−1
VOHD−th
2.4
2.5
2.6
V
EHT OVP SECTION (Note 9)
Negative Clamp Level (Isynch−in = −2.0 mA)
EHT OVP Input Threshold
EHT OVP Input Bias Current (VEHT OVP(pin 9) = 0 V)
9
WINDING SHORT CIRCUIT DETECTION SECTION
WSCD Threshold with Ipin15 = 200 A
MPL & OHD SECTION
MPL Parameter (Note 10)
MPL Comparator Threshold (Note 11)
OHD Parameter (Note 12)
OHD Comparator Threshold (Note 13)
5. Adjust VCC above the startup threshold before setting to 12 V. Low duty cycle pulse techniques are used during test to maintain junction
temperature as close to ambient as possible.
6. This function can be inhibited by connecting pin 8 to GND. In this case, there is a minimum off−time equal to TDEM−GND.
7. The MC44605 can be shut down by connecting soft−start pin (pin 11) to GND.
8. The circuit is shutdown if the soft−start pin voltage is lower than this level.
9. This function can be inhibited by connecting pin 9 to GND. In this case, the synchronization block is inhibited too and the MC44605 works
in free mode.
10. This parameter is defined in the MPL §. This parameter is obtained by measuring the MPL pin average current and dividing this result by the
corresponding squared VCS, the measured frequency value and the CT value deducted from the measured frequency value.
Measurement conditions: VFeedback(pin 14) = 2.3 V, Vsoft−start(pin 11) = 0.5 V and pins 7, 8, and 9 connected to GND (the working frequency
is typically equal to 18 kHz − Rref = 10 k "1%, CT = 2.2 nF).
11. The MPL comparator output is DisMPL.
12. This parameter is defined in the OHD §. This parameter is obtained by measuring the OHD pin average current and dividing this result by
the corresponding squared VCS value and multiplying it by the Rref value.
Measurement conditions: VFeedback(pin 14) = 2.3 V, Vsoft−start(pin 11) = 0.5 V and pins 7, 8, and 9 connected to GND (the working frequency
is typically equal to 18 kHz − Rref = 10 k "1%, CT = 2.2 nF).
13. The OHD comparator output is DisOHD.
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MC44605
ELECTRICAL CHARACTERISTICS (VCC and VC = 12 V, Rref = 10 k, CT = 2.2 nF, for typical values TA = 25°C, for
min/max values TA = −25° to +85°C unless otherwise noted.) (Note 14)
Pin #
Characteristic
Symbol
Min
Typ
Max
Unit
DISABLING BLOCK SECTION
TWSCD
−
4.0
−
s
Ratio (EHTOVP and WSCD Disabling Capacitor Charge Current)Iref
IDis−H/Iref
90
100
110
%
Ratio (MPL and OHD Disabling Capacitor Charge Current)Iref
IDis−L/Iref
2.7
3.1
3.5
%
VCCDis
1.0
−
5.0
V
Delay Pulse Width
Minimum VCC Value Enabling the Disabling Block Latch (Note 15)
TOTAL DEVICE
Power Supply Current
ICC
mA
Startup−Up (VCC = 5.0 V with VCC increasing)
−
0.35
0.55
Startup−Up (VCC = 9.0 V with VCC increasing)
−
0.35
0.55
Startup−Up (VCC = 12 V with VCC increasing)
−
0.35
0.55
Operating TA = −25°C to +85°C (Note 16)
−
20
25
Disabling Mode (VCC = 6.0 V) (Note 17)
Power Supply Zener Voltage (ICC = 35 mA)
Thermal Shutdown
−
−
0.55
VZ
18.5
−
−
V
−
−
155
−
°C
14. Adjust VCC above the startup threshold before setting to 12 V. Low duty cycle pulse techniques are used during test to maintain junction
temperature as close to ambient as possible.
15. Once a fault detection activated it, the Disabling Block Latch gets reset when the VCC becomes lower than this threshold.
16. Refer to Note 14.
17. This consumption is measured while the circuit is inhibited by the Definitive Latch.
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MC44605
Pin
Name
1
VCC
This pin is the positive supply of the IC.
2
VC
The output high state, VOH, is set by the voltage applied to this pin. With a separate
connection to the power source, it gives the possibility to set by means of an external
resistor the output source current at a different value than the sink current.
3
Output
4
GND
5
Maximum Power Limitation
6
Over−Heating Detection
7
Current Sense Input
A voltage proportional to the current flowing into the power switch is connected to this input.
The PWM latch uses this information to terminate the conduction of the output buffer. A
maximum level of 1 V allows to limit the inductor current.
8
Demagnetization Detection
A voltage delivered by an auxiliary transformer winding provides to the demagnetization pin
an indication of the magnetization state of the flyback energy reservoir. A zero voltage
detection corresponds to a complete core demagnetization. The demagnetization detection
prevents the oscillator from a re−start and so the circuit from a new conduction phase, if the
fly−back is not in a dead−time state. This function can be inhibited by connecting Pin 8 to
GND but in this case, there is a minimum off−time typically equal to 3 s.
9
Synchronization and
E.H.T.OVP Input
Activating the synchronization input pin with a pulse higher or equal to the negative
threshold (typically −200 mV) allows the next switching period to be reinitialized. The
oscillator is free when connecting Pin 9 to GND.
When the E.H.T.OVP pin receives a voltage that is greater than 7.5 V, the disabling block
Cext capacitor is charged so that the circuit gets definitively disabled if the Cext voltage
becomes higher than Vref. This block is incorporated to detect and disable the device when
the synchronization pulses are too high.
10
Oscillator Capacitor CT
The free mode oscillator frequency is programmed by the capacitor CT choice together with
the Rref resistance value. CT, connected between pin 10 and GND, generates the oscillator
sawtooth.
11
Soft−Start
A capacitor connected to this pin can temporary reduce the maximum inductor peak
current. By this way, a soft−start can be performed. By connecting pin 11 to Ground, the
MC44605 is shutdown.
12
Cext (Disabling Block)
When a too high synchronization pulse voltage (E.H.T.OVP) or a winding short circuit
(WSCD) is detected, the capacitor Cext is charged using a current source IDis− H. In the
case of a MPL or OHD fault detection, Cext is charged using IDis−L. If the Cext capacitor
voltage gets higher than Vref, the circuit is definitively disabled. Then, to re−start, the
converter must be switched off in order to make VCC decrease down to about 0 V.
13
E/A Output
14
Voltage Feedback
This is the inverting input of the Error Amplifier. It can be connected to the Switching Mode
Power Supply output through an optical (or else) feedback loop or to the subdivided VCC
voltage in case of primary sensing technic.
15
Winding Short Circuit
Detection Programmation
The W.S.C.D. block is incorporated to detect the transformer Winding Short Circuits. This
function is performed by detecting the inductor overcurrents thanks to a comparator which
threshold is programmable to be well adapted to any application.
16
Rref
Pin Description
The output current capability is suited for driving a power MOSFET.
The ground pin is a single return typically connected back to the power source. It is used as
control and power ground.
This block enables to estimate the input power. When this calculated power is detected as
too high, a fault information is sent to the disabling block in order to definitively disable the
circuit.
This block estimates the MOSFET heating. When this calculated heating is too high, the
device gets definitively disabled (disabling block action).
The error amplifier output is made available for loop compensation.
The Rref value fixes the internal reference current that is particularly used to perform the
precise oscillator waveform. The current range goes from 100 A up to 500 A.
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MC44605
Summary of the Main Design Equations
The following table consists of equations enabling to dimension a multisynchronized SMPS operating in discontinuous
mode.
Poutmax is the maximum power the load may draw in normal working.
Pin max +
ƪ
Lp max +
Ǹ2·Vac
2
Pon max + 1
3
ƫ
NVo 2
min
min
Pin max
Ǹ
2
L
ǸPin max
The maximum input power Pinmax is easily deducted by dividing Poutmax by the
efficiency (η). In this kind of application, the efficiency is generally taken equal to
80%.
η
Ǹ2·Vac
Ipk max +
d max +
Pout max
)NVo
fsync max
Vac
fsync max
min
Ipk max2
Rds on
In effect, if Lp was higher than Lpmax, a synchronized and discontinuous working
could not be guaranteed (in some cases, the demagnetization phase would not be
finished while a new conduction phase should start to follow the synchronization).
Ipkmax is the maximum inductor peak current. This current is obtained when the
power to transfer is maximum at the minimum synchronization frequency (60 W
output, 30 kHz in the proposed application).
Pin max
fsync
min
Lp
The inductor value Lp must be chosen lower than Lpmax or ideally equal to this
value (to optimize the application design−in).
d max
dmax is the maximum duty cycle. The duty cycle is maximum at the lowest input
voltage when the power demand is maximum while the synchronization frequency
also is maximum.
Ponmax is the maximum MOSFET on−time losses that are proportional to Ipkmax,
dmax and Rdson (on−time MOSFET resistor).
This conduction losses estimation enables to dimension the power MOSFET.
(V ) max + ǒǸ2
DS
Vac maxǓ ) (N
ǒ
(V ) max + Ǹ2
D
Vout)
Ǔ
Vac max
) Vout
N
(VDS)max is the maximum voltage the power switch must be able to face. In fact,
this calculation does not take into account the turnings off spikes. So, it is
necessary to take a margin of at least about 50 V.
(VD)max is the maximum voltage the high voltage secondary diode must be able
to face. Because of the turning off spikes, a margin must also be taken.
(AL) and (ni) are the magnetic parameters.
(ni) max + N
A +
L
(N
n
L
Vout
P
n
)2
Vout
Ipk max
(ni)max must not exceed the ferrite (ni). Otherwise, the transformer may get
saturated when the peak current is high.
(AL) is the ferrite constant that links the primary inductor value to the squared
number of primary turns: Lp = AL x np2.
Error Amplifier
+
1.0 mA
Compensation
A fully compensated Error Amplifier with access to the
inverting input and output is provided. It features a typical
DC voltage gain of 70 dB. The non inverting input is
internally biased at 2.5 V and is not pinned out. The
converter output voltage is typically divided down and
monitored by the inverting input. The maximum input bias
current with the inverting input at 2.5 V is −2.0 A. This can
cause an output voltage error that is equal to the product of
the input bias current and the equivalent input divider source
resistance.
RFB
Rf
Error
Amplifier
13
14 2.5 V
Cf
Voltage
Feedback
Input
MC44605
R1
R2
2R
R
1.0 V
Current
Sense
Comparator
GND
4
From Power Supply Output
Figure 1. Error Amplifier Compensation
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MC44605
The Error Amp Output (Pin 13) is provided for external
loop compensation. The output voltage is offset by two
diodes drops ([1.4 V) and divided by three before it
connects to the inverting input of the Current Sense
Comparator. This guarantees that no drive pulses appear at
the Source Output (Pin 3) when Pin 13 is at its lowest state
(VOL). This occurs when the power supply is operating and
the load is removed, or at the beginning of a soft−start
interval. The Error Amp minimum feedback resistance is
limited by the amplifier’s minimum source current (0.2 mA)
and the required output voltage (VOH) to reach the current
sense comparator’s 1.0 V clamp level:
R1(min) +
(3
I
pk
I
I
ref
+
V
R
Rref
CSTARTUP
14
1 0
R2
Q1
Q
0
Vdisable
7.5 V
STARTUP
14.5 V
CUVLO1
Substrate
Current Sense
Comparator
1
R3
PWM
Latch
Current
Sense
7
Pin 16
Vref enable
3
R
1V
R
S
ref where V + 2.5 V (typically)
ref
ref
VC
Thermal
Protection
+
VCC
(Pin 1)
Vin
S
pk(max)
As depicted in Figure 3, an undervoltage lockout has been
incorporated to guarantee that the IC is fully functional
before allowing the system working.
In effect, the VCC is connected to the non inverting input
of a comparator that has an upper threshold equal to 14.5 V
(typical Vstup−th) and a lower one equal to 7.5 V (typical
Vdisable 2). This hysteresis comparator enables or disables
the reference block that generates the voltage and current
sources required by the system.
This block particularly, produces Vref (pin 16 voltage) and
Iref that is determined by the resistor Rref connected between
pin 16 and the ground:
The MC44605 operates as a current mode controller. The
circuit uses a current sense comparator to compare the
inductor current to the threshold level established by the
Error Amplifier output (Pin 13). When the current reaches
the threshold, the current sense comparator terminates the
output switch conduction that has been initiated by the
oscillator, by resetting the PWM Latch. Thus the error signal
controls the peak inductor current on a cycle−by−cycle
basis. This configuration ensures that only one single pulse
appears at the Source Output during the appropriate
oscillator cycle.
VS
* 1.4 V
(pin13)
3
R
S
Undervoltage Lockout Section
Current Sense Comparator and PWM Latch
Vdemag out
V
The Current Sense Comparator threshold is internally
clamped to 1.0 V. Therefore the maximum peak switch
current is:
1 V) ) 1.4 V
+ 22 kΩ
0.2 mA
UVLO
Disout
[
C
Vdisable1
9.0 V
R
Reference Block:
Voltage and Current
Sources Generator
(Vref, Iref, ...)
UVLO1
(to SOFT−START)
MC44605
RS
Figure 3. VCC Management
Figure 2. Output Totem Pole
In addition to this, VCC is compared to a second threshold
level that is nearly equal to 9.0 V (Vdisable1) so that a signal
UVLO1 is generated to reset the soft−start block and so, to
disable the output stage (refer to the Soft−Start §) as soon as
VCC becomes lower than Vdisable 1. In this way, the circuit
is reset and made ready for a next startup, before the
reference block is disabled (refer to Figure 3). Thus, finally
the upper limit for the minimum normal operating voltage
The inductor current is converted to a voltage by inserting
the ground referenced sense resistor RS in series with the
power switch Q1.
This voltage is monitored by the Current Sense Input
(Pin 7) and compared to a level derived from the Error Amp
output. The peak inductor current under normal operating
conditions is controlled by the voltage at Pin 13 where:
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9
MC44605
The MC44605 oscillator achieves four functions:
— it fixes the free mode frequency
— it takes into account the synchronization signal
— it does not allow a new power switch conduction if the
flyback is not in a dead−time state when the circuit
works in demagnetization mode (pin 8 connected)
— it builds the Sf pulse required by the MPL block
During the operating mode, the oscillator sawtooth can
vary between a valley value (1.6 V typically) and a peak one
(3.6 V typically) and presents three distinct phases:
— the CT charge
— the CT discharge
— the phase during which the oscillator voltage is
maintained equal to its valley value. This happens at
the end of a discharge cycle when the synchronization
or demagnetization condition does not allow a new CT
charge phase. During this sequence, IREGUL
compensates the charge current Icharge.
The oscillator has two working modes:
— a free one when there is no synchronization
— a synchronized one.
In the free working, the oscillator grows up from its valley
value to its peak one for the charge phase and when once the
peak value is reached, a discharge sequence makes the CT
voltage decrease down to its valley value. When the
decrease phase is finished, a new charge cycle occurs if the
demagnetization condition is achieved (VDT high).
Otherwise there is a REGUL phase until VDT gets high.
In the synchronized mode, the charge cycle is only
allowed when the synchronization signal gets high while a
dead time has been detected (VDT high). This charge phase
is stopped when the synchronization signal has got low and
when the oscillator voltage is higher than Vint, the
intermediary voltage level used to generate the calibrated
pulse Sf by comparing the CT voltage to this threshold. So,
when these two conditions are performed, a discharge
sequence is set until the oscillator voltage is equal to its
valley value. Then, the CT voltage is maintained constant
thanks to the “REGUL” arrangement until the next
synchronization pulse.
In both cases, during the charge phase, a signal VS is
generated. When Sf becomes high. VS gets high and remains
in this state until the PWN latch is set of Sf is low. Then, VS
keeps low until the next Sf high level. This oscillator
behavior is obtained using the process described in
Figure 5b.
is 9.4 V (maximum value of Vdisable 1) and so the minimum
hysteresis is 4.2 V. [(Vstup−th)min = 13.6 V].
The large hysteresis and the low startup current of the
MC44605 make it ideally suited for off−line converter
applications where efficient bootstrap startup techniques are
required.
Soft−Start Control Section
The Vcs value is clamped down to the pin 11 voltage.
So, if a capacitor is connected to this pin, its voltage
increases slowly at the startup (the capacitor is charged by
an internal current source 0.4 Iref). So, Vcs is limited during
the startup and then a soft−start is performed.
This pin can be used to inhibit the circuit by applying a
voltage that is lower than VSSinhi (refer to page 4).
Particularly, the MC44605 can be shutdown by connecting
the soft−start pin to ground.
As soon as Vdis1 is detected (that is Vcc lower than
Vdisable1), a signal UVLO1 is generated until the Vcc falls
down to Vdis2 (refer to the undervoltage lockout section §).
During the delay between the disable1 and the disable2,
using a transistor controlled by UVLO1, the pin 11 voltage
is made equal to zero in order to make the soft−start
arrangement ready to work for the next re−start.
Vref
Vcs
0.4 Iref
Pin 11
Soft
Start
Capacitor
DZ
2.4 V
UVLO1
Output
Inhibition
VSSlnhi
MC44605
Figure 4. Soft−Start
Oscillator Section (Figures 5 & 5b)
The oscillator and synchronization
represented in Figure 5b.
behavior
is
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10
MC44605
a − Free mode
Inductor
current
V DT
Oscillator
Vint
Sf
Output
b − Synchronized mode
Synchro
input
Inductor
current
V DT
Oscillator
Vint
Sf
Output
Figure 5b. Oscillator Behavior
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MC44605
In effect, the output of the latch L1 is:
— high during the oscillator capacitor charge and during
the REGUL phase
— low for the oscillator capacitor discharge
Now, the latch L2 is set when the L1 output is high and the
synchronization condition is performed (that is: sync = 1 −
free mode or synchro signal high state) and during the
dead−time (VDT high). So, this latch is set for the CT charge.
On the other hand, this latch is reset by the signal used to
reset L1. Consequently, it is reset at the end of the charge
phase.
So, in any case, QL2 is:
— high during the CT charge cycle
— low in the other cases
Thus, this latch enables to obtain a signal that is high for
the charge phase and low in the other cases, whatever the
mode (synchronized or free) and whatever the
synchronization pulses width (higher than the delay
necessary for the oscillator to reach its intermediary value or
lower than this delay) in the synchronized mode.
That is why:
— the discharge current source must be connected to the
oscillator capacitor when QL1 is low. The condition (CT
voltage higher than the valley value) is added to stop
the discharge phase as soon as the oscillator voltage is
detected as lower than the valley value (without any
delay due to the L1 latch propagation time).
— the REGUL current source must be connected when:
• QL1 is high (charge or REGUL phase)
• QL2 is low (the oscillator is not in a charge phase)
On the other hand, the oscillator charge is stopped when:
— the oscillator voltage reaches the peak value in the
free mode
— the oscillator voltage is higher than the intermediary
value (Vint) and the synchronization signal is negative,
in the synchronized mode.
Consequently, in any case, QL2 that is high during the
oscillator charge phase, is high for the delay during which
the oscillator voltage grows from the valley value up to the
intermediary one. That is why the signal Sf (refer to the MPL
block) that must be high when the oscillator voltage is
between the valley value and the intermediary one during
the charge phase (QL2 high), is obtained using an AND gate
with the following inputs:
— QL2 (QL2 high <=> charge phase)
— COSCINT (COSCINT high <=> the CT voltage is lower
than the intermediary value).
So, using the output of this AND gate, Sf is obtained.
This signal Sf is connected to a logic block consisting of
two AND gates and an OR one. This block aims at supplying
a signal VS that:
— gets high as soon as Sf becomes high if the PWM
latch output is low
— gets low as soon as the PWM latch is set and then
remains low until the next cycle.
Vref
Icharge
COSCINT
Vint
&
sync
DISCH
PWM
Latch
Output
COSC HIGH
3.6 V
&
Sf
COSCINT
PWM
VS Latch
Set
&
&
COSC LOW
QL2
1.6 V
VDT (from demag
CT<1.6 V
block)
sync
SQ
&
SQ
10
CT
L1
DISCH
R
DISCH
L2
Q
R
COSC REGUL
1
0
0
&
1
QL2
Iregul
MC44605
Idischarge
Figure 5. Oscillator
Synchronization Section (Note 1)
The synchronization block consists of a protection
arrangement similar to the demagnetization block one (a
diode + a negative active clamping system (Note 2)). In
addition to this, a high value resistor (R − about 50 k) is
incorporated as the pin 9 input is also used by the EHTOVP
section.
The signal obtained at the output of this protection
arrangement, is compared to a negative threshold (−200 mV,
typically) so that when the synchronization pulse applied to
the pin 9 (through a resistor or a resistors divider to adapt this
input to the EHTOVP function), is higher than this
threshold, the system considers that the synchronization
condition is performed (free mode or synchronization signal
high level).
Note 1. The synchronization can be inhibited by connecting the
pin 9 to the ground. By this means, a free mode is
obtained.
Note 2. This negative active clamping system works even if the
circuit is off. This feature is really useful as
synchronization pulses may be applied while the product
is off.
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MC44605
VCC
Synchro.
Signal
A diode D is incorporated to clamp the positive applied
voltages while an active clamping system limit the negative
voltages to typically −0.33 V. This negative clamp level is
high enough to avoid the substrate diode switching on.
A latch system is incorporated to keep the
demagnetization block output level low as soon as a voltage
lower than 65 mV is detected and as long as a new restart is
produced (high level on the output (refer to Figure 8). This
process avoids that any ringing on the signal used on the
pin 8, disrupts the demagnetization detection (refer to
Figure 7). Finally, this method results in a very accurate
demagnetization phase detection, and the signal VDT drawn
from this block is high only for the dead time. Therefore, an
oscillator re−start and so, a new power switch conduction is
only allowed during the dead−time.
For a higher safety, the Vdemagout output of the
demagnetization block is also directly connected to the
output, to disable it during the demagnetization phase (refer
to the block diagram).
The demagnetization detection can be inhibited by
connecting pin 8 to the ground but in this case, a timer (about
3 s) that is incorporated to set the latch when it can not be
set by Vdemagout, results in a minimum off−time (refer to
Figure 8).
E.H.T. OVP
Block
Negative Active
Clamping System
Pin 9
R
sync
−200 mV
MC44605
Figure 6. Synchronization
Demagnetization Section
This block is incorporated to detect the complete core
demagnetization in order to prevent the power MOSFET
from switching on if the converter is not in a dead time
phase. That is why this block inhibits any oscillator re−start
as long as the inductor current is not finished (from the
beginning of the on−time to the end of the demagnetization
phase).
In a fly−back, a good means to detect the demagnetization
phase consists in using the VCC winding voltage. In effect,
this voltage is:
— negative during the on−time,
— positive during the off−time,
— equal to zero for the dead−time with generally a
ringing (refer to Figure 7).
Zero
Current
Detection
Output
Buffer
3 s
R
Q
Demag
Q
S
Negative Active
Clamping System
Vdemag out
0.75 V
VCC
Vpin 8
Pin 8
65 mV
C DEM
65 mV
Oscillator
D
VDT
−0.33 V
Figure 8. Demagnetization Block
On−Time
Off−Time Dead−Time
Overvoltage Protection Section
Figure 7. Demagnetization Detection
The overvoltage arrangement compares a portion Vcc to
Vref (2.5 V) (refer to Figure 9). In fact, this threshold
corresponds to a VCC equal to to 17 V. When the Vcc is
higher than this level, the output is latched off until a new
circuit re−start.
That is why, the MC44605 demagnetization detection
consists of a comparator that compares the VCC winding
voltage to a reference that is typically equal to 65 mV.
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MC44605
Vref
For instance, if this threshold value is required to be equal
to 30 V, Vpin9 must be equal to 7.5 V when the
synchronization pulse value is 30 V.
So, in this case:
VCC
In
Delay τ 5.0 s
T
0
30
Out
2.5 V
Then, the ratio (r1/r2) can be deducted:
Enable
In
COVLO
VOVP out
τ Out
Delay
r1 + 3
r2
So, as r1 and r2 must be negligible in relation to R (about
50 k), the couple of resistors can be chosen as follows:
2.0 s
2.5 V
(Vref)
r2 + 7.5
r1 ) r2
(If VOVP out = 1.0,
the Output is Disabled)
r1 + 3 kΩ
and:
Figure 9. Overvoltage Protection
r2 + 1 kΩ
A delay (2 s) is incorporated in order to avoid any
activation due to interferences by only taking into account
the overvoltages that last at least 2 s.
The VCC is connected when once the circuit has
started−up in order to limit the circuit startup consumption
(T is switched on when once Vref has been generated).
The overvoltage section is enabled 5 s after the regulator
has started to allow the reference Vref to stabilize.
Winding Short Circuit Detection Section (WSCD)
The MC44605 being designed to control a Fly−Back
SMPS, this block is incorporated to detect a short circuit on
a transformer winding or on an output diode (refer to
Figure 11).
+
E.H.T. Overvoltage Protection Section
AC Line
+
Lp
This block uses the synchronization input as this section
is incorporated to detect too high synchronization pulses and
then to activate the device definitive latch in this case.
r1
Lleak
MC44605
VCC
Synchro.
Pulses
Negative Active
Clamping System
RS
Synchronization
Block
Disabling
Block
Pin 9
2R
r2
4V
+
CEHTOVP
Figure 11. Winding Short Circuit Fault
In the case of a Winding Short Circuit, the primary
inductor Lp is short circuited and then the current increase
is only controlled by the leakage inductor Lleak.
In current mode, the power switch conduction is stopped
when the inductor current is detected as high enough, by the
controller. In fact, when the current sense resistor (Rs)
voltage gets equal to Vcs, the current sense comparator
switches to reset the output.
Now, the circuit has a propagation delay and the power
switch needs some time to turn off. Consequently, there is a
delay t between the moment at which the Rs voltage gets
equal to Vcs and the actual current increase stop. So, this
results in an overcurrent (refer to Figure 12).
E.H.T.
OVP
R
Vref
MC44605
Figure 10. E.H.T. OVP
This block consists of a high impedance resistors bridge
(R is nearly equal to 50 k − refer to Figure 10) so that the
EHTovp threshold is 7.5 V. So, using an external resistors
bridge (r1, r2 <<R), the synchronization pulse level above
which the working must be considered as wrong, can be
adjusted.
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MC44605
Power Switch Current
(VCS + Vshift)/RS
Finally, when there is a winding short circuit, an
overcurrent is detected by the WSCD comparator. The
output of this comparator, VWSCD, is connected to the
disabling block (refer to the disabling block §).
(Vin x t/Lleak
Vin x t/Lp
VCS/RS
Maximum Power Limitation Section (MPL)
The MPL block is designed to calculate this input power
using the following equation:
Pin + 1
2
t
Figure 12. Overcurrent in a WSCD Case
δt
P
where: Vin is the input voltage (rectified a.c. line)
While in a WSCD case:
(Ipk)
WSCD
+ Vin δt
L
Leak
V
Consequently, as the leakage inductor value is generally
much lower than the primary one (less than 5% generally),
the overcurrent is much higher in the WSCD case. That is
why this fault can be detected by detecting the high
overcurrents.
So, the WSCD block consists of comparing the sensed
current to a reference equal to: (Vcs + Vshift), where Vshift is
a voltage proportional to the current injected in the pin 15
(refer to Figure 13).
Isense
CWSCD
VWSCD
(Sf)
T
Vcs 2
MPL
R
C
ref
T
where: k1 is a constant
On the other hand, kMPL that is depending on the reference
current source Iref, is proportional to 1/Rref:
k
MPL
+ k2
1
R
ref
where: k2 is a constant
Pin 15
Vshift shift = 500 k
MPL
(Sf) + k1
R
Disabling
Block
+R
Now, as Sf is built comparing the oscillator to a constant
level, (Sf) is proportional to Rref and CT:
3.75 Vshift
MPL
where: kMPL is the multiplier gain
(Sf) is the width of the calibrated pulse
T is the switching (oscillator) period
Vin
Pin 7
f
As Vcs is proportional to the inductor peak current
(Vcs = Rs x Ipk), the squared Ipk value is estimated by
building a current source proportional to Vcs2. This current
is chopped by a calibrated pulse Sf, generated at each new
oscillator cycle (refer to Figure 14).
Finally, using an external resistor and capacitor network
(RMPL, CMPL) on the MPL pin, a voltage VMPL,
proportional to the input power can be obtained.
In effect,
Now, in normal working, this overcurrent Ipk is equal to:
Ipk + Vin
L
Ipk 2
P
where: Lp is the inductor value
Ipk is the inductor peak current
f is the switching frequency
time
t
L
So:
V
Ishift
MPL
+R
MPL
k1
Vcs 2
k2
f
C
T
where: CT is the oscillator capacitor
Vcs
Finally:
MC44605
V
Figure 13. WSCD
MPL
+R
MPL
Γ
MPL
Vcs 2
f
C
T
where: ΓMPL is the MPL parameter as defined in the
specification. This is a constant equal to the product
(k1 x k2).
Now, as the overcurrent level depends on the input voltage
Vin, it is preferable to use a Vshift proportional to this input
voltage instead of a constant Vshift. So, the WSCD pin must
be connected to Vin through a resistor that fixes Vshift by
adjusting the current injected in this pin 15.
Now, as:
Pin + 1
2
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15
L
P
Ipk 2
f
MC44605
As in the MPL section, the squared Ipk term is estimated
by building a current source proportional to Vcs2.
The duty cycle is taken into account thanks to the action
on this current source of a “chopper” controlled by the
circuit output. By this means, the pin 6 average current is
proportional to the squared peak current multiplied to the
duty cycle (refer to Figure 14).
So, using an external resistor and capacitor network
(ROHD, COHD) on this pin, a voltage VOHD, proportional to
the conduction losses can be obtained.
Like in the MPL block, this voltage VOHD, is compared to
2.5 V. If VOHD gets higher than this threshold, the disabling
block is activated by DisOHD (output of the comparator).
The external resistor ROHD choice enables to obtain a
calculated VOHD equal to 2.5 V when the conduction losses
are equal to their maximum value.
In effect,
and:
Vcs + R
Ipk
S
So:
V
MPL
+
2
R
C
Γ
MPL
MPL
L
P
R 2
S
T
Pin
A comparator is used to compare VMPL to Vref, the output
of which, DisMPL, is connected to the “definitive inhibition
latch” of the disabling block. So, when the calculated power
is higher than the threshold, the circuit is definitively
disabled (the system considers that there is an overload
condition).
Finally, replacing VMPL by 2.5 V (the threshold value),
the RMPL value to be used, can be deducted:
R
MPL
+
1.25
Γ
C
MPL
T
L
P
R 2 (Pin) max
S
V
Vcs
OHD
k
kMPLVcs2
TOHD
TMPL
Output
R MPL
C MPL
C OHD
R OHD
DisOHD
OHD
OHD
+R
OHD
+
Vcs 2
R
ref
k2
OHD
d
Finally:
V
Disabling
Block
R
Γ
OHD
OHD
R
ref
Vcs 2
d
where: ΓOHD is the OHD parameter as defined in the
specification. This is a constant equal to k2.
VMPL
DisMPL
Now, as:
Vcs + R
MC44605
V
Ipk
OHD
+
3
R
OHD
R
ref
Γ
R
OHD
dson
R 2
S
p on
So, by choosing the value of ROHD, the heating
corresponding to Vref is determined. If the MOSFET
dissipation is such that the heating is higher than this
threshold, the “definitive inhibition latch” of the Disabling
Block is activated and so, the output gets definitively
disabled.
Overheating Detection Section (O.H.D.)
In the MPL block, the converter input power is calculated.
In the O.H.D. block, that is the power MOSFET heating
which is calculated, using the following equation:
Ipk 2
S
So, replacing Vcs and using the pon equation:
Figure 14. OHD and MPL
dson
1
R
ref
+ k2
So:
Sf
VMPL
R
d
where: k2 is a constant
V
p on + 1
3
Vcs 2
OHD
Now, as kOHD that is depending on the reference current
source Iref, is proportional to 1/Rref:
kOHDVcs2
2.5 V
k
OHD
where: kOHD is the multiplier gain
x
2.5 V
+R
d
where: pon are the power switch on−time losses
Rdson is the conduction MOSFET resistor
d is the duty cycle
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16
MC44605
Consequently, by replacing VOHD by 2.5 V (threshold
value) in the last equation, the value ROHD to use, can be
deducted:
OHD
+
3
Γ
R
ref
R 2
OHD
S
R
Vref
104% Iref
3.4% Iref
dson
(p on) max
1
where: (pon)max are the maximum on time losses that are
0
C ext
acceptable.
Disabling Block Section
This section consists of a “definitive inhibition latch”
(directly supplied by the Vcc) that disables the output (the
output is forced to zero).
In effect, this block aims at definitively disabling the
circuit when one of the following faults is detected:
— a Winding Short Circuit
— too high synchronization pulses
— a too high input power
— a too high power switch (MOSFET) heating
The signals corresponding to these faults are high when a
fault is detected (for instance, when the input power is
detected as too high, DisMPL is high).
When one (or several) of these four faults is detected, a
current source charges Cext (with a certain duty cycle) and
when its voltage becomes higher than Vref, the definitive
inhibition latch is activated. Thus, the circuit gets
definitively disabled after a delay depending on Cext.
According to the detected fault, the current that charges
Cext is not the same:
The typical values are:
— 260 A for EHTOVP and WSCD
— 8.5 A for OHD and MPL
DisOHD
1
0
E.H.T.OVP
Q S
DisMPL
Pin 12
VWSCD
R
VCC
R ext
R
2.5
Vref
Delay
4S
Definitive
Inhibition
Latch
2.5 V
Output
Buffer
MC44605
Figure 15. Disabling Block
This latch is reset when the Vcc falls down to about 3.0 V.
In this case, if a new startup is performed, the circuit will
work normally (until this fault or another one is detected).
Practically, to re−start after a fault has shutdown the
circuit, the converter must be turned off for a time long
enough to enable the Vcc capacitor discharge (repair time...).
Note: As VWSCD is generally a really narrow pulse, it is
necessary to add a latch and a delay to build a 4 s width
pulse when VWSCD becomes high.
when Rref is equal to 10 k.
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MC44605
Application Schematic
90 Vac to
264 Vac
1nF / 1KV
RFI
Filter
R1
1 / 5W
4.7 M
C4....C7
1nF/500V
Vin
D1 ... D4
1N4007
160 V/0.1 A
100 F
400 V
MR856
47 K
1.8 M
100 F
25 V
3.3 k
2.2 nF
1 F
10 nF
470
k
9
8
10
7
11
6
12
1N4148
1 nF
14
22 k
22 k
13
MC44605P
1.2 k
3
15
2
16
1
10 k
2x150 K//
70 V/0.2 A
1N4937
1N4937
100 F
1F
120 pF 27 K
100 k
Laux
40 V/0.5 A
1nF
1N4937
105
k
4.7 F
5
4
47 nF
1N4934
47 k/2W
SYNC
100 F
4.7 F
Lp
340 K
470 F
470 pF
MTA4N60E
39 1N4937
1 k
100 220 nF
1305 V/0.65 A
1N4934
1000 F
10 k
470 330 1 k
100 k
8 V/0.5 A
0.22 1N4934
1000 F
2.2 k
270 226 k
MOC8103
10 k
Vin
2.2 k
1N4733
100 nF
6.8 nF
33 nF
TL431
3.6 k
65 W output SMPS controlled by the MC44605
Mains input range: 90 Vac <−> 264 Vac
Synchronization range: 30 kHz <−> 100 kHz
Orega Transformer ref. G5984−00
(Lp = 195 H)
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18
MC44605
Performances
Input Voltage
90−260 Vac
Synchronization Range
30 to 100 kHz
160 V
100 mA
70 V
200 mA
40 V
500 mA
13.5 V
650 mA
8.0 V
500 mA
Outputs
30 kHz
Measured Efficiency
(Pout = 64 W)
60 kHz
100 kHz
Standby Losses
(No Load − Pout = 0)
83%
110 Vac
81%
220 Vac
82%
110 Vac
80%
220 Vac
80%
110 Vac
2.0 W
3.2 W
28 V
30 kHz
60 kHz
100 kHz
Overheating Detection
(Pout = 64 W):
The input rms levels at which
the circuit detects an OHD case.
Winding Short Circuit
Detection
80%
220 Vac
220 Vac
EHTovp Threshold
Maximum Power
Limitation
110 Vac (Input)
110 Vac (Input)
86 W (Input)
220 Vac
87 W
110 Vac
90 W
220 Vac
95 W
110 Vac
94 W
220 Vac
110 W
30 kHz
85 V
60 kHz
76 V
100 kHz
76 V
Fully Functional
(Tested by short circuiting one output diode or one transformer winding)
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19
MC44605
PACKAGE DIMENSIONS
PDIP−16
CASE 648−08
ISSUE T
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION L TO CENTER OF LEADS
WHEN FORMED PARALLEL.
4. DIMENSION B DOES NOT INCLUDE
MOLD FLASH.
5. ROUNDED CORNERS OPTIONAL.
−A−
16
9
1
8
B
F
C
L
S
−T−
H
SEATING
PLANE
K
G
D
M
J
16 PL
0.25 (0.010)
M
T A
M
DIM
A
B
C
D
F
G
H
J
K
L
M
S
INCHES
MIN
MAX
0.740 0.770
0.250 0.270
0.145 0.175
0.015 0.021
0.040
0.70
0.100 BSC
0.050 BSC
0.008 0.015
0.110 0.130
0.295 0.305
0_
10 _
0.020 0.040
MILLIMETERS
MIN
MAX
18.80 19.55
6.35
6.85
3.69
4.44
0.39
0.53
1.02
1.77
2.54 BSC
1.27 BSC
0.21
0.38
2.80
3.30
7.50
7.74
0_
10 _
0.51
1.01
GreenLine is a trademark of Motorola, Inc.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
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associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
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20
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For additional information, please contact your local
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MC44605/D