A519HRT HART Modem

AND8346/D
A519HRT HART) Modem
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APPLICATION NOTE
Physical Layer Specification2. The A5191HRT is intended
to replace the 20C15 for all existing and future HART
applications with no circuit topology changes. Only the
values of four external resistors in the receive filter need to
be adapted.
Introduction
This application note describes a demonstration circuit
that permits a user to implement a HART slave or master
interface between a microprocessor and a process loop using
the ON Semiconductor A5191HRT HART modem
integrated circuit. The information in this Application Note
is correct to the best of our knowledge. Development of a
circuit suitable to the user’s particular system and
application environment is the responsibility of the user.
The HART (Highway Addressable Remote Transmitter)
communication protocol provides digital communication
for microprocessor−based process control instruments.
HART uses the Bell−202 forward channel signaling
frequencies and bit rate (1200 bits/second) as making it a
subset of the Bell−202 standard. HART−speaking devices
can use virtually any Bell−202 standard modem. However,
the ON Semiconductor A5191HRT single−chip modem has
been designed to meet the low power requirements of
2−wire process instruments.
The ON Semiconductor A5191HRT modem is designed
to allow the user to easily implement a HART compliant
Physical Layer design conforming to the HART FSK
Features
• Same modem design as 20C15 from LSI Logic
(formerly NCR and Symbios)
Transmits a trapezoidal signal
Internal oscillator cell
Internal receive filter
Carrier detect
Available in 28−pin PLCC, 32−pin QFN and
32−pin LQFP Packages
• These Devices are Pb−Free and are RoHS Compliant
This application note shows how to interface the
A5191HRT modem to the HART network, as well as other
general advice on using the modem and on designing HART
devices. A block diagram showing a typical application of
the A5191HRT in a HART Slave is shown in Figure 1.
•
•
•
•
•
PC20101210.4
RxD
RxA
+
TxD
UART
A5191HRT
RTS
TxA
CD
+
4 − 20 mA
Loop
Current
mP
4 − 20 mA
DAC OUT
DAC
−
Figure 1. HART Slave Application Block Diagram
© Semiconductor Components Industries, LLC, 2013
August, 2013 − Rev. 4
1
Publication Order Number:
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OVERVIEW OF HART COMMUNICATIONS
Analog Signaling
The HART Master or Controller (Primary Master) detects
this current variation by measuring the DC voltage across
the current sense resistor. The loop current varies from 4 to
20 mA at frequencies usually under 10 Hz.
HART devices are connected in a conventional current
loop arrangement shown in Figure 2. The HART Slave or
Process Transmitter or Field Instrument (terminology used
by HART specification) signals by varying the amount of
current flowing through itself.
Process Variable
HART Master
Network
24 V
Loop Current
ADC
4 − 20 mA
Current
Sense
Resistor
Receive
Amplifier
Receive
Amplifier
Tx Current
Modem
Tx Enable
Modem
HART Slave
PC20101207.1
Tx Voltage
Figure 2. Current Loop with HART Signal Sources. Analog Signaling is Marked in Black. Digital Signaling is
Marked in Blue.
Digital Signaling
HART Waveform
The HART (digital) signal is superimposed on the
4−20 mA (analog) signal as shown in Figure 2 (marked in
blue). The Master transmits HART signals by applying a
voltage signal across the current sense resistor and it
receives a voltage signal by detecting the HART current
signal across the sense resistor. Conversely, the Slave
transmits by modulating the loop current with HART signals
and receives HART signals by demodulating the loop
current.
HART
signals
using
phase−continuous
frequency−shift−keying
(FSK)
at
1200
bit/s.
Phase−continuous frequency−shift−keying requires the
phase angle of the mark (1200 Hz) and the space (2200 Hz)
to remain continuous at the 1200 bit/s bit boundaries. A
HART Slave or Field Instrument transmits a HART signal
by modulating a high−frequency carrier current of about
1 mA p−p onto its normal output current. This is illustrated
in Figure 3 for a 6 mA analog signal.
Loop Current
Phase continuous at bit boundary
Mark = 1200 Hz
Space = 2200 Hz
1 mA
6 mA
t
t BIT
Data
833 ms
PC20101207.2
“0”
“1”
“0”
“1”
Figure 3. Field Instrument Current versus Time
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“0”
“0”
AND8346/D
HART Slave Device
HART network can only have one Primary and one
Secondary Master connected a time.
HART Slaves transmit by modulating the process
4−20 mA DC loop current with a 1 mA p−p AC current
signal as shown in Figure 3. Since the average value of the
HART signal is zero, the DC value of the process loop
remains unchanged. Receive circuits in a HART Slave
device amplify, filter, and demodulate the current signal.
Multiplexing a HART Master
To reduce the design complexity of a multiple loop HART
Master, the physical layer can be multiplexed to two or more
process loops. This is usually done with analog switches that
allow signals as high as 6 Vp−p to pass and exhibit an
extremely low ‘on resistance’. The added impedance of the
switch directly affects the output impedance of the Master
device.
The multiplexer can switch only the HART signal or it can
switch both the HART signal and the associated signal
return. Switching the signal return insures the physical layer
interface will be non−intrusive to the HART network if a
failure were to occur. Typically, the analog switches connect
to each process loop through a coupling capacitor (about
2.2 mF).
The greatest disadvantage of multiplexing HART signals
is the reduction in communication throughput to each Slave
device.
HART Multi−dropped Slave Devices
Some current loops (called networks in HART
documents) use only digital signaling. The field instrument
current is fixed at 4 mA or some other convenient value, and
only digital communication occurs. Up to 15 such field
instruments with unique addresses of 1 through 15 may be
connected in parallel. More address space is available for
devices compliant with HART specification rev. 5 or higher
(up to 38 bits). A device that is not multi−dropped will
usually have its address set to 0.
HART Master Device
HART Masters transmit by driving the loop with a low
impedance voltage source as shown in Figure 2. Regardless
whether a master or field device is transmitting, a signal
voltage of about 500 mVp−p is developed across the
conductors of the current loop (assuming a 500 W current
sense resistor), and is seen by both devices. Receive circuits
in each device filter and demodulate the signal voltage.
HART Cabling
Because of the relatively low HART frequencies, there is
little cable attenuation and delay distortion. This results in
very few restrictions on constructing networks. The
complete topology requirements and electrical requirements
for HART devices are given in the HART Physical Layer
Specification2.
In most applications, HART communications can be
performed up to a distance of 1500 meter using existing field
wiring for a 2−wire process instrument.
HART Primary Master
In general, a HART Primary Master is the device that
provides the communications between the control system
(DCS) and the remote process instruments with the intent to
receive process information and perform maintenance
operations. A HART network that has a HART Master
interface integrated into the DCS will usually be configured
as a Primary Master.
HART Data Link Layer
Normally, one HART device talks while others listen.
Talking means that the device applies the modulated carrier
to the network cable. A given device applies the carrier in
one unbroken segment called a frame. Between frames the
network is silent. Field instrument frames are usually
responses to commands by a Master. Further information on
network protocol is found in the HART Data Link Layer
Specification1.
HART Secondary Master
In general, a second HART interface connected to a
network that contains a Master will be a Secondary Master.
An example of a Secondary Master is a hand−held
communicator that would be connected directly across a
HART. Such a network may have a Primary Master. A
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A5191HRT FUNCTIONAL BLOCKS
VDD
VDDA
RxAFI
RxAF
PC20101211.1
RESET
RxD
RxA
Demodulator
Logic
Rx Comp
FSK_IN
Rx HP Filter
AREF
CD
Carrier Detect
Counter
CDREF
DEMODULATOR
Carrier Comp
RTS
TxD
MODULATOR
Numeric
Controlled
Oscillator
Sine
Shaper
Clock
Oscillator
BIAS
XOUT XIN
TxA
FSK_OUT
A5191HRT
CBIAS
VSS
VSSA
Figure 4. A5191HRT Block Diagram
HART MODEM
Demodulator
The bit rate is nominally 1200 bits/second. The output of
the modulator RxD is qualified with the carrier detect signal
CD. Therefore only RxA signals large enough to be detected
(100 mVp−p typical) by the carrier detect function will
produce demodulated output at RxD.
The demodulator accepts an FSK signal at its RxA input
and reproduces the original modulating signal at its RxD
output shown below:
The modem uses shift frequencies of nominally 1200 Hz
(logical one, mark) and 2200 Hz (logical zero, space).
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FSK_IN
RxD
LSB
IDLE (mark)
PC20101207.4
MSB
IDLE (mark)
Start
D0
D1
D2
D3
D4
D5
D6
D7
Par
“0”
“1”
“0”
“1”
“0”
“0”
“1”
“0”
“1”
“0”
tBIT
Stop
t
8 data bits
BIT
Figure 5. Demodulator Signal Timing
Modulator
TxA output. RTSB must be a logic low for the modulator
to be active.
The modulator accepts digital data in NRZ form at its TxD
input and generates the FSK modulated signal at its
RTS
IDLE (mark)
t
TxD
tBIT
Start
“0”
Par
“1”
“0”
“1”
“0”
“0”
“1”
“0”
“1”
Stop
t
“0”
TxA
0.5 V
t
PC20101208.3
Figure 6. Modulator Signal Timing
Transmit Waveshaping
For
VAREF = 1.235 V
VTxA = 1.235 x 0.417 = 0.515 V p−p
The DC bias voltage of TxA is VTxA = 0.5 V.
This means that when:
RTSB = “1” VTxA = 0.5 V
RTSB = “0” VTxA has a voltage swing from 0.16 V
to 0.77 V.
The A5191HRT generates a HART compliant trapezoidal
FSK modulated signal at its TxA output. Shown in Figure 6
are actual transmit signals from a A5191HRT.
The amplitude of TxA is proportional to the analog
reference voltage as follows:
V TxA,p*p + V AREF
0.417
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VTxA
VTxA
“1” = Mark; fm =1.2 kHz
0.5 V
0.5 V
SRm = 1860 V/s
0
1
“0” = Space; fs =2.2 kHz
0.5 V
0.5 V
0
2
t (ms)
SRs = 3300 V/s
t (ms)
1
2
PC20101209.4
Figure 7. TxA Waveforms
Carrier Detect
FSK_IN
Low HART input signal levels increases the risk for the
generation of bit errors. If the received signal is below this
100 mV p−p level the demodulator is disabled.
NOTE: HART Physical Layer Specification specifies a
signal level > 120 mVp−p as a valid carrier; signal level
< 80 mVp−p as noise.
This level detection is done in the Carrier Detector. The
output of the demodulator is qualified with the carrier detect
signal (CD), therefore, only RxA signals large enough to be
detected (100 mVp−p typically) by the carrier detect circuit
produce received serial data at RxD.
1
2
3
4
50 mV
0
CD
tCD
2.2 ms
PC20101211.2
Figure 9. Carrier Detect Timing
Once CD goes inactive, it takes four consecutive pulses
out of the comparator to assert CD again. Four consecutive
pulses equals tCD = 3.33 ms when the received signal is
1200 Hz and tCD = 1.82 ms when the received signal is 2200
Hz.
Clock Oscillator
The A5191HRT requires a 460.8 kHz clock. This
frequency is generated in the internal clock oscillator block,
designed to use either a quartz crystal, ceramic resonator, or
an external clock.
Figure 8. Demodulator Carrier and Signal
Comparator
The carrier detect comparator shown in Figure 8
generates logic Low output if the RxAFI voltage is below
CDREF. The comparator output is fed into a carrier detect
block. The carrier detect block drives the carrier detect
output pin CD high if nRTS is high and four consecutive
pulses out of the comparator have arrived. CD stays high as
long as nRTS is high and the next comparator pulse is
received in less than 2.2 ms.
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input to the oscillator cell. Table 5 lists commonly available
clock frequencies for crystals suited to this purpose. The
interface between the microprocessor clock and the
A5191HRT could be as simple as a direct connection or a
single integrated circuit.
Clock
Oscillator
XIN
XOUT
R1
PC20101208.5
Xtal
Clock
Oscillator
C1
C2
XOUT
Figure 10. Oscillator Connection for Ceramic
Resonator or Quartz Crystal
460.8 kHz
When using a crystal or resonator the accuracy should be
at least 1%. The oscillator requires two external capacitors
and one external resistor, which values varies depending on
the used type. The specifications are listed in Table 4. Care
should be taken to keep the circuit board traces between the
A5191HRT and the external oscillator components as short
as possible.
Figure 11. External Clock
NOTE: Output XOUT is driven by an external source
Table 2. COMMONLY AVAILABLE FREQUENCY
MULTIPLES
Ceramic Resonator Sources
Ceramic resonators are less expensive than quartz
crystals, but are not as accurate. Unfortunately, ceramic
resonators at the needed frequency require special ordering
in very large quantities. Ceramic resonators that oscillate at
460.8 kHz are available from:
http://www.ecsxtal.com/store/dc-4-ecs-resonators.aspx
www.raltron.com/products/resonators/default.asp
Quartz crystals are available from:
www.aelcrystals.co.uk
www.statek.com/prod_thruholecrystals.php
Ceramic
Quartz
Type
ECS Crystal
ZTB Series
Statek CX−1V
Series
Frequency
460.8
460.8
kHz
C1
220
47
pF
C2
220
22
pF
R1
0
10
kW
Maximum ESR
N.A.
12
kW
Frequency
Multiple
1.8432 MHz
x4
3.6864 MHz
x8
7.3728 MHz
x16
14.7456 MHz
x32
29.4912 MHz
x64
Clock Skew
If one uses the same time base for both the modem and the
UART, a 1% accurate time base may not be good enough.
The problem is a combination of receive data jitter and clock
skew between transmitting and receiving HART devices. If
the transmit time base is at 99% of nominal and the receive
time base in another device is at 101% of nominal, the
receive data (at the receiving UART) will be skewed by
roughly 21% of one bit time at the end of each 11−bit byte.
This is shown in Figure 12. The skew time is measured from
the initial falling edge of the start bit to the center of the 11th
bit cell. This 21% skew by itself isn’t bad. However, there
is another error source for bit boundary jitter. The Phase
Lock Loop demodulator in the A5191HRT produces jitter in
the receive data that can be as large as 12% of a bit time.
Therefore, a bit boundary can be shifted by as much as 24%
of a bit time relative to its ideal location based on the start−bit
transition. (The start−bit transition and a later transition can
be shifted in opposite directions for a total of 24%.)
The clock skew and jitter added together is 45%, which is
the amount that a bit boundary could be shifted from its
expected position. UARTs that sample at mid−bit will not be
affected. However, there are UARTs that take multiple
samples during each bit to try to improve on error
Table 1. CERAMIC RESONATOR AND QUARTZ
CRYSTAL VALUES
Description
PC20101118.6
XIN
Unit
External Clock
It may be desirable to use an external clock of 460.8 kHz
rather than the internal oscillator cell because of the cost and
availability of quartz crystals or ceramic resonators. In
addition, the A5191HRT consumes less current when an
external clock is used as shown in Figure 11. An external
clock associated with the microprocessor (running at a
frequency that is a multiple of 460.8 kHz) can be used as an
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AND8346/D
performance. These UARTs may not be satisfactory,
depending on how close the samples are to each other, and
how samples are interpreted. A UART that takes a majority
vote of 3 samples is acceptable.
Transmitter
tBIT
tBIT
2
tBIT@ 99% nominal CLK
Start
Receiver
Even if your own time base is perfect, you still must plan
on a possible 35% shift in a bit boundary, since you don’t
have control over time bases in other HART devices.
D0
D1
D2
D3
D4
D5
D6
D7
Par
Stop
t
45% t BIT = 1200-1
tBIT@ 101 % nominal CLK
21% tBIT = 1200-1
t
PC20101209.1
UART mid bit sample moment
±12% jitter
Figure 12. Clock Skew
RECEIVE ANALOG CIRCUITRY
Receive High−pass Filter
The interfering analog signal should be reduced to at least
ten times smaller than the smallest HART signal, or under
about 7.5 mV. Therefore, the high−pass filter should have an
attenuation of 63 dB at 25 Hz. The HART signal band covers
approximately 950 Hz to 2500 Hz, which means that the
high−pass filter should begin rolling off somewhere below
950 Hz and be 63 dB down at 25 Hz. This is illustrated in
Figure 13.
To remove the interfering analog signal, a high−pass filter
is required in the HART signal receive path. The filter
requirements are found as follows. From section 7.1 of the
HART Physical Layer Specification2, the interfering signal
can be as high as 16 V p−p at 25 Hz.
NOTE: This is directly related to limits on analog
signaling. The difference in specifications for analog
interference as output versus analog interference as input is
the result of the loads being different.
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Figure 13. Receive Filter Bounds
The A5191HRT has an internal active filter to attenuate
the frequencies outside the HART band. In addition to the
internal active filter, an external passive filter is necessary to
complete the filtering requirements. The external capacitors
and resistor were too large in size to cost effectively integrate
into the A5191HRT silicon.
The external components required for the receive filter is
shown in Figure 14. All the external capacitors are ±5% and
the resistors are ±1% components (except the 3 MW which
is ±5%).
The external components on the receiver create a third
order high pass filter with a pole at 624 Hz and a first order
low pass filter with a pole at 2500 Hz. Internally, the
A5191HRT has a high pass pole at 35 Hz and a low pass pole
at 90 kHz, each of which can vary by as much as ±30 %. The
input impedance to the entire filter is greater than 150 kW at
frequencies less than 50 kHz.
A5191HRT is a pin to pin replacement for 20C15. Only
the values of four resistors in the receiver need to be
changed. (See Table 5 values marked in italic).The different
resistor values change the shape of the lower pass band for
the receive filter. It creates a more robust and noise immune
receiver and as a results provides more margin in passing the
out−of−band noise interference tests.
Table 3. RECEIVE FILTER VALUES FOR A5191HRT
AND 20C15
Symbol
A5191HRT
20C15
Tolerance
Unit
R1
215
402
1%
kW
R2
215
453
1%
kW
R3
499
825
1%
kW
R4
787
787
1%
kW
R5
422
732
1%
kW
R6
215
215
1%
kW
R7
3
3
5%
MW
C1
470
470
5%
pF
C2
1
1
5%
nF
C3
1
1
5%
nF
C4
220
220
5%
pF
The values shown for all external components in
Figures 14 and Table 6 and all other circuits in this
application note are those used in the circuitry which was
used to pass the HART physical layer conformance test for
the A5191HRT.
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C4
R5
R6
RxAF
RxAFI
PC20101124.1
HART IN
300 pF
R3
RxA
15 MW
Rx Comp
R4
214 kW
C3
C2
R2
C1
R1
40 pF
R7
44 kW
Rx HP Filter
Carrier Comp
AREF
CDREF
1.235 VDC
AREF − 80 mV
DEMODULATOR
Figure 14. Receive Filter
Voltage References
The bias current controls the operating parameters of the
internal operational amplifiers and comparators and should
be set to 2.5 mA.
The value of the bias current resistor is determined by the
reference voltage AREF and the following formula:
The A5191HRT requires two voltage references, AREF
and CDREF. AREF sets the DC operating point of the
internal operational amplifiers and is the reference for the
Rx comparator.
If A5191HRT operates at VDD = 3.3 V the
ON Semiconductor LM285D−1.2 1.235 V reference is
recommended. In the case VDD = 5 V, AREF is typically
2.5 VDC. LM285D−2.5 is recommended.
The level at which CD (Carrier Detect) becomes active is
determined by the DC voltage difference (CDREF − AREF).
Selecting a voltage difference of 80 mV will set the carrier
detect to a nominal 100 mVp−p.
R BIAS +
The recommended bias current resistor is 499 kW when
AREF is equal to 1.235 V or 100 kW when AREF is 2.5 V.
Supply Current Budget and Transmitter Lift−Off
Voltage
Current consumption of internal circuits is important in
any 2−wire field instrument. It becomes critically important
in a microprocessor− based field instrument featuring both
analog and digital signaling. The available current is derived
here and some techniques are examined for reducing current
consumption.
The nominal HART signal transmitted by a field
instrument is 0.5 mA peak. When this is superimposed upon
a 4 mA analog signal, the terminal current must vary
between 3.5 mA and 4.5 mA. During the peak of the HART
waveform, the instrument has 4.5 mA available, but during
the valley it has only 3.5 mA. Energy storage techniques can
be used to allow the internal circuits to draw a steady 4 mA
at all times. However, to be effective at HART frequencies,
the storage capacitor should be quite large. A large capacitor
(or any form of energy storage device) complicates the
circuit design if intrinsic safety is required.
Therefore, the circuit is normally designed to run
everything on 3.5 mA. Another 200 to 400 mA is often
subtracted from this to allow some margin and to satisfy
other conditions. Assuming a guard value of 200 mA, the
Analog Bias Resistor
The A5191HRT requires a bias current resistor RBIAS to
be connected between CBIAS and VSS.
2.5 mA
BIAS
AREF
OPA
PC20101118.4
AREF
2.5 mA
CBIAS
RBIAS
Figure 15. Bias Circuit
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internal circuits of the 2−wire field instrument have to live
on a diet of 3.3 mA during transmit. While receiving 3.8 mA
is available.
temperature. One way of reducing the A5191HRT current
consumption is to operate it at reduced voltage. Since the
A5191HRT is a CMOS part, current consumption is roughly
proportional to supply voltage. Operation at VDD = 3.3 V is
common.
In applying the A5191HRT be careful not to let inputs
float. The nRTS pin of the A5191HRT will often be driven
by an I/O pin of a microprocessor. During power−up or after
reset an I/O pin may be tri−stated, allowing it to float. This
can cause the A5191HRT to draw excess current. In some
cases the current may be large enough to prevent the field
instrument from starting up properly. There should be a
1 MW pull−up resistor on RTSB.
The transmitter lift−off voltage is the minimum terminal
voltage at which it is guaranteed to operate. When only
analog signaling is involved, lift−off voltage is an
unambiguous quantity. But when HART digital signaling is
added, the available voltage can swing by as much as 0.75 V
above and below the DC level. The minimum applied
voltage is the DC level minus 0.75 volt. You should either
design the field instrument to accommodate the dips in
voltage, or else specify the lift−off voltage to include them.
Loop Current (mA)
1 mA
HART signal
4
3.5
3.3
200 mA guardband
3.3 mA
Supply Current Budget
t
PC20101212.1
Figure 16. Supply Current Budget
A characteristic of the A5191HRT is that its current
consumption is approximately 350 to 450 mA. This leads to
the fortunate circumstance that the remaining (non−modem)
circuits always have at least 2.85 mA available. Margins
may be needed to cover current consumption over
INTERFACING TO THE HART NETWORK
Slave Device
capacitively coupled into the summing junction, to preserve
DC accuracy of the 4 − 20 mA analog signal. Also shown in
Figure 17 is the receive path, consisting of a capacitor from
the collector of the current regulator transistor to a
high−impedance amplifier. Both the receive amplifier and
the transmit current source should present a high impedance
(greater than 100 kW) to the network.
A5191HRT
The Slave interface to the network is typically a current
regulator, as illustrated in simplified form in Figure 17. Its
output current is controlled by varying a much smaller
current into an op−amp summing junction. This junction is
a convenient point at which to sum the analog and digital
signals, thereby achieving superposition of the digital signal
onto the analog. The digital transmit signal (TxA) is
Receive
Amplifier
+
RxA
TxA
Loop
Amplifier
Loop
Current
Vref
D/A
Converter
−
PC20101125.2
Figure 17. Simplified HART Slave Interface Circuit
Transmit Interface
amplifier in the current loop regulator or at the regulator
itself. It is recommended that the amplitude can be adjusted
with voltage gain circuits. Using a small series capacitance
to attenuate the signal may cause distortion of the
transmitted signal.
The amplitude of TxA is nominally 500 mVp−p. The
HART Physical Layer requires the Slave to modulate the
loop with a 1 mA p−p signal. The amplitude of modulation
of the loop current will have to be adjusted before the
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Receive Interface
network is driven by a low output impedance (600 W or less)
voltage amplifier circuit that can be switched to a high
impedance state using RTSB. The output of the voltage
amplifier needs to be high impedance while the HART
Master receives, to insure a high input impedance for the
receiver. Typically a HART Master uses a 2.2 mF coupling
capacitor to insure the transmitter circuit meets the output
impedance requirements specified in the HART Physical
Layer Specification.
All HART receivers require non−disruptive coupling to
the network current loop. This connection can be made with
capacitive or inductive coupling without corruption of the
process loop current. Typically a HART Slave uses a small
value (1 nF) capacitor to couple an adequate signal level to
the receive filter.
Master Device
The HART Master interfaces to the network as a voltage
source, as illustrated in simplified form in Figure 18. The
Receive
Amplifier
+
A5191HRT
RxA
RTS
Loop
Current
TxA
PC20101125.3
Loop
Amplifier
Vref
Figure 18. Simplified HART Master Interface Circuit
Transmit Interface
could have as much as 1 kW of loop resistance. This results
in (20 – 4)10−3 x 1.103 = 16 V p−p low frequency carrier with
HART superimposed. The transmit switch must be designed
to prevent this 16 V p−p signal from sinking current through
the HART transmitter’s output stage. Any clipping of the
16 Vp−p signal through the HART transmitter or the use of
transient suppressors will result in the clipping of the
superimposed HART signal. The last solution will block and
not clip the 16 Vp−p process signal and will not corrupt the
input impedance. The third solution can be implemented
easily as shown in Figure 18.
The amplitude of TxA is nominally 500 mVp−p. The
500 mVp−p meets the requirements for a HART Master.
However, the A5191HRT is unable to source enough current
to drive a HART network, therefore a low impedance
voltage driver build around the Loop Amplifier is required
between the A5191HRT and the network.
Transmit Switch
During a HART message transmission from the Master,
the output impedance of the voltage source is quite low (0 −
600 W). However during reception of a HART message, the
input impedance of the receive section must be quite high.
Therefore, the transmitter section must be disabled when not
active (nRTS = “High”).
Several methods can be used to achieve a high output
impedance of the transmitter while nRTS is inactive. One
method is choosing an op−amp with an “active low” current
programming resistor that will basically “turn−off” the
op−amp during receive operations. A second method can be
implemented with a correctly biased discrete FET in series
with the output stage. Lastly, a discrete solid−state relay with
a characteristically low “on resistance” can be used.
All three above techniques will provide an adequate
solution for the impedance requirements. However, only the
third solution will allow proper circuit operation per the
“bit−error−rate” requirements and does not need to be
powered by a dual supply.
Under worse case conditions, a transmitter could
dynamically change from 4 − 20 mA and a HART Master
Receiving the HART Signal
HART Signal Coupling
All HART receivers require non−disruptive coupling to
the network current loop. This connection can be made with
capacitive or galvanic coupling without corruption of the
process loop current. A receiver in a Master may use a single
2.2 mF capacitor to couple the HART signal in which the
Master is always connected across a current sense resistor
and share the same ground. Where ground isolation is
required, another coupling capacitor may be necessary in the
signal return connection.
To insure complete isolation from the network, galvanic
isolation is the preferred method. Typically a transformer
and a series capacitor are used to couple the HART signal.
With this type of isolation, the master can be connected
across the field instrument or the current sense resistor
regardless of the polarity or grounding configuration.
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AND8346/D
Current Sense Resistor
PC20101125.5
The current sense resistor is considered an integral part of
a HART Master. In operation, a HART message response
from a Slave is superimposed on the 4−20 mA current loop.
The Slave’s 1 mA p−p HART signal is dropped across the
Master’s sense resistor. For a typical sense resistor of 250 W,
one can expect 250 mVp−p signal extracted from a HART
Slave device. A HART Master can have a sense resistor
ranging from 230 W to 600 W.
4.7 mF
HART
Modem
+
Current
Sense
Resistor
Loop
Current
−
4.7 mF
Figure 20. Dual Capacitive Coupling
Master Connection to a HART Network
A HART Master must be able to receive voltage signals
that are developed across the sense impedance. In addition,
it must be able to apply a transmit signal across the current
sense impedance. In some cases the Master will signal
across the field device as well. In all cases, the presence of
the HART Master must not disrupt the analog signaling of
the process loop. Therefore, the HART signals must be
non−intrusively coupled from the current sense resistor to
the Master through one of the various coupling techniques
listed below.
Capacitive coupling can work well when Masters have
their power isolated from ground. When the system power
configuration is unknown, the use of transformer coupling
will insure the elimination of ground effects. An example of
transformer coupling is shown below in Figure 21.
PC20101125.6
2.2 mF
HART
Modem
Coupling Techniques
+
Current
Sense
Resistor
Coupling to the network is most commonly done with
capacitors and transformers. The simplest form of coupling
is a single capacitor as shown in Figure 19.
This coupling technique will only work if one side of the
Current Sense Resistor is connected to the analog ground.
Loop
Current
−
Figure 21. Transformer Coupling
Grounding Effects
With the use of capacitive coupling, AC and DC ground
loops can be created if one is not aware of the grounding
techniques of both the Master and the process loop. If the
Master is powered from a battery or a galvanically isolated
power source, then single capacitive coupling will work in
any HART network.
If a Master (that does not have an isolated power source)
with single capacitive coupling is connected to a sense
resistor or field device that is not at ground potential, a DC
ground loop will occur as shown in below:
In this case the Field Device will have a direct short across
it.
PC20101125.4
2.2 mF
HART
Modem
+
Current
Sense
Resistor
Loop
Current
This coupling technique will only work if one side
of the Current Sense Resistor is connected to the
analog ground.
Figure 19. Single Capacitive Coupling
Using two capacitors will allow a connection across a
sense resistor or field device that does not have one of its
connections at analog ground as show below in Figure 20.
When two coupling capacitors are used, lager values are
required to meet the impedance requirements.
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AND8346/D
illustrated in Figure 24. This means, for example, that a
sinusoidal output at 25 Hz must have an amplitude into a
500 W test load of less than 8 Vp−p. It also means that a
25 Hz square wave of 8 Vp−p would not be acceptable, since
its harmonics do not decrease at a −40 dB rate.
The required roll−off of the 4 – 20 mA analog signal can
be achieved by various means, including:
• Analog filters.
• Digital filters (software).
• Inherent filtering in the instrument sensor.
Often it is a combination of these. Any convenient method
may be used to insure that changes in instrument output
current fall within the specification. If the field instrument
uses a D/A converter (DAC) to generate its analog output,
the output steps of the DAC must be sufficiently small or else
the high−frequency content of the steps must be removed by
filtering. When large changes in the DAC output occur due
to calibration operations, caution must be used not to have
these DAC changes active during a HART command or
response. Any large and fast current changes that fall outside
the HART specifications can cause unreliable
communications with the HART Master or Slave.
The analog signaling of typical field instruments is
usually band−limited to the range of 1 to 10 Hz, so that the
roll−off starts well below the 25 Hz point of Figure 24. This
can lead to simpler low− pass filtering. For example, a
single−pole filter that begins rolling off at 1 Hz falls below
the curve of Figure 24.
The dashed line illustrated the use of a simple first order
filter with a pole at 1 Hz.
PC20101209.1
2.2 mF
Loop
Current
HART
Modem
+
Current
Sense
Resistor
HART
Field
Device
In this case the Field Device will have a direct
short across it.
Figure 22. Single Capacitor Coupled DC Ground
Loop
If the Master has a ground connection that is not at the
ground potential as the analog ground from the process loop,
it is possible to create AC ground loops. A noisy 240 VAC
ground signal could be coupled directly unto the process
loop ground (single capacitive coupling) or coupled unto the
process loop (dual capacitive coupling) ground through the
connection. The AC noise potentially could impede HART
communications and corrupt the accuracy of the DCS
measuring the analog current as shown in Figure 23.
PC20101209.2
Loop
Current
HART
Modem
+
Current
Sense
Resistor
Digital
Ground
Analog
Ground
The dashed line shows the ground loop when a
single coupling capacitor is used..
Figure 23. Capacitor Coupled AC Ground Loop
Limiting the Analog Signal Frequency Bandwidth
Digital signaling can potentially interfere with analog
signaling. A much worse problem is the analog signal
interfering with digital signaling. This is due to the relative
size of the two signals. For example, a change from 4 to
20 mA can produce a voltage change of as high as 16 V
across the field instrument terminals, depending on the
amount of resistance in the loop. At the same time, the
instrument may be trying to detect a HART (digital) signal
as low as 80 mVp−p.
Separating the two is usually done with a combination of
low−pass filtering of the analog signal and high−pass
filtering of the HART signal. The HART Physical Layer
Specification1 (section 7.2) limits the analog signal to
16 mA p−p at frequencies below 25 Hz and constrains the
output above 25 Hz to fall within a −40 dB/decade slope, as
Figure 24. Analog Signal Bandwidth Limitation
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AND8346/D
Miscellaneous
UARTs differ in when they indicate empty. Some indicate
empty at the time of the last shift clock − that is,
simultaneously with the stop bit being shifted out. But if
carrier were to be turned off at that time, the modem would
cease transmitting and the stop bit wouldn’t be sent.
Therefore, it is important to determine which kind of UART
you have. If it behaves as described, then you should wait at
least one bit time after the empty indication until you turn off
carrier. Note that adding a 1 bit delay can’t do any harm, even
if it isn’t needed.
Start of Frame
Because the HART carrier must be started and stopped,
the receiving UART and processor must become
synchronized to each HART frame. During the
synchronization period at the beginning of the frame, it is
normal for some transmitted bytes to be corrupted or lost.
Various HART requirements have been devised to insure
that synchronization will occur. They are:
• The transmitting device must send at least five
preamble bytes (hex FF).
• The transmitting device must load the first preamble
byte into the transmit UART within 5−bit times of
starting carrier.
• The receiving device must recognize carrier within
30−bit times.
Correct start−up is based on the fact that, during the stream
of preamble bytes, the start bits (applied to the UART) are
the only 0 bits.
Everything else (stop, data, and parity bits), including idle
time, is a 1. Although some extra 0 bits may be generated by
the start−up transient, after a short time only valid start bits
will remain and the device will be synchronized.
Misuse of Carrier Detect
To maximize speed, a field instrument will often be
designed to begin its response frame immediately after the
Master’s command frame. These frames are close enough
together that there is not enough time for carrier detect to
drop out. This brings out an important point regarding
carrier detect: It sole purpose is to indicate the presence of
carrier, so that receivers know that they have sufficient
signal to work with. Carrier detect is not intended to reliably
indicate the start or end of a particular frame. Start of frame
detection is a Data Link Layer function and must occur
through examination of the frame content.
More information
More information can be found under following
weblinks:
Datasheets and Application notes of devices used above
http://www.onsemi.com
HART Standards:
http://www.hartcomm.org/
End of Frame
The UART can cause a possible problem at the end of
frame. Normally, it is necessary to wait until the last bit of
the last character of the frame has been sent until carrier is
turned off. The UART usually tells you it is empty, which is
an indication that you should turn carrier off. However,
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AND8346/D
APPENDIX
HART Slave
only. For detailed information on the Hart Physical Layer
specification and requirements see the HART Physical
Layer Specification1.
The schematic in Figure 25 is a possible implementation
of the HART Slave physical layer. The Loop Current
Regulator Circuit build around the op amp is for information
3.3 V
C4
C6
21, 30
RESET
CAT808
VDDA
VDD
R14
IC1
13
R5
R6
RxAFI
24 27 29 31 32
RxAF
16
15
14
2
C3
R3
RxA
R4
C2
C1
R2
R1
R7
VDDA
RxD
25
A5191HRT
CD
26
3.3 V
mC
R9
8
Z1
R11
R13
R10
C5
+
9
RTS
TxD
PC20101210.1
LM285
AREF
CDREF
22
C9
TxA
23
17
R15
Opamp
Q1
7
18
1
3
4
5
11 28 6, 20 12,19
VSS VSSA CBIAS
XIN
XOUT
10
*
X1
R12
C7
R19
C10
HART &
4 - 20 mA OUT
R17
C8
4 ć 20 mA
DAC OUT
R20
R16
R18
−
**
Figure 25. Possible Implementation for HART Slave Physical Layer
Table 4. COMPONENT LIST HART SLAVE IMPLEMENTATION
Symbol
Value
Tolerance
Units
R1
215
1%
kW
R2
215
1%
kW
R3
499
1%
kW
R4
787
1%
kW
R5
422
1%
kW
R6
215
1%
kW
R7
3
5%
MW
R8
470
5%
pF
R9
10
1%
kW
R10
200
1%
kW
R11
14.7
1%
kW
R12
499
1%
kW
R13
1
5%
MW
R14
100
5%
kW
*Value depends on used operational amplifier
* *Value depends on VMSB of DAC
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AND8346/D
Table 4. COMPONENT LIST HART SLAVE IMPLEMENTATION
Symbol
Value
Tolerance
Units
R15
47
1%
kW
R16
11.3
1%
kW
R17
120
1%
W
R18
120
1%
kW
R19
3.6
5%
kW
R20
6.2**
1%
kW
C1
470
5%
pF
C2
1
5%
nF
C3
1
5%
nF
C4
220
5%
pF
C5
10
20%
nF
C6
100
20%
nF
C7
220
5%
pF
C8
220
5%
pF
C9
100
5%
nF
C10
Z1
*
LM285
http://www.onsemi.com/pub/Collateral/LM285−D.PDF
Q1
BC547
http://www.onsemi.com/pub/Collateral/BC546−D.PDF
IC1
CAT808
http://www.onsemi.com/pub/Collateral/CAT808−D.PDF
X1
ZTBF−460.8−E
http://www.ecsxtal.com/store/pdf/ztb_ztbfr.pdf
*Value depends on used operational amplifier
* *Value depends on VMSB of DAC
HART Master
coupling circuit build around the op amp is for information
only. For detailed information on the Hart Physical Layer
specification and requirements see HCF_SPEC−54.
The schematic in Figure 26 is a possible implementation
of the HART Master physical layer. The HART signal
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AND8346/D
3.3 V
C4
C6
21, 30
RESET
CAT808
VDDA
VDD
R14
IC1
13
R5
R6
RxAFI
24 27 29 31 32
16
RxAF
R3
RxA
15
C3
C2
C1
14
2
R4
R2
R1
R7
VDDA
RxD
25
A5191HRT
CD
26
3.3 V
mC
R9
8
R11
R13
Z1
C5
R10
9
RTS
TxD
PC20101210.1
LM285
AREF
CDREF
22
Opamp
TxA
23
17
18
1
3
4
5
11 28 6, 20 12,19
C9
10
VSS VSSA CBIAS
XIN
XOUT
Loop
Current
IC2
X1
R15
R12
C7
MC74VHC1GT66
7
RTS
C8
Figure 26. Possible Implementation for HART Master Physical Layer
Table 5. COMPONENT LIST HART MASTER IMPLEMENTATION
Symbol
Value
Tolerance
Unit
R1
215
1%
kW
R2
215
1%
kW
R3
499
1%
kW
R4
787
1%
kW
R5
422
1%
kW
R6
215
1%
kW
R7
3
5%
MW
R8
470
5%
pF
R9
10
1%
kW
R10
200
1%
kW
R11
14.7
1%
kW
R12
499
1%
kW
R13
1
5%
MW
R14
100
5%
kW
R15
499*
1%
W
C1
470
5%
pF
C2
1
5%
nF
C3
1
5%
nF
C4
220
5%
pF
C5
10
20 %
nF
*Value depends on the required HART master Current Sense sensitivity
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AND8346/D
Table 5. COMPONENT LIST HART MASTER IMPLEMENTATION
Symbol
Value
Tolerance
Unit
C6
100
20 %
nF
C7
220
5%
pF
C8
220
5%
pF
C9
2,2
20 %
mF
Z1
LM285
http://www.onsemi.com/pub/Collateral/LM285−D.PDF
Q1
BC547
http://www.onsemi.com/pub/Collateral/BC546−D.PDF
IC1
CAT808
http://www.onsemi.com/pub/Collateral/CAT808−D.PDF
IC2
1GT66
http://www.onsemi.com/pub/Collateral/MC74VHC1GT66−D.PDF
X1
ZTBF−460.8−E
http://www.ecsxtal.com/store/pdf/ztb_ztbfr.pdf
*Value depends on the required HART master Current Sense sensitivity
A5191HRT Current Consumption
different operating conditions. Other possibilities to reduce
current consumption include shutting down the master
during dead time, and careful design of reference voltage
generation.
The current consumption of the 15191HRT can be
drastically reduced by reducing operating voltage and using
an external clock. Table 6 lists typical current usage under
Table 6. CURRENT CONSUMPTION IN DIFFERENT CIRCUIT CIRCUMSTANCES
Operating Condition
Operating Voltage
Current Consumption
Idle, External oscillator
2.8 V
220 mA (Typ.)
Idle, External oscillator
3V
240 mA (Typ.)
Normal Operation
3V
350 mA (Typ.)
Absolute maximum current consumption
3V
450 mA (Max.)
REFERENCES
Layer Specification, Revision 8.1; 9390 Research
Blvd., Suite I−350, Austin Texas, 78759.
3. HART Communication Foundation Document
Number HCF_TEST_2, HART FSK Physical
Layer Test Specification, Revision 2.1; 9390
Research Blvd., Suite I−350, Austin Texas, 78759.
1. HART Communication Foundation Document
Number HCF_SPEC−13, HART FSK
Communication Protocol Specification, Revision
7.3; 9390 Research Blvd., Suite I−350, Austin
Texas, 78759.
2. HART Communication Foundation Document
Number HCF_SPEC−54, HART FSK Physical
HART is a registered trademark of the HART Communication Foundation of Austin, Texas, USA. Any time that the term ‘HART’ is used in
this document or in any document referenced by this document, that term implies the registered trademark.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC owns the rights to a number of patents, trademarks,
copyrights, trade secrets, and other intellectual property. A listing of SCILLC’s product/patent coverage may be accessed at www.onsemi.com/site/pdf/Patent−Marking.pdf. SCILLC
reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any
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