3-in-1 PWM Dual Buck and Linear Power Controller

NCP5220
3−in−1 PWM Dual Buck and
Linear Power Controller
The NCP5220 3−in−1 PWM a Dual Buck and Linear Power
Controller, is a complete power solution for MCH and DDR memory.
This IC combines the efficiency of PWM controllers for the VDDQ
supply and the MCH core supply voltage with the simplicity of linear
regulator for the VTT termination voltage.
This IC contains two synchronous PWM buck controller for driving
four external N−Ch FETs to form the DDR memory supply voltage
(VDDQ) and the MCH regulator. The DDR memory termination
regulator (VTT) is designed to track at the half of reference voltage
with sourcing and sinking current.
Protective features include, soft−start circuitry, undervoltage
monitoring of 5VDUAL, BOOT voltage and thermal shutdown. The
device is housed in a thermal enhanced space−saving DFN−20
package.
Features
• Pb−Free Package is Available*
• Incorporates Synchronous PWM Buck Controllers for VDDQ and
•
•
•
•
•
•
•
•
•
•
•
•
•
VMCH
Integrated Power FETs with VTT Regulator Source/Sink up to 2.0 A
All External Power MOSFETs are N−Channel
Adjustable VDDQ and VMCH by External Dividers
VTT Tracks at Half the Reference Voltage
Fixed Switching Frequency of 250 kHz for VDDQ and VMCH
Doubled Switching Frequency of 500 kHz for VDDQ Controller in
Standby Mode to Optimize Inductor Current Ripple and Efficiency
Soft−Start Protection for All Controllers
Undervoltage Monitor of Supply Voltages
Overcurrent Protections for DDQ and VTT Regulators
Fully Complies with ACPI Power Sequencing Specifications
Short Circuit Protection Prevents Damage to Power Supply Due to
Reverse DIMM Insertion
Thermal Shutdown
5x6 DFN−20 Package
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NCP5220
AWLYYWW
1
DFN−20
MN SUFFIX
CASE 505AB
• DDR I and DDR II Memory and MCH Power Supply
1
NCP5220 = Specific Device Code
A
= Assembly Location
WL
= Wafer Lot
YY
= Year
WW
= Work Week
PIN CONNECTIONS
COMP
FBDDQ
SW_DDQ
BG_DDQ
TG_DDQ
BOOT
SS
PGND
VTT
VDDQ
5VDUAL
COMP_1P5
SLP_S3
TG_1P5
BG_1P5
GND_1P5
AGND
FBVTT
SLP_S5
FB1P5
NOTE: Pin 21 is the thermal pad
on the bottom of the device.
ORDERING INFORMATION
Device
Typical Applications
MARKING
DIAGRAM
20
NCP5220MNR2
NCP5220MNR2G
Package
Shipping†
DFN−20
2500 Tape & Reel
DFN−20
(Pb−Free)
2500 Tape & Reel
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specification
Brochure, BRD8011/D.
*For additional information on our Pb−Free strategy and soldering details, please
download the ON Semiconductor Soldering and Mounting Techniques
Reference Manual, SOLDERRM/D.
© Semiconductor Components Industries, LLC, 2004
November, 2004 − Rev. 4
1
Publication Order Number:
NCP5220/D
NCP5220
12 V
SLP_S3
SLP_S3
SLP_S5
BOOT
SLP_S5
13 V
Zener
SS
5VDUAL
5VDUAL
CSS
VTT
VTT
1.25 V,
2.0 Apk
M1
COUT2
TG_DDQ
VDDQ
FBVTT
2.5 V, 20 A
L
AGND
SW_DDQ
COUT1
M2
NCP5220
BG_DDQ
CZM2
R5
CZM1
PGND
COMP_1P5
RZM2
RZM1
CPM1
FB1P5
COMP
5VDUAL
CZ1
R6
M3
VMCH
CZ2
CP1
TG_1P5
RZ1
L
R1
RZ2
FBDDQ
1.5 V, 10 A
COUT3
M4
R2
BG_1P5
VDDQ
PGND
Figure 1. Application Diagram
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2
NCP5220
VREF
VCC
VOLTAGE
and CURRENT
REFERENCE
_VREFGD
SLP_S5
THERMAL
SHUTDOWN
TSD
12 V
BOOT
13 V
ZENER
SLP_S3
VCC
_BOOTGD
R10
5VDUAL
S0
CONTROL
LOGIC
BOOT_
5VDUAL
S3
UVLO
VREF
R11
VOCP
+
ILIM
5VDUAL
+
−
R12
5VDUAL_
VREF
R13
VCC
TG_DDQ
VDDQ
UVLO
_5VDLGD
L
V1P5
VDDQ
SW_DDQ
PGND
VCC
BG_DDQ
SS
M2
COUT1
PGND
PGND
OSC
M1
and
PWM
LOGIC
CSS
5VDUAL
S0
S3
COMP
VREF
CZ1
AMP
CZ2
CP1
RZ1
A1
FBDDQ
R1
RZ2
R2
5VDUAL
VCC
M3
TP_1P5
VMCH
L2
PGND
1805 Phase
Shift
VCC
BG_1P5
M4
COUT3
GND_1P5
PGND
COMP_1P5
AMP_MCH
VREF
CZM1
CZM2
CPM1
RZM1
A1
FB1P5
5VDUAL
S0
RM2
VDDQ
R16
M2
R17
VTT
Regulation
Control
AGND
VTT
VTT
R18
5VDUAL
COUT2
M3
R19
AGND
AGND
PGND
FBVTT
Figure 2. Internal Block Diagram
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3
RM1
RZM2
NCP5220
PIN DESCRIPTION
Pin
Symbol
1
COMP
VDDQ error amplifier compensation node.
Description
2
FBDDQ
DDQ regulator feedback pin.
3
SS
4
PGND
5
VTT
6
VDDQ
Power input for VTT linear regulator.
7
AGND
Analog ground connection and remote ground sense.
8
FBVTT
VTT regulator pin for closed loop regulation.
9
SLP_S5
Active LOW control signal to activate S5 Power OFF State.
10
FB1P5
11
GND_1P5
12
BG_1P5
Gate driver output for V1P5 regulator low side N−Channel Power FET.
13
TG_1P5
Gate driver output for V1P5 regulator high side N−Channel Power FET.
14
SLP_S3
Active LOW control signal to activate S3 sleep state.
15
COMP_1P5
16
5VDUAL
17
BOOT
18
TG_DDQ
Gate driver output for DDQ regulator high side N−Channel Power FET.
19
BG_DDQ
Gate driver output for DDQ regulator low side N−Channel Power FET.
20
SW_DDQ
DDQ regulator switch node and current limit sense input.
21
TH_PAD
Copper pad on bottom of IC used for heatsinking. This pin should be connected to the ground plane under
the IC.
Soft−start pin of DDQ and MCH.
Power ground.
VTT regulator output.
V1P5 switching regulator feedback pin.
Power ground for V1P5 regulator.
V1P5 error amplifier compensation node.
5.0 V dual supply input, which is monitored by undervoltage lock out circuitry.
Gate driver input supply, which is monitored by undervoltage lock out circuitry, and a boost capacitor
connection between SWDDQ and this pin.
MAXIMUM RATINGS
Rating
Symbol
Value
Unit
5VDUAL
−0.3, 6.0
V
BOOT
−0.3, 14
V
Gate Drive (Pins 12, 13, 18, 19) to AGND (Pin 7)
Vg
−0.3 DC,
−4.0 for t100 ns; 14
V
Input / Output Pins to AGND (Pin 7)
Pins 1−3, 5, 6, 8−10, 14−15, 20
VIO
−0.3, 6.0
V
Power Supply Voltage (Pin 16) to AGND (Pin 7)
BOOT (Pin 17) to AGND (Pin 7)
PGND (Pin 4), GND_1P5 (Pin 11) to AGND (Pin 7)
VGND
−0.3, 0.3
V
Thermal Characteristics
DFN−20 Plastic Package
Thermal Resistance Junction−to−Air
RqJA
35
°C/W
Operating Junction Temperature Range
TJ
0 to + 150
°C
Operating Ambient Temperature Range
TA
0 to + 70
°C
Storage Temperature Range
Tstg
− 55 to +150
°C
Moisture Sensitivity Level
MSL
2.0
Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit
values (not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied,
damage may occur and reliability may be affected.
1. This device series contains ESD protection and exceeds the following tests: Human Body Model (HBM) " 2.0 kV per JEDEC standard:
JESD22–A114. Machine Model (MM) " 200 V per JEDEC standard: JESD22–A115.
2. Latchup Current Maximum Rating: " 150 mA per JEDEC standard: JESD78.
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NCP5220
ELECTRICAL CHARACTERISTICS (5VDUAL = 5.0 V, BOOT = 12 V, TA = 0°C to 70°C, L = 1.7 mH, COUT1 = 3770 mF,
COUT2 = 470 mF, COUT3 = NA, CSS = 33 nF, R1 = 2.166 kW, R2 = 2.0 kW, RZ1 = 20 kW, RZ2 = 8.0 W, CP1 = 10 nF,
CZ1 = 6.8 nF, CZ2 = 100 nF, RM1 = 2.166 kW, RM2 = 2.0 kW, RZM1 = 20 kW, RZM2 = 8.0 W, CPM1 = 10 nF, CZM1 = 6.8 nF,
CZM2 = 100 nf for min/max values unless otherwise noted). Duplicate component values of MCH regulator from DDQ.
Characteristic
Symbol
Test Conditions
Min
Typ
Max
Unit
4.5
5.0
5.5
V
12.0
13.2
V
SUPPLY VOLTAGE
5VDUAL Operating Voltage
BOOT Operating Voltage
V5VDUAL
VBOOT
SUPPLY CURRENT
S0 Mode Supply Current from 5VDUAL
I5VDL_S0
SLP_S5 = HIGH, SLP_S3 = HIGH,
BOOT = 12 V, TG_1P5 and BG_1P5
Open
10
mA
S3 Mode Supply Current from 5VDUAL
I5VDL_S3
SLP_S5 = HIGH, SLP_S3 = LOW,
TG_1P5 and BG_1P5 Open
5.0
mA
S5 Mode Supply Current from 5VDUAL
I5VDL_S5
SLP_S5 = LOW, BOOT = 0 V,
TG_1P5 and BG_1P5 Open
1.0
mA
S0 Mode Supply Current from BOOT
IBOOT_S0
SLP_S5 = HIGH, SLP_S3 = HIGH,
BOOT = 12 V, TG_1P5 and BG_1P5
Open
25
mA
S3 Mode Supply Current from BOOT
IBOOT_S3
SLP_S5 = HIGH, SLP_S3 = LOW,
TG_1P5 and BG_1P5 Open
25
mA
4.4
V
550
mV
10.4
V
UNDERVOLTAGE−MONITOR
5VDUAL UVLO Upper Threshold
5VDUAL UVLO Hysteresis
V5VDLUV+
V5VDLhys
BOOT UVLO Upper Threshold
VBOOTUV+
BOOT UVLO Hysteresis
VBOOThys
250
400
1.0
V
THERMAL SHUTDOWN
Tsd
(Note 3)
145
°C
Tsdhys
(Note 3)
25
°C
VFBQ
TA = 25°C
TA = 0°C to 70°C
Feedback Input Current
IDDQFB
V(FBDDQ) = 1.3 V
Oscillator Frequency in S0 Mode
FDDQS0
217
Oscillator Frequency in S3 Mode
FDDQS3
434
Thermal Shutdown
Thermal Shutdown Hysteresis
DDQ SWITCHING REGULATOR
FBDDQ Feedback Voltage, Control Loop in
Regulation
Oscillator Ramp Amplitude
dVOSC
(Note 3)
Current Limit Blanking Time in S0 Mode
1.178
1.166
1.190
1.202
1.214
V
1.0
mA
250
283
KHz
500
566
KHz
1.3
Vp−p
TDDQbk
(Note 3)
400
nS
Current Limit Threshold Offset from 5VDUAL
VOCP
(Note 3)
0.8
V
Minimum Duty Cycle
Dmin
0
%
Maximum Duty Cycle
Dmax
Soft−Start Pin Current for DDQ
100
Iss1
V(SS) = 0.5 V
4.0
%
mA
DDQ ERROR AMPLIFIER
DC Gain
GAINDDQ
(Note 3)
70
dB
Gain−Bandwidth Product
GBWDDQ
COMP PIN to GND = 220 nF,
1.0 W in Series (Note 3)
12
MHz
SRDDQ
COMP PIN TO GND = 10 pF
8.0
V/uS
Slew Rate
3. Guaranteed by design, not tested in production.
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NCP5220
ELECTRICAL CHARACTERISTICS (5VDUAL = 5.0 V, BOOT = 12 V, TA = 0°C to 70°C, L = 1.7 mH, COUT1 = 3770 mF,
COUT2 = 470 mF, COUT3 = NA, CSS = 33 nF, R1 = 2.166 kW, R2 = 2.0 kW, RZ1 = 20 kW, RZ2 = 8.0 W, CP1 = 10 nF,
CZ1 = 6.8 nF, CZ2 = 100 nF, RM1 = 2.166 kW, RM2 = 2.0 kW, RZM1 = 20 kW, RZM2 = 8.0 W, CPM1 = 10 nF, CZM1 = 6.8 nF,
CZM2 = 100 nf for min/max values unless otherwise noted). Duplicate component values of MCH regulator from DDQ.
Characteristic
Symbol
Test Conditions
Min
DVTTS0
IOUT= 0 to 2.0 A (sink current)
IOUT= 0 to –2.0 A (source current)
−30
Typ
Max
Unit
30
mV
VTT ACTIVE TERMINATION REGULATOR
VTT tracking DDQ_REF/2 at S0 mode
VTT Source Current Limit
ILIMVTsrc
2.0
A
VTT Sink Current Limit
ILIMVTsnk
2.0
A
SLP_S5, SLP_S3 Input Logic HIGH
Logic_H
2.0
SLP_S5, SLP_S3 Input Logic LOW
Logic_L
0.8
V
Ilogic
1.0
mA
CONTROL SECTION
SLP_S5, SLP_S3 Input Current
V
GATE DRIVERS
TGDDQ Gate Pull−HIGH Resistance
RH_TG
VCC = 12 V, V(TGDDQ) = 11.9 V
3.0
W
TGDDQ Gate Pull−LOW Resistance
RL_TG
VCC = 12 V, V(TGDDQ) = 0.1 V
2.5
W
BGDDQ Gate Pull−HIGH Resistance
RH_BG
VCC = 12 V, V(BGDDQ) = 11.9 V
3.0
W
BGDDQ Gate Pull−LOW Resistance
RL_BG
VCC = 12 V, V(BGDDQ) = 0.1 V
1.3
W
TG1P5 Gate Pull−HIGH Resistance
RH_TPG
VCC = 12 V, V(TG1P5) = 11.9 V
3.0
W
TG1P5 Gate Pull−LOW Resistance
RL_TPG
VCC = 12 V, V(TG1P5) = 0.1 V
2.5
W
BG1P5 Gate Pull−HIGH Resistance
RH_BPG
VCC = 12 V, V(BG1P5) = 11.9 V
3.0
W
BG1P5 Gate Pull−LOW Resistance
RL_BPG
VCC = 12 V, V(BG1P5) = 0.1 V
1.3
W
VFB1P5 Feedback Voltage, Control Loo
in Regulation
VFB1P5
TA = 0°C to 70°C
Feedback Input Current
I1P5FB
MCH SWITCHING REGULATOR
Oscillator Frequency
F1P5
Oscillator Ramp Amplitude
dV1P5OSC
Minimum Duty Cycle
Dmin_1P5
Maximum Duty Cycle
Dmax_1P5
Soft−Start Pin Current for V1P5 Regulator
0.784
217
(Note 4)
0.8
250
0.816
V
1.0
mA
283
1.3
0
%
100
ISS2
(Note 4)
4. Guaranteed by design, not tested in production.
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6
KHz
Vp−p
8.0
%
mA
NCP5220
TYPICAL OPERATING CHARACTERISTICS
550
1.195
SWITCHING FREQUENCY (kHz)
VFBQ, FEEDBACK VOLTAGE (V)
1.196
1.194
1.193
1.192
1.191
1.190
1.189
1.188
1.187
500
S3 MODE
450
400
350
300
250
S0 MODE
200
0
20
40
60
80
0
Figure 3. VFBQ Feedback Voltage vs. Ambient
Temperature
VFB1P5, FEEDBACK VOLTAGE (V)
0.809
0.807
0.805
0.803
0.801
0.799
0.797
0.795
0
20
40
60
80
−19
2.0 A Sourcing Current
with 10 ms period and
1.0 ms pulse width
−21
80
31.2
2.0 A Sinking Current
with 10 ms period and
1.0 ms pulse width
31
30.8
30.6
30.4
30.2
30
29.8
0
20
40
60
80
TA, AMBIENT TEMPERATURE (°C)
Figure 6. VTT Sink Current Load Regulation
vs. Ambient Temperature
0.03
0.02
0.01
0
−0.01
−22
Sourcing/Sinking Current
with 10 ms period and
1.0 ms pulse width
TA = 25°C
−0.02
−23
−24
60
Figure 4. Oscillation Frequency in S0/S3 vs.
Ambient Temperature
Figure 5. VFB1P5 Feedback Voltage vs. Ambient
Temperature
DVTT, OUTPUT VOLTAGE (VDDQ/2 V)
DVTT, SOURCE CURRENT LOAD REGULATION (mVp−p)
TA, AMBIENT TEMPERATURE (°C)
−20
40
TA, AMBIENT TEMPERATURE (°C)
DVTT, SINK CURRENT LOAD REGULATION (mVp−p)
TA, AMBIENT TEMPERATURE (°C)
20
−0.03
0
20
40
60
80
−0.04
−2.5
−1.5
−0.5
0.5
1.5
2.5
TA, AMBIENT TEMPERATURE (°C)
IVTT, OUTPUT LOAD CURRENT (A)
Figure 7. VTT Source Current Load Regulation vs.
Ambient Temperature
Figure 8. VTT, Output Voltage vs. Load Current
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NCP5220
TYPICAL OPERATING WAVEFORMS
Channel 1: VDDQ Output Voltage, 1.0 V/div
Channel 2: VTT Output Voltage, 1.0 V/div
Channel 3: V1P5 Output Voltage, 1.0V/div
Time Base: 5.0 ms/div
Channel 1: SLP_S3 Pin Voltage, 5.0 V/div
Channel 2: VDDQ Output Voltage, AC−Coupled, 20 mV/div
Channel 3: VTT Output Voltage, AC−Coupled, 100 mV/div
Channel 4: V1P5 Output Voltage, 50 mV/div
Time Base: 10 ms/div
Figure 9. Power−Up Sequence
Figure 10. S0−S3−S0 Transition
500 mA Applied
to VDDQ
417 mA Applied
to VTT
288 mA Applied
to V1P5
Channel 1: SLP_S5 Pin Voltage, 5.0 V/div
Channel 2: VDDQ Output Voltage, 1.0 V/div
Channel 3: VTT Output Voltage, 1.0 V/div
Channel 4: V1P5 Output Voltage, 1.0 V/div
Time Base: 10 ms/div
Channel 1: Current sourced out of VTT, 2.0 A/div
Channel 2: VDDQ Output Voltage, AC−Coupled, 100 mV/div
Channel 3: VTT Output Voltage, AC−Coupled, 20 mV/div
Channel 4: V1P5 Output Voltage, AC−Coupled, 100 mV/div
Time Base: 200 ms/div
Figure 12. VTT Source Current Transient,
0A−2A–0A
Figure 11. S0−S5−S0 Transition
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NCP5220
TYPICAL OPERATING WAVEFORMS
Channel 1: Current Sunk into VTT, 2.0 A/div
Channel 2: VDDQ output Voltage, AC−Coupled, 100 mV/div
Channel 3: VTT Output Voltage, AC−Coupled, 50 mV/div
Channel 4: V1P5 Vutput Voltage, AC−Coupled, 100 mV/div
Time Base: 200 ms/div
Channel 1: Current Sourced into V1P5, 10 A/div
Channel 2: VDDQ Output Voltage, AC−Coupled, 100 mV/div
Channel 3: VTT Output Voltage, AC−Coupled, 100 mV/div
V1P5 Output Voltage, AC−Coupled, 100 mV/div
Time Base: 1.0 ms/div
Figure 13. VTT Sink Current Transient,
0A−2A−0A
Figure 14. VDDQ Source Current Transient,
0A–20A–0A
Channel 1: Current Sourced into VDDQ, 10 A/div
Channel 2: VDDQ Output Voltage, AC−Coupled, 50 mV/div
Channel 3: VTT Output Voltage, AC−Coupled, 100 mV/div
Channel 4: V1P5 Output Voltage, AC−Coupled, 100 mV/div
Time Base: 1.0 ms/div
Channel 1: Current Sourced into VDDQ, 2.0 A/div
Channel 2: VDDQ Output Voltage, AC−Coupled, 50 mV/div
Time Base: 200 ms/div
Figure 16. S3 Mode without 12VATX,
0A–2A–0A
Figure 15. V1P5 Source Current Transient,
0A–12A–0A
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NCP5220
DETAILED OPERATION DESCRIPTIONS
General
S5 to S0 Mode Power−Up Sequence
The NCP5220 3−in−1 PWM Dual Buck Linear DDR
Power Controller contains two high efficiency PWM
controllers and an integrated two−quadrant linear regulator.
The VDDQ supply is produced by a PWM switching
controller with two external N−Ch FETs. The VTT
termination voltage is an integrated linear regulator with
sourcing and sinking current capability which tracks at ½
VDDQ. The MCH core voltage is created by the secondary
switching controller.
The inclusion of soft−start, supply undervoltage monitors
and thermal shutdown, makes this device a total power
solution for the MCH and DDR memory system. This device
is packaged in a DFN−20.
The ACPI control logic is enabled by the assertion of
_VREFGD. Once the ACPI control is activated, the power−
up sequence starts by waking up the 5VDUAL voltage
monitor block. If the 5VDUAL supply is within the preset
levels, the BOOT undervoltage monitor block is then
enabled. After 12VATX is ready and the BOOT UVLO is
asserted LOW, the ACPI control triggers this device from S5
shutdown mode into S0 normal operating mode by
activating the soft−start of DDQ switching regulator,
providing SLP_S3 and SLP_S5 remain HIGH.
Once the DDQ regulator is in regulation and the soft−start
interval is completed, the _InRegDDQ signal is asserted
HIGH to enable the VTT regulator as well as the V1P5
switching regulator.
ACPI Control Logic
The ACPI control logic is powered by the 5VDUAL
supply. It accepts external control at the SLP_S3 input and
internal supply voltage monitoring signals from two
UVLOs to decode the operating mode in accordance with
the state transition diagram in Figure 18.
These UVLOs monitor the external supplies, 5VDUAL
and 12VATX, through 5VDUAL and BOOT pins
respectively. Two control signals, _5VDUALGD and
_BOOTGD, are asserted when the supply voltages are good.
When the device is powered up initially, it is in the S5
shutdown mode to minimize the power consumption. When
all three supply voltages are good with SLP_S3 and SLP_S5
remaining HIGH, the device enters the S0 normal operating
mode. The transition of SLP_S3 from HIGH to LOW while
in the S0 mode, triggers the device into the S3 sleep mode.
In S3 mode the 12VATX supply collapses. On transition of
SLP_S3 from LOW to HIGH, the device returns to S0 mode.
The IC can re−enter S5 mode by setting SLP_S5 LOW. A
timing diagram is shown in Figure 17.
Table 1 summarizes the operating states of all the
regulators, as well as the conditions of the output pins.
DDQ Switching Regulator
In S0 mode the DDQ regulator is a switching synchronous
rectification buck controller driving two external power
N−Ch FETs to supply up to 20 A. It employs voltage mode
fixed frequency PWM control with external compensation
switching at 250kHz ± 13.2%. As shown in Figure 2, the
VDDQ output voltage is divided down and fed back to the
inverting input of an internal amplifier through the FBDDQ
pin to close the loop at VDDQ = VFBQ × (1 + R1/R2). This
amplifier compares the feedback voltage with an internal
reference voltage of 1.190 V to generate an error signal for
the PWM comparator. This error signal is compared with a
fixed frequency RAMP waveform derived from the internal
oscillator to generate a pulse−width−modulated signal. This
PWM signal drives the external N−Ch FETs via the
TG_DDQ and BG_DDQ pins. External inductor L and
capacitor COUT1 filter the output waveform. When the IC
leaves the S5 state, the VDDQ output voltage ramps up at a
soft−start rate controlled by the capacitor at the SS pin.
When the regulation of VDDQ is detected in S0 mode,
_INREGDDQ goes HIGH to notify the control block.
In S3 standby mode, the switching frequency is doubled
to reduce the conduction loss in the external N−Ch FETs.
Internal Bandgap Voltage Reference
An internal bandgap reference is generated whenever
5VDUAL exceeds 2.7 V. Once this bandgap reference is in
regulation, an internal signal _VREFGD will be asserted.
Table 1. Mode, Operation and Output Pin Conditions
OPERATING CONDITIONS
OUTPUT PIN CONDITIONS
MODE
DDQ
VTT
MCH
TG_DDQ
BG_DDQ
TP_1P5
BG_1P5
S0
Normal
Normal
Normal
Normal
Normal
Normal
Normal
S3
Standby
H−Z
OFF
Standby
Standby
Low
Low
S5
OFF
H−Z
OFF
Low
Low
Low
Low
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NCP5220
pin directly, VTT voltage is designed to automatically track
at the half of VDDQ. This regulator is stable with any value
of output capacitor greater than 470 mF, and is insensitive to
ESR ranging from 1 mW to 400 mW.
For enhanced efficiency, an active synchronous switch is
used to eliminate the conduction loss contributed by the
forward voltage of a diode or Schottky diode rectifier.
Adaptive non−overlap timing control of the complementary
gate drive output signals is provided to reduce
shoot−through current that degrades efficiency.
Fault Protection of VTT Active Terminator
To provide protection for the internal FETs, bi−directional
current limit preset at 2.4 A magnitude is implemented. The
VTT provides a soft−start function during start up.
Tolerance of VDDQ
Both the tolerance of VFBQ and the ratio of the external
resistor divider R1/R2 impact the precision of VDDQ. With
the control loop in regulation, VDDQ = VFBQ × (1 +
R1/R2). With a worst case (for all valid operating
conditions) VFBQ tolerance of ±1.5%, a worst case range of
±2% for VDDQ will be assured if the ratio R1/R2 is
specified as 1.100 ±1%.
MCH Switching Regulator
The secondary switching regulator is identical to the DDQ
regulator except the output is 10 A. No fault protection is
implemented and the soft−start timing is twice as fast with
respect to CSS.
BOOT Pin Supply Voltage
Fault Protection of VDDQ Regulator
In typical application, a flying capacitor is connected
between SWDDQ and BOOT pins. In S0 mode, 12VATX is
tied to BOOT pin through a Schottky diode as well. A 13 V
Zener clamp circuit must clamp this boot strapping voltage
produced by the flying capacitor in S0 mode.
In S3 mode the 12VATX is collapsed and the BOOT
voltage is created by the Schottky diode between 5VDUAL
and BOOT pins as well as the flying capacitor. The
BOOT_UVLO works specially. The _BOOTGD goes low
and the IC remains in S3 mode.
In S0 mode, an internal voltage (VOCP) = 5VDUAL – 0.8
sets the current limit for the high−side switch. The voltage
VOCP pin is compared to the voltage at SWDDQ pin when
the high−side gate drive is turned on after a fixed period of
blanking time to avoid false current limit triggering. When
the voltage at SWDDQ is lower than VOCP, an overcurrent
condition occurs and all regulators are latched off to protect
against overcurrent. The IC will be powered up again if one
of the supply voltages, 5VDUAL, SLP_S5 or 12VATX, is
recycled. The main purpose is for fault protection, not for
precise current limit.
In S3 mode, this overcurrent protection feature is
disabled.
Thermal Consideration
Assuming an ambient temperature of 50°C, the maximum
allowed dissipated power of DFN−20 is 2.8 W, which is
enough to handle the internal power dissipation in S0 mode.
To take full advantage of the thermal capability of this
package, the exposed pad underneath must be soldered
directly onto a PCB metal substrate to allow good
thermal contact.
Feedback Compensation of VDDQ Regulator
The compensation network is shown in Figure 2.
VTT Active Terminator
The VTT active terminator is a 2 quadrant linear regulator
with two internal N−Ch FETs to provide current sink and
source capability up to 2.0 A. It is activated only when the
DDQ regulator is in regulation in S0 mode. It draws power
from VDDQ with the internal gate drive power derived from
5VDUAL. While VTT output is connecting to the FBVTT
Thermal Shutdown
When the junction temperature of the IC exceeds 145°C,
the entire IC is shutdown. When the junction temperature
drops below 120°C, the chip resumes normal operation.
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NCP5220
5VSTBY
or
5VDUAL
12 V
SLP_S5
Switching
Frequency
Doubles
SLP_S3
SS Pin
DDQ−S0
VTT
MCH
State
1
2
3
4
5
6
7
8
9
SO
10
S3
11 12 13
14
SO
15 16
17
S5
2. 5VSTBY or 5VSTB is the ultimate chip enable, SLP_S5 and SLP_S3 go HIGH. This supply has to be up first to
ensure gates are in known state.
3. 12 V supply ramp.
4. DDQ will ramp with the tracking of SS pin, timing is 1.2 * CSS / 4 m (sec).
5. DDQ SS is completed, then SS pin is released from DDQ. SS pin is shorted to ground.
5. MCH ramps with the tracking of SS pin ramp, timing is 0.8 * CSS / 8 m (sec). VTT start up with current limit.
6. MCH SS is completed, then SS pin is released from MCH, SS pin is shorted to ground. S0 Mode.
7. S3 MODE −− SLP_S3 = L.
8. VTT and MCH will be turned off.
9. 12 V ramps to 0 V.
10. Standard S3 State.
11. SLP_S3 goes HIGH.
12. 12 V ramps back to regulation.
13. 12 V UVLO = L and SLP_S3 = H. MCH ramps with SS pin, timing is 0.8 * CSS / 8 m (sec). VTT rises.
14. S0 Mode.
15. S5 Mode −− SLP_S5 = L.
16. DDQ, VTT and MCH Turned OFF.
17. S5 Mode.
Figure 17. NCP5220 Power−Up and Power−Down
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NCP5220
S5
SLP_S3 = 1 AND
SLP_S5 = 1 AND
_BOOTGD = 1
SLP_S5 = 0 OR
(SLP_S3 = 1 AND
_BOOTGD = 0)
SLP_S5 = 0
S0
SLP_S3 = 1 AND
SLP_S5 = 1 AND
_BOOTGD = 1
SLP_S3 = 0 AND
SLP_S5 = 1
NOTE: 5VDUAL is assumed to be in good conditions in any mode.
All possible state transitions are shown.
All unspecified inputs do not cause any state change.
S3
Figure 18. Transitions State Diagram of NCP5220
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13
NCP5220
APPLICATION INFORMATION
Application Circuit
Power MOSFET Selection
Figure 20, on the following page, shows the typical
application circuit for NCP5220. The NCP5220 is
specifically designed as a total power solution for the MCH
and DDR memory system. This diagram contains NCP5220
for driving four external N−Ch FETs to form the DDR
memory supply voltage (VDDQ) and the MCH regulator.
Power MOSFETs are chosen by balancing the cost with
the requirements for the current load of the memory system
and the efficiency of the converter provided. The selections
criteria can be based on drain−source voltage, drain current,
on−resistance RDS(on) and input gate capacitance. Low
RDS(on) and high drain current power MOSFETs are usually
preferred to achieve the high current requirement of the
DDR memory system and MCH, as well as the high
efficiency of the converter. The tradeoff is a corresponding
increase in the input gate capacitor of the power MOSFETs.
Output Inductor Selection
The value of the output inductor is chosen by balancing
ripple current with transient response capability. A value of
1.7 mH will yield about 3.0 A peak−peak ripple current when
converting from 5.0 V to 2.5 V at 250 kHz. It is important
that the rated inductor current is not exceeded during full
load, and that the saturation current is not less than the
expected peak current. Low ESR inductors may be required
to minimize DC losses and temperature rise.
PCB Layout Considerations
Input capacitors for PWM power supplies are required to
provide a stable, low impedance source node for the buck
regulator to convert from. The usual practice is to use a
combination of electrolytic capacitors and multi−layer
ceramic capacitors to provide bulk capacitance and high
frequency noise suppression. It is important that the
capacitors are rated to handle the AC ripple current at the
input of the buck regulators, as well as the input voltage. In
the NCP5220 the DDQ and MCH regulators are interleaved
(out of phase by 180 degrees) to reduce the peak AC input
current.
With careful PCB layout the NCP5220 can supply 20 A or
more of current. It is very important to use wide traces or
large copper shapes to carry current from the input node
through the MOSFET switches, inductor and to the output
filters and load. Reducing the length of high current nodes
will reduce losses and reduce parasitic inductance. It is
usually best to locate the input capacitors the MOSFET
switches and the output inductor in close proximity to
reduce DC losses, parasitic inductance losses and radiated
EMI.
The sensitive voltage feedback and compensation
networks should be placed near the NCP5220 and away
from the switch nodes and other noisy circuit elements.
Placing compensation components near each other will
minimize the loop area and further reduce noise
susceptibility.
Output Capacitor Selection
Optional Boost Voltage Configuration
Output capacitors are chosen by balancing the cost with
the requirements for low output ripple voltage and transient
voltage. Low ESR electrolytic capacitors can be effective at
reducing ripple voltage at 250 kHz. Low ESR ceramic
capacitors are most effective at reducing output voltage
excursions caused by fast load steps of system memory and
the memory controller.
The charge pump circuit in Figure 19 can be used instead
of boost voltage scheme of Figure 20. The advantage in
Figure 19 is the elimination of the requirement for the Zener
clamp.
Input Capacitor Selection
12VATX
TP2
5VDUAL
TP2
D2
BAT54HT1
D1
BAT54HT1
D2
NCP5220
SW_DDQ 20
BG_DDQ 19
TG_DDQ 18
BOOT 17
5VDUAL 16
15
COMP_1P5
SLP_S3 14
TG_1P5 13
12
BG_1P5
11
BAT54HT1
5VDUAL
4
R2
4.7 1
Q2
3 NTD40N03
C4
100 nF
L
R3
1k
R4
4.7
TP5
VDDQ
1
4 DPAK
Q2
NTD40N03
3
C6
4.7
mF
GND_1P5
Figure 19. Charge Pump Circuit at BOOT Pin
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14
+
C7
2200
mF
+
C25
2200
mF
R15 2.5 VDDQ
1k
VTT
1.25 VTT
TP7
R16
1k
VDDQ
C12
4.7
mF
R5
2.2 k
+
C8
10 nF
C13
470
mF
+
C24
470
mF
SGND
R8
2k
+ C1
33
nF
C10 R7
6.8 nF 20 k
SGND
R6 C9
8 100 nF
R18
51 k
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11
12
SGND
4 DPAK
Q2
85N02R
3
1
1
D1
BAT54HT1
C18 R13
100 nF 8
R14
2k
C6
4.7
mF
VDDQ
L3
1.8 mH
R6
2.2 k
C3
3300
mF
SGND
C17 R12
6.8 nF 20 k
+
Filtered 5VDUAL
C22
10
mF
D2
+
+
C25
2200
mF
VCMH
+
C26
C14
+
2200
4.7
mF
mF
C7
2200
mF
C15
2200
mF
R15
1k
C5
470 mF
12VATX
TP2
ZENER
BAT54HT1
MMSZ13T1
AGND to PGND
L2 1.8 mH
C4
22 nF
4 DPAK
Q5
C16
40N03R 10 nF
3
DPAK
Q4
3 40N03R
4
+ C21
100
mF
COMP_1P5
R10
4.7
R9
4.7
C11
220
nF
R4
1 1
VREF = 800 mV
AGND to PGND
GND_1P5
BG_1P5
TG_1P5 13
COMP_1P5 15
SLP_S3 14
R3
1k
DPAK
Q1
3 85N02R
4
C23
10 mF
Filtered 5VDUAL
R2
2.2 1
+ C2
3300
mF
BOOT 17
5VDUAL
5VDUAL 16
TG_DDQ 18
BG_DDQ 19
SW_DDQ 20
L1
1 mF
Figure 20. NCP5220 Typical Application Circuit
7 AGND
SGND 8 FBVTT
C20
9 SLP_S5
470
10 FB1P5
mF
VDDQ
4 PGND
5 VTT
6 VDDQ
1 COMP
VREF = 1.20 V 2
FBDDQ
3 SS
U1
NCP5220
5VDUAL
5VDUAL
TP2
R17
1k
GND
TP16
1.5 VMCH
TP8
2.5 VDDQ
TP5
NCP5220
NCP5220
Table 2. Bill of Material of NCP5220 Application Circuit
Reference
Design
Description
Value
Qty
Part Number
Manufactur
Q1, Q2
Power MOSFET N−Channel
24 V, 4.8 mW, 85 A
2
NTD85N02R
ON Semiconductor
Q3, Q4
Power MOSFET N−Channel
25 V, 12.6 mW, 40 A
2
NTD40N03R
ON Semiconductor
D1, D2
Rectifier Schottky Diode
30 V
2
BAT54HT1
ON Semiconductor
3−in−1 PWM Dual Buck
and Linear Power Controller
1
NCP5220
ON Semiconductor
Zener Diode
13 V, 0.5 W
1
MMSZ13T1
ON Semiconductor
U1
Zener
Controller
L1
Toroidal Choke
1.0 mH, 25 A
1
T60−26(6T)
L2, L3
Toroidal Choke
1.8 mH, 25 A
2
T50−26B(6T)
C2, C3
Aluminum Electrolytic Capacitor
3300 mF, 6.3 V
2
EEUFJ0J332U
Panasonic
C5
Aluminum Electrolytic Capacitor
470 mF, 35 V
1
EEUFC1V471
Panasonic
C21
Aluminum Electrolytic Capacitor
100 mF, 50 V
1
EEUFC1H101
Panasonic
C20
Aluminum Electrolytic Capacitor
470 mF, 16 V
1
EEUFC1C471
Panasonic
C13, C24
Aluminum Electrolytic Capacitor
470 mF, 10 V
2
EEUFC1A471
Panasonic
C7, C25,
C15, C26
Aluminum Electrolytic Capacitor
2200 mF, 6.3 V
4
EEUFC0J222S(H)
Panasonic
C11
Ceramic Capacitor
220 nF, 10 V
1
ECJ1VB1A224K
Panasonic
C6, C12,
C14
Ceramic Capacitor
4.7 mF, 6.3 V
3
ECJHVB0J475M
Panasonic
C22, C23
Ceramic Capacitor
10 mF, 25 V
2
ECJ4YB1E106M
Panasonic
C4
Ceramic Capacitor
22 nF, 25 V
1
ECJ1VB1E223K
Panasonic
C10, C17
Ceramic Capacitor
6.8 nF, 50 V
2
ECJ1VB1H682K
Panasonic
C9, C18
Ceramic Capacitor
100 nF, 16 V
2
ECJ1VB1C104K
Panasonic
C8, C16
Ceramic Capacitor
10 nF, 50 V
2
ECJ1VB1H103K
Panasonic
C1
Ceramic Capacitor
33 nF, 25 V
1
ECJ1VB1E333K
Panasonic
R2
Resistor
2.2 W
1
R4
Resistor
1.0 W
1
R9, R10
Resistor
4.7 W
2
R3, R15,
R16, R17
Resistor
1.0 kW
4
R7, R12
Resistor
20 kW
2
R6, R13
Resistor
8.2 W
2
R8, R14
Resistor
2.0 kW
2
R5, R11
Resistor
2.2 kW
2
R18
Resistor
51 kW
1
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16
NCP5220
PACKAGE DIMENSIONS
20 PIN DFN, DUAL−SIDED, 5x6 mm
MN SUFFIX
CASE 505AB−01
ISSUE A
NOTES:
1. DIMENSIONS AND TOLERANCING PER
ASME Y14.5M, 1994.
2. DIMENSIONS IN MILLIMETERS.
3. DIMENSION b APPLIES TO PLATED
TERMINALS AND IS MEASURED BETWEEN
0.25 AND 0.30 MM FROM TERMINAL
4. COPLANARITY APPLIES TO THE EXPOSED
PAD AS WELL AS THE TERMINALS.
A
D
B
PIN 1 LOCATION
E
2X
DIM
A
A1
A2
A3
b
D
D2
E
E2
e
K
L
0.15 C
2X
TOP VIEW
0.15 C
0.10 C
A2
A
0.08 C
A1
SIDE VIEW (A3)
C
SEATING
PLANE
D2
20X
20X
L
e
1
10
E2
K
20
11
20X
b
0.10 C A B
0.05 C
NOTE 3
BOTTOM VIEW
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MILLIMETERS
MIN
MAX
0.80
1.00
0.00
0.05
0.65
0.75
0.20 REF
0.23
0.28
6.00 BSC
3.98
4.28
5.00 BSC
2.98
3.28
0.50 BSC
0.20
−−−
0.50
0.60
NCP5220
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death
associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
LITERATURE FULFILLMENT:
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P.O. Box 61312, Phoenix, Arizona 85082−1312 USA
Phone: 480−829−7710 or 800−344−3860 Toll Free USA/Canada
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Phone: 81−3−5773−3850
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For additional information, please contact your
local Sales Representative.
NCP5220/D
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