1.5 A, 170 kHz, Low Voltage Buck Regulator with Synchronization Capability

NCP1546
1.5 A, 170 kHz, Buck
Regulator with
Synchronization Capability
The NCP1546 is a 1.5 A buck regulator IC operating at a
fixed−frequency of 170 kHz. The device uses the V2t control
architecture to provide unmatched transient response, the best overall
regulation and the simplest loop compensation for today’s high−speed
logic. The NCP1546 accommodates input voltages from 4.0 V to 40 V
and contains synchronization circuitry.
The on−chip NPN transistor is capable of providing a minimum of
1.5 A of output current, and is biased by an external “boost” capacitor
to ensure saturation, thus minimizing on−chip power dissipation.
Protection circuitry includes thermal shutdown, cycle−by−cycle
current limiting and frequency foldback.
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MARKING
DIAGRAM
1
18
NCP1546
AWLYYWW G
G
1
Features
•
•
•
•
•
•
•
•
•
•
•
18−LEAD DFN
MN SUFFIX
CASE 505
V2
Architecture Provides Ultra−Fast Transient Response, Improved
Regulation and Simplified Design
Wide Operating Range: 4 V to 40 V
2.0% Error Amp Reference Voltage Tolerance
Switch Frequency Decrease of 4:1 in Short Circuit Conditions
Reduces Short Circuit Power Dissipation
BOOST Lead Allows “Bootstrapped” Operation to Maximize
Efficiency
Sync Function for Parallel Supply Operation or Noise Minimization
Shutdown Lead Provides Power−Down Option
1.0 mA Quiescent Current During Power−Down
Thermal Shutdown
Soft−Start
These are Pb−Free Devices
18
8
SOIC−8
D SUFFIX
CASE 751
8
1
1
P1546
ALYWE
G
A
= Assembly Location
WL, L
= Wafer Lot
YY, Y
= Year
WW, W = Work Week
E
= Automotive Grade
G
= Pb−Free Package
(Note: Microdot may be in either location)
ORDERING INFORMATION
Device
Package
Shipping†
NCP1546DG
SOIC−8
(Pb−Free)
98 Units / Rail
NCP1546DR2G
SOIC−8 2500 / Tape & Reel
(Pb−Free)
NCP1546MNR2G
DFN18 2500 / Tape & Reel
(Pb−Free)
†For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.
© Semiconductor Components Industries, LLC, 2011
July, 2011 − Rev. 5
1
Publication Order Number:
NCP1546/D
NCP1546
R6
100k
D1
Vout (3.3 V)
C3
100 mF
R3
162
+
C1
1 mF
U1
Vsw
SHDNB
NC
NC
NC
NC
SYNC
GND
D2
EPAD
L1
27 mH
Vin (7 V to 16 V)
Vin
BOOST
NC
NC
NC
NC
Vc
Vfb
C5
0.1 mF
+ C2
NCP1546
330 mF
C4
0.1 mF
R2
100
SHDNB
SYNC
Figure 1. Application Diagram, 4.5 V − 16 V to 3.3 V @ 1.0 A Converter
MAXIMUM RATINGS*
Rating
Value
Unit
45
V
0 to +150
°C
260 peak
(Note 2)
°C
−65 to +150
°C
(Human Body Model)
(Machine Model)
(Charge Device Model)
2.0
200
>1.0
kV
V
kV
18−Lead DFN Junction−to−Ambient, RqJA
SO−8 Junction−to−Ambient, RqJA (Note 3)
16
100
°C/W
Peak Transient Voltage (31 V Load Dump @ VIN = 14 V)
Operating Junction Temperature Range, TJ
Lead Temperature Soldering:
Reflow: (Note 1)
Storage Temperature Range, TS
ESD
Package Thermal Resistance
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended Operating Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
*The maximum package power dissipation must be observed.
1. 60 second maximum above 183°C.
2. −5°C/0°C allowable conditions.
3. 1 in2, 1 oz copper area used for heatsinking.
MAXIMUM RATINGS (Voltages are with respect to GND)
Pin Name
VMax
VMIN
ISOURCE
ISINK
VIN (DC)*
40 V
−0.3 V
N/A
4.0 A
BOOST
40 V
−0.3 V
N/A
100 mA
VSW
40 V
−0.6 V/−1.0 V, t < 50 ns
4.0 A
10 mA
VC
7.0 V
−0.3 V
1.0 mA
1.0 mA
SHDNB
7.0 V
−0.3 V
1.0 mA
1.0 mA
SYNC
7.0 V
−0.3 V
1.0 mA
1.0 mA
VFB
7.0 V
−0.3 V
1.0 mA
1.0 mA
*See table above for load dump.
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NCP1546
PACKAGE PIN DESCRIPTION
SO−8
DFN−18
PIN SYMBOL
1
1
BOOST
2
2−4
VIN
This pin is the main power input to the IC.
3
5−7
VSW
This is the connection to the emitter of the on−chip NPN power transistor and serves as the
switch output to the inductor. This pin may be subjected to negative voltages during switch off−
time. A catch diode is required to clamp the pin voltage in normal operation. This node can
stand −1.0 V for less than 50 ns during switch node flyback.
4
8
SHDNB
The shutdown pin is active low and TTL compatible. The IC goes into sleep mode, drawing less
than 1.0 mA when the pin voltage is pulled below 1.0 V.
This pin should be pulled up to VCC with a resistor.
5
10
SYNC
This pin provides the synchronization input.
6
13
GND
Power return connection for the IC.
7
16
VFB
The FB pin provides input to the inverting input of the error amplifier. If VFB is lower than 0.29 V,
the oscillator frequency is divided by four, and current limit folds back to about 1 ampere. These
features protect the IC under severe overcurrent or short circuit conditions.
8
17
VC
The VC pin provides a connection point to the output of the error amplifier and input to the PWM
comparator. Driving of this pin should be avoided because on−chip test circuitry becomes active
whenever current exceeding 0.5 mA is forced into the IC.
−
9, 11, 12,
14, 15, 18
NC
No Connection
−
−
−
FUNCTION
The BOOST pin provides additional drive voltage to the on−chip NPN power transistor. The
resulting decrease in switch on voltage increases efficiency.
Exposed die attach pad. Internally connected to GND. External connection to GND is optional.
PIN CONNECTIONS
1
8
BOOST
VIN
VFB
VSW
GND
SHDNB
BOOST
VIN
VIN
VIN
Vsw
VSW
VSW
SHDNB
NC
VC
SYNC
SO−8
1
2
3
4
5
6
7
8
9
18
17
16
15
14
13
12
11
10
18−Lead DFN
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3
NC
VC
VFB
NC
NC
GND
NC
NC
SYNC
NCP1546
ELECTRICAL CHARACTERISTICS (0°C < TJ < 70°C, 4.5 V< VIN < 40 V; unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
Oscillator
Operating Frequency
−
153
170
187
kHz
Frequency Line Regulation
−
−
0.05
0.15
%/V
Maximum Duty Cycle
−
85
90
95
%
VFB Frequency Foldback Threshold
−
0.29
0.32
0.36
V
5.0
9.0
17
mV/ms
−
100
200
ns
PWM Comparator
Slope Compensation Voltage
Fix VFB, DVC/DTON
Minimum Output Pulse Width
VFB to VSW
Power Switch
Current Limit
VFB > 0.36 V
1.6
2.3
3.0
A
Foldback Current
VFB < 0.29 V
0.9
1.5
2.1
A
Saturation Voltage
IOUT = 1.5 A, VBOOST = VIN + 2.5 V
0.4
0.7
1.0
V
Current Limit Delay
Note 4
−
120
160
ns
1.244
1.270
1.296
V
−
40
−
dB
−
0.02
0.1
mA
Error Amplifier
Internal Reference Voltage
Reference PSRR
−
Note 4
FB Input Bias Current
−
Output Source Current
VC = 1.270 V, VFB = 1.0 V
15
25
35
mA
Output Sink Current
VC = 1.270 V, VFB = 2.0 V
15
25
35
mA
Output High Voltage
VFB = 1.0 V
1.39
1.46
1.53
V
Output Low Voltage
VFB = 2.0 V
5.0
20
60
mV
Unity Gain Bandwidth
Note 4
−
500
−
kHz
Open Loop Amplifier Gain
Note 4
−
70
−
dB
Amplifier Transconductance
Note 4
−
6.4
−
mA/V
190
−
355
kHz
−
360
485
mA
−
0.9
1.5
1.9
V
−
1.0
1.3
1.6
V
Sync
Sync Frequency Range
Sync Pin Bias Current
−
VSYNC = 5.0 V
Sync Threshold Voltage
Shutdown
Shutdown Threshold Voltage
Thermal Shutdown
Overtemperature Trip Point
Note 4
175
185
195
°C
Thermal Shutdown Hysteresis
Note 4
−
42
−
°C
4. Guaranteed by design, not 100% tested in production.
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NCP1546
ELECTRICAL CHARACTERISTICS (continued) (0°C < TJ < 70°C, 4.5 V< VIN < 40 V; unless otherwise specified.)
Characteristic
Test Conditions
Min
Typ
Max
Unit
General
Quiescent Current
ISW = 0 A
−
4.0
7.5
mA
Shutdown Quiescent Current
VSHDNB = 0 V
−
1.0
5.0
mA
Boost Operating Current
VBOOST − VSW = 2.5 V
6.0
15
40
mA/A
Minimum Boost Voltage
Note 5
−
−
2.5
V
Startup Voltage
−
3.0
3.5
4.0
V
Minimum Output Current
−
−
7.0
12
mA
5. Guaranteed by design, not 100% tested in production.
SHDNB
SYNC
VIN
2.9 V LDO
Voltage
Regulator
Shutdown
Comparator
+
Artificial
Ramp
−
Thermal
Shutdown
Oscillator
BOOST
+
1.3 V −
S
Q
R
Output
Driver
VSW
∑
+
Current
Limit Comparator
−
PWM Comparator
1.46 V
+
SHDNB
IREF
−
−
VFB
−
+
+
0.32 V −
+
1.27 V
+
−
Frequency
and Current
Limit Foldback
Error
Amplifier
VC
Figure 2. Block Diagram
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IFOLDBACK
GND
NCP1546
APPLICATIONS INFORMATION
THEORY OF OPERATION
current of the inductor L1 and the ESR (equivalent series
resistor) of the output capacitor C1.
The slope compensation signal is a fixed voltage ramp
provided by the oscillator. Adding this signal eliminates
subharmonic oscillation associated with the operation at
duty cycle greater than 50%. The artificial ramp also ensures
the proper PWM function when the output ripple voltage is
inadequate. The slope compensation signal is properly sized
to serve it purposes without sacrificing the transient
response speed.
Under load and line transient, not only the ramp signal
changes, but more significantly the DC component of the
feedback voltage varies proportionally to the output voltage.
FFB path connects both signals directly to the PWM
comparator. This allows instant modulation of the duty cycle
to counteract any output voltage deviations. The transient
response time is independent of the error amplifier
bandwidth. This eliminates the delay associated with error
amplifier and greatly improves the transient response time.
The error amplifier is used here to ensure excellent DC
accuracy.
V2 Control
The NCP1546 buck regulator provides a high level of
integration and high operating frequencies allowing the
layout of a switch−mode power supply in a very small board
area. This device is based on the proprietary V2 control
architecture. V2 control uses the output voltage and its ripple
as the ramp signal, providing an ease of use not generally
associated with voltage or current mode control. Improved
line regulation, load regulation and very fast transient
response are also major advantages.
S1
VIN
L1
VO
Duty Cycle
C1
D1
R1
Buck
Controller
Oscillator
Slope
Comp
Error Amplifier
FFB
The NCP1546 has a transconductance error amplifier,
whose non−inverting input is connected to an Internal
Reference Voltage generated from the on−chip regulator.
The inverting input connects to the VFB pin. The output of
the error amplifier is made available at the VC pin. A typical
frequency compensation requires only a 0.1 mF capacitor
connected between the VC pin and ground, as shown in
Figure 1. This capacitor and error amplifier’s output
resistance (approximately 8.0 MW) create a low frequency
pole to limit the bandwidth. Since V2 control does not
require a high bandwidth error amplifier, the frequency
compensation is greatly simplified.
The VC pin is clamped below Output High Voltage. This
allows the regulator to recover quickly from over current or
short circuit conditions.
)
Latch S
R
R2
SFB
−
+
VC
−
+
PWM Comparator
V2
Control
Error
Amplifier
VREF
+
−
Figure 3. Buck Converter with V2 Control.
As shown in Figure 3, there are two voltage feedback
paths in V2 control, namely FFB(Fast Feedback) and
SFB(Slow Feedback). In FFB path, the feedback voltage
connects directly to the PWM comparator. This feedback
path carries the ramp signal as well as the output DC voltage.
Artificial ramp derived from the oscillator is added to the
feedback signal to improve stability. The other feedback
path, SFB, connects the feedback voltage to the error
amplifier whose output VC feeds to the other input of the
PWM comparator. In a constant frequency mode, the
oscillator signal sets the output latch and turns on the switch
S1. This starts a new switch cycle. The ramp signal,
composed of both artificial ramp and output ripple,
eventually comes across the VC voltage, and consequently
resets the latch to turn off the switch. The switch S1 will turn
on again at the beginning of the next switch cycle. In a buck
converter, the output ripple is determined by the ripple
Oscillator and Sync Feature
The on−chip oscillator is trimmed at the factory and
requires no external components for frequency control. The
high switching frequency allows smaller external
components to be used, resulting in a board area and cost
savings. The tight frequency tolerance simplifies magnetic
components selection. The switching frequency is reduced
to no more than 25% of the nominal value when the VFB pin
voltage is below Frequency Foldback Threshold. In short
circuit or over−load conditions, this reduces the power
dissipation of the IC and external components.
The oscillator frequency varies with junction
temperature, as seen in the following graph.
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NCP1546
When the switch turns on, the VSW voltage is equal to the
VIN minus switch Saturation Voltage. In the buck regulator,
the VSW voltage swings to one diode drop below ground
when the power switch turns off, and the inductor current is
commutated to the catch diode. Due to the presence of high
pulsed current, the traces connecting the VSW pin, inductor
and diode should be kept as short as possible to minimize the
noise and radiation. For the same reason, the input capacitor
should be placed close to the VIN pin and the anode of the
diode.
The saturation voltage of the power switch is dependent
on the switching current, as shown in Figure 7.
fOSC, FREQUENCY (%)
102.5
100
97.5
95
92.5
90
−40 −20
0.7
0
20 40 60 80 100 120 140
TJ, JUNCTION TEMPERATURE (°C)
0.6
Figure 4. Oscillator Frequency Versus Junction
Temperature
Sync
10k
±33%
VIN − VSW (V)
An external clock signal can sync the NCP1546 to a higher
frequency. The SYNC pin equivalent input circuit is shown
in Figure 5.
0.5
0.4
0.3
0.2
50k
±33%
VZ = 11V
to 20V
0.1
50k
±33%
0
GND
0
0.5
1.0
SWITCHING CURRENT (A)
1.5
Figure 7. The Saturation Voltage of the Power Switch
Increases with the Conducting Current
Figure 5.
The NCP1546 contains pulse−by−pulse current limiting
to protect the power switch and external components. When
the peak of the switching current reaches the Current Limit,
the power switch turns off after the Current Limit Delay. The
switch will not turn on until the next switching cycle. The
current limit threshold is independent of switching duty
cycle. The maximum load current, given by the following
formula under continuous conduction mode, is less than the
Current Limit due to the ripple current.
The rising edge of the sync pulse turns on the power
switch to start a new switching cycle, as shown in Figure 6.
There is approximately 0.5 ms delay between the rising edge
of the sync pulse and rising edge of the VSW pin voltage. The
sync threshold is TTL logic compatible, and duty cycle of
the sync pulses can vary from 10% to 90%. The frequency
foldback feature is disabled during the sync mode.
V (V * VO)
IO(MAX) + ILIM * O IN
2(L)(VIN)(fs)
where:
fS = switching frequency,
ILIM = current limit threshold,
VO = output voltage,
VIN = input voltage,
L = inductor value.
When the regulator runs under current limit, the
subharmonic oscillation may cause low frequency
oscillation, as shown in Figure 8. Similar to current mode
control, this oscillation occurs at the duty cycle greater than
50% and can be alleviated by using a larger inductor value.
The current limit threshold is reduced to Foldback Current
when the FB pin falls below Foldback Threshold. This
feature protects the IC and external components under the
power up or over−load conditions.
Figure 6. A NCP1546 Buck Regulator is Synchronized
to an External 350 kHz Pulse Signal
Power Switch and Current Limit
The collector of the built−in NPN power switch is
connected to the VIN pin, and the emitter to the VSW pin.
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NCP1546
BOOST PIN CURRENT (mA)
30
25
20
15
10
5
0
0
0.5
1.0
SWITCHING CURRENT (A)
1.5
Figure 9. The Boost Pin Current Includes 7.0 mA
Pre−Driver Current and Base Current when the
Switch is Turned On. The Beta Decline of the
Power Switch Further Increases the Base
Current at High Switching Current
Figure 8. The Regulator in Current Limit
BOOST Pin
The BOOST pin provides base driving current for the
power switch. A voltage higher than VIN provides required
headroom to turn on the power switch. This in turn reduces
IC power dissipation and improves overall system
efficiency. The BOOST pin can be connected to an external
boost−strapping circuit which typically uses a 0.1 mF
capacitor and a 1N914 or 1N4148 diode, as shown in Figure
1. When the power switch is turned on, the voltage on the
BOOST pin is equal to
Shutdown
The internal power switch will not turn on until the VIN
pin rises above the Startup Voltage. This ensures no
switching will occur until adequate supply voltage is
provided to the IC. Refer to Figure 10 for the SHDNB
(shutdown−bar) pin input circuit.
SHDNB
VBOOST + VIN ) VO * VF
1 mA to 10 mA
where:
VF = diode forward voltage.
The anode of the diode can be connected to any DC
voltage as well as the regulated output voltage (Figure 1).
However, the maximum voltage on the BOOST pin shall not
exceed 40 V.
As shown in Figure 9, the BOOST pin current includes a
constant 7.0 mA pre−driver current and base current
proportional to switch conducting current. A detailed
discussion of this current is conducted in Thermal
Consideration section. A 0.1 mF capacitor is usually adequate
for maintaining the Boost pin voltage during the on time.
VIN
20k
±33%
1.2V
+
VZ = 6V to 8V
GND
Figure 10.
The IC enters a sleep mode when the SHDNB pin is pulled
below the Shutdown Threshold Voltage. In sleep mode, the
power switch is kept open and the supply current reduces to
Shutdown Quiescent Current ( 1 mA typically). This pin has
an internal pull−down current. When not in use, pull this pin
up to VCC with a resistor (See Figure 1). A 100 kW pullup
resistor will ensure safe operation from below 9 V and
during a 40 V load dump condition.
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NCP1546
Startup
minimum duty cycle by extending the switching cycle. This
protects the IC from overheating, and also limits the power
that can be transferred to the output. The current limit
foldback effectively reduces the current stress on the
inductor and diode. When the output is shorted, the DC
current of the inductor and diode can approach the current
limit threshold. Therefore, reducing the current limit by
40% can result in an equal percentage drop of the inductor
and diode current. The short circuit waveforms are captured
in Figure 12, and the benefit of the foldback frequency and
current limit is self−evident.
During power up, the regulator tends to quickly charge up
the output capacitors to reach voltage regulation. This gives
rise to an excessive in−rush current which can be detrimental
to the inductor, IC and catch diode. In V2 control , the
compensation capacitor provides Soft−Start with no need
for extra pin or circuitry. During the power up, the Output
Source Current of the error amplifier charges the
compensation capacitor which forces VC pin and thus output
voltage ramp up gradually. The Soft−Start duration can be
calculated by
V
CCOMP
TSS + C
ISOURCE
where:
VC = VC pin steady−state voltage, which is approximately
equal to error amplifier’s reference voltage.
CCOMP = Compensation capacitor connected to the VC pin
ISOURCE = Output Source Current of the error amplifier.
Using a 0.1 mF CCOMP, the calculation shows a TSS over
5.0 ms which is adequate to avoid any current stresses.
Figure 11 shows the gradual rise of the VC, VO and envelope
of the VSW during power up. There is no voltage over−shoot
after the output voltage reaches the regulation. If the supply
voltage rises slower than the VC pin, output voltage may
over−shoot.
Figure 12. In Short Circuit, the Foldback Current and
Foldback Frequency Limit the Switching Current to
Protect the IC, Inductor and Catch Diode
Thermal Considerations
A calculation of the power dissipation of the IC is always
necessary prior to the adoption of the regulator. The current
drawn by the IC includes quiescent current, pre−driver
current, and power switch base current. The quiescent
current drives the low power circuits in the IC, which
include comparators, error amplifier and other logic blocks.
Therefore, this current is independent of the switching
current and generates power equal to
WQ + VIN
IQ
where:
IQ = quiescent current.
Figure 11. The Power Up Transition of NCP1546
Regulator
The pre−driver current is used to turn on/off the power
switch and is approximately equal to 12 mA in worst case.
During steady state operation, the IC draws this current from
the Boost pin when the power switch is on and then receives
it from the VIN pin when the switch is off. The pre−driver
current always returns to the VSW pin. Since the pre−driver
current goes out to the regulator’s output even when the
power switch is turned off, a minimum load is required to
prevent overvoltage in light load conditions. If the Boost pin
voltage is equal to VIN + VO when the switch is on, the power
dissipation due to pre−driver current can be calculated by
Short Circuit
When the VFB pin voltage drops below Foldback
Threshold, the regulator reduces the peak current limit by
40% and switching frequency to 1/4 of the nominal
frequency. These features are designed to protect the IC and
external components during over load or short circuit
conditions. In those conditions, peak switching current is
clamped to the current limit threshold. The reduced
switching frequency significantly increases the ripple
current, and thus lowers the DC current. The short circuit can
cause the minimum duty cycle to be limited by Minimum
Output Pulse Width. The foldback frequency reduces the
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NCP1546
IRMS + IO ǸD(1 * D)
V 2
(VIN * VO ) O )
VIN
WDRV + 12 mA
where:
D = switching duty cycle which is equal to VO/VIN.
IO = load current.
The base current of a bipolar transistor is equal to collector
current divided by beta of the device. Beta of 60 is used here
to estimate the base current. The Boost pin provides the base
current when the transistor needs to be on. The power
dissipated by the IC due to this current is
V 2
WBASE + O
VIN
IS
60
where:
IS = DC switching current.
When the power switch turns on, the saturation voltage
and conduction current contribute to the power loss of a
non−ideal switch. The power loss can be quantified as
WSAT +
VO
VIN
IS
VSAT
where:
VSAT = saturation voltage of the power switch which is
shown in Figure 7.
Figure 13. Input Voltage Ripple in a Buck Converter
To calculate the RMS current, multiply the load current
with the constant given by Figure 14 at each duty cycle. It
is a common practice to select the input capacitor with an
RMS current rating more than half the maximum load
current. If multiple capacitors are paralleled, the RMS
current for each capacitor should be the total current divided
by the number of capacitors.
The switching loss occurs when the switch experiences
both high current and voltage during each switch transition.
This regulator has a 30 ns turn−off time and associated
power loss is equal to
I
WS + S
2
VIN
30 ns
fS
The turn−on time is much shorter and thus turn−on loss is
not considered here.
The total power dissipated by the IC is sum of all the above
0.6
0.5
WIC + WQ ) WDRV ) WBASE ) WSAT ) WS
The IC junction temperature can be calculated from the
ambient temperature, IC power dissipation and thermal
resistance of the package. The equation is shown as follows,
IRMS (XIO)
TJ + WIC
0.4
RqJA ) TA
0.3
0.2
Minimum Load Requirement
As pointed out in the previous section, a minimum load is
required for this regulator due to the pre−driver current
feeding the output. Placing a resistor equal to VO divided by
12 mA should prevent any voltage overshoot at light load
conditions. Alternatively, the feedback resistors can be
valued properly to consume 12 mA current.
0.1
0
0
0.2
0.4
0.6
DUTY CYCLE
0.8
1.0
Figure 14. Input Capacitor RMS Current can be
Calculated by Multiplying Y Value with Maximum Load
Current at any Duty Cycle
COMPONENT SELECTION
Selecting the capacitor type is determined by each
design’s constraint and emphasis. The aluminum
electrolytic capacitors are widely available at lowest cost.
Their ESR and ESL (equivalent series inductor) are
relatively high. Multiple capacitors are usually paralleled to
achieve lower ESR. In addition, electrolytic capacitors
usually need to be paralleled with a ceramic capacitor for
filtering high frequency noises. The OS−CON are solid
aluminum electrolytic capacitors, and therefore has a much
lower ESR. Recently, the price of the OS−CON capacitors
has dropped significantly so that it is now feasible to use
Input Capacitor
In a buck converter, the input capacitor witnesses pulsed
current with an amplitude equal to the load current. This
pulsed current and the ESR of the input capacitors determine
the VIN ripple voltage, which is shown in Figure 13. For VIN
ripple, low ESR is a critical requirement for the input
capacitor selection. The pulsed input current possesses a
significant AC component, which is absorbed by the input
capacitors. The RMS current of the input capacitor can be
calculated using:
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NCP1546
them for some low cost designs. Electrolytic capacitors are
physically large, and not used in applications where the size,
and especially height is the major concern.
Ceramic capacitors are now available in values over 10 mF.
Since the ceramic capacitor has low ESR and ESL, a single
ceramic capacitor can be adequate for both low frequency
and high frequency noises. The disadvantage of ceramic
capacitors are their high cost. Solid tantalum capacitors can
have low ESR and small size. However, the reliability of the
tantalum capacitor is always a concern in the application
where the capacitor may experience surge current.
The output ripple voltage is the sum of a triangular wave
caused by ripple current flowing through ESR, and a square
wave due to ESL. Capacitive reactance is assumed to be
small compared to ESR and ESL. The peak to peak ripple
current of the inductor is:
V (V * VO)
IP * P + O IN
(VIN)(L)(fS)
VRIPPLE(ESR), the output ripple due to the ESR, is equal
to the product of IP−P and ESR. The voltage developed
across the ESL is proportional to the di/dt of the output
capacitor. It is realized that the di/dt of the output capacitor
is the same as the di/dt of the inductor current. Therefore,
when the switch turns on, the di/dt is equal to (VIN − VO)/L,
and it becomes VO/L when the switch turns off. The total
ripple voltage induced by ESL can then be derived from
Output Capacitor
In a buck converter, the requirements on the output
capacitor are not as critical as those on the input capacitor.
The current to the output capacitor comes from the inductor
and thus is triangular. In most applications, this makes the
RMS ripple current not an issue in selecting output
capacitors.
V * VO
V
V
VRIPPLE(ESL) + ESL( IN) ) ESL( IN
) + ESL( IN)
L
L
L
The total output ripple is the sum of the VRIPPLE(ESR) and
VRIPPLE(ESR).
Figure 16. The Output Voltage Ripple Using One
100 mF POSCAP Capacitor
Figure 15. The Output Voltage Ripple Using Two 10 mF
Ceramic Capacitors in Parallel
Figure 17. The Output Voltage Ripple Using
One 100 mF OS−CON
Figure 18. The Output Voltage Ripple Using
One 100 mF Tantalum Capacitor
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NCP1546
Figure 15 to Figure 18 show the output ripple of a 5.0 V
to 3.3 V/500 mA regulator using 22 mH inductor and various
capacitor types. At the switching frequency, the low ESR
and ESL make the ceramic capacitors behave capacitively
as shown in Figure 15. Additional paralleled ceramic
capacitors will further reduce the ripple voltage, but
inevitably increase the cost. “POSCAP”, manufactured by
SANYO, is a solid electrolytic capacitor. The anode is
sintered tantalum and the cathode is a highly conductive
polymerized organic semiconductor. TPC series, featuring
low ESR and low profile, is used in the measurement of
Figure 16. It is shown that POSCAP presents a good balance
of capacitance and ESR, compared with a ceramic capacitor.
In this application, the low ESR generates less than 5.0 mV
of ripple and the ESL is almost unnoticeable. The ESL of the
through−hole OS−CON capacitor give rise to the inductive
impedance. It is evident from Figure 17 which shows the
step rise of the output ripple on the switch turn−on and large
spike on the switch turn−off. The ESL prevents the output
capacitor from quickly charging up the parasitic capacitor of
the inductor when the switch node is pulled below ground
through the catch diode conduction. This results in the spike
associated with the falling edge of the switch node. The D
package tantalum capacitor used in Figure 18 has the same
footprint as the POSCAP, but doubles the height. The ESR
of the tantalum capacitor is apparently higher than the
POSCAP. The electrolytic and tantalum capacitors provide
a low−cost solution with compromised performance. The
reliability of the tantalum capacitor is not a serious concern
for output filtering because the output capacitor is usually
free of surge current and voltage.
The worse case of the diode average current occurs during
maximum load current and maximum input voltage. For the
diode to survive the short circuit condition, the current rating
of the diode should be equal to the Foldback Current Limit.
See Table 1 for Schottky diodes from ON Semiconductor
which are suggested for use with the NCP1546 regulator.
Inductor Selection
When choosing inductors, one might have to consider
maximum load current, core and copper losses, component
height, output ripple, EMI, saturation and cost. Lower
inductor values are chosen to reduce the physical size of the
inductor. Higher value cuts down the ripple current, core
losses and allows more output current. For most
applications, the inductor value falls in the range between
2.2 mH and 22 mH. The saturation current ratings of the
inductor shall not exceed the IL(PK), calculated according to
V (V * VO)
IL(PK) + IO ) O IN
2(fS)(L)(VIN)
The DC current through the inductor is equal to the load
current. The worse case occurs during maximum load
current. Check the vendor’s spec to adjust the inductor value
under current loading. Inductors can lose over 50% of
inductance when it nears saturation.
The core materials have a significant effect on inductor
performance. The ferrite core has benefits of small physical
size, and very low power dissipation. But be careful not to
operate these inductors too far beyond their maximum
ratings for peak current, as this will saturate the core.
Powered Iron cores are low cost and have a more gradual
saturation curve. The cores with an open magnetic path,
such as rod or barrel, tend to generate high magnetic field
radiation. However, they are usually cheap and small. The
cores providing a close magnetic loop, such as pot−core and
toroid, generate low electro−magnetic interference (EMI).
There are many magnetic component vendors providing
standard product lines suitable for the NCP1546. Table 2
lists three vendors, their products and contact information.
Diode Selection
The diode in the buck converter provides the inductor
current path when the power switch turns off. The peak
reverse voltage is equal to the maximum input voltage. The
peak conducting current is clamped by the current limit of
the IC. The average current can be calculated from:
I (V * VO)
ID(AVG) + O IN
VIN
Table 1.
Part Number
VBREAKDOWN (V)
IAVERAGE (A)
V(F) (V) @ IAVERAGE
Package
1N5817
20
1.0
0.45
Axial Lead
1N5818
30
1.0
0.55
Axial Lead
1N5819
40
1.0
0.6
Axial Lead
MBR0520
20
0.5
0.385
SOD−123
MBR0530
30
0.5
0.43
SOD−123
MBR0540
40
0.5
0.53
SOD−123
MBRS120
20
1.0
0.55
SMB
MBRS130
30
1.0
0.395
SMB
MBRS140
40
1.0
0.6
SMB
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NCP1546
Table 2.
Vendor
Product Family
Web Site
Telephone
Coiltronics
UNI−Pac1/2: SMT, barrel
THIN−PAC: SMT, toroid, low profile
CTX: Leaded, toroid
www.coiltronics.com
(516) 241−7876
Coilcraft
DO1608: SMT, barrel
DS/DT 1608: SMT, barrel, magnetically shielded
DO3316: SMT, barrel
DS/DT 3316: SMT, barrel, magnetically shielded
DO3308: SMT, barrel, low profile
www.coilcraft.com
(800) 322−2645
Pulse
−
www.pulseeng.com
(619) 674−8100
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NCP1546
PACKAGE DIMENSIONS
DFN18 6x5, 0.5P
CASE 505−01
ISSUE D
NOTES:
1. DIMENSIONS AND TOLERANCING PER
ASME Y14.5M, 1994.
2. DIMENSIONS IN MILLIMETERS.
3. DIMENSION b APPLIES TO PLATED
TERMINAL AND IS MEASURED BETWEEN
0.25 AND 0.30 MM FROM TERMINAL
4. COPLANARITY APPLIES TO THE EXPOSED
PAD AS WELL AS THE TERMINALS.
A
D
B
PIN 1 LOCATION
E
2X
DIM
A
A1
A3
b
D
D2
E
E2
e
K
L
0.15 C
2X
TOP VIEW
0.15 C
(A3)
0.10 C
A
18X
0.08 C
A1
C
SIDE VIEW
SEATING
PLANE
D2
18X
L
MILLIMETERS
MIN
MAX
0.80
1.00
0.00
0.05
0.20 REF
0.18
0.30
6.00 BSC
3.98
4.28
5.00 BSC
2.98
3.28
0.50 BSC
0.20
−−−
0.45
0.65
SOLDERING FOOTPRINT*
e
1
5.30
9
1
18X
E2
K
18
10
BOTTOM VIEW
18X
18X
0.75
0.50
PITCH
b
0.10 C A B
0.05 C
4.19
NOTE 3
18X
0.30
3.24
DIMENSIONS: MILLIMETERS
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
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14
NCP1546
PACKAGE DIMENSIONS
SOIC−8 NB
CASE 751−07
ISSUE AK
−X−
NOTES:
1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
6. 751−01 THRU 751−06 ARE OBSOLETE. NEW
STANDARD IS 751−07.
A
8
5
S
B
0.25 (0.010)
M
Y
M
1
4
−Y−
K
G
C
N
DIM
A
B
C
D
G
H
J
K
M
N
S
X 45 _
SEATING
PLANE
−Z−
0.10 (0.004)
H
D
0.25 (0.010)
M
Z Y
S
X
M
J
S
MILLIMETERS
MIN
MAX
4.80
5.00
3.80
4.00
1.35
1.75
0.33
0.51
1.27 BSC
0.10
0.25
0.19
0.25
0.40
1.27
0_
8_
0.25
0.50
5.80
6.20
INCHES
MIN
MAX
0.189
0.197
0.150
0.157
0.053
0.069
0.013
0.020
0.050 BSC
0.004
0.010
0.007
0.010
0.016
0.050
0 _
8 _
0.010
0.020
0.228
0.244
SOLDERING FOOTPRINT*
1.52
0.060
7.0
0.275
4.0
0.155
0.6
0.024
1.270
0.050
SCALE 6:1
mm Ǔ
ǒinches
*For additional information on our Pb−Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
V2 is a trademark of Switch Power, Inc.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications
intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should
Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates,
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associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal
Opportunity/Affirmative Action Employer. This literature is subject to all applicable copyright laws and is not for resale in any manner.
PUBLICATION ORDERING INFORMATION
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Phone: 303−675−2175 or 800−344−3860 Toll Free USA/Canada
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For additional information, please contact your local
Sales Representative
NCP1546/D
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