LT3798 Isolated No Opto-Coupler Flyback Controller with Active PFC DESCRIPTION FEATURES Isolated PFC Flyback with Minimum Number of External Components n V and V IN OUT Limited Only by External Components n Active Power Factor Correction n Low Harmonic Distortion n No Opto-Coupler Required n Constant-Current and Constant-Voltage Regulation n Accurate Regulated Voltage and Current (±5% Typical) n Energy Star Compliant (<0.5W No Load Operation) n Thermally Enhanced 16-lead MSOP Package The LT®3798 is a constant-voltage/constant-current isolated flyback controller that combines active power factor correction (PFC) with no opto-coupler required for output voltage feedback into a single-stage converter. A LT3798 based design can achieve a power factor of greater than 0.97 by actively modulating the input current, allowing compliance with most Harmonic Current Emission requirements. n The LT3798 is well suited for a wide variety of off-line applications. The input range can be scaled up or down, depending mainly on the choice of external components. Efficiencies higher than 86% can be achieved with output power levels up to 100W. In addition, the LT3798 can easily be designed into high DC input applications. APPLICATIONS Offline 5W to 100W+ Applications High DC VIN Isolated Applications n Offline Bus Converter (12V, 24V or 48V Outputs) n n L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499 and 7471522. TYPICAL APPLICATION Universal Input 24W PFC Bus Converter VOUT vs IOUT 24.50 90V TO 265V AC 0.1µF 499k 100k 499k 100k D2 20Ω 4:1:1 4.7pF 10µF 2k D3 VIN VIN_SENSE 1M DCM Z1 221k 40.2k 16.5k CTRL3 GATE CTRL2 SENSE CTRL1 OVP INTVCC GND VC 2.2µF 23.50 VAC = 90V VAC = 120V VAC = 220V VAC = 265V 0 0.2 0.4 0.6 IOUT (A) 0.8 1 3798 TA01b 20Ω Z2 0.05Ω 4.7µF COMP+ 24V 1A 560µF ×2 22pF D1 LT3798 100k D4 90.9k 4.99k VREF 24.00 23.75 FB EN/UVLO 95.3k VOUT (V) 24.25 2.2nF COMP– 0.1µF 3798 TA01a 3798fa 1 LT3798 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) EN/UVLO....................................................................30V VIN.............................................................................42V INTVCC.......................................................................12V CTRL1, CTRL2, CTRL3.................................................4V FB, VREF, COMP +.........................................................3V VC, OVP, COMP–..........................................................4V SENSE.......................................................................0.4V VIN_SENSE..................................................................1mA DCM........................................................................±3mA Operating Temperature Range (Note 2) LT3798E/LT3798I................................... –40°C to 125°C Storage Temperature Range................... –65°C to 150°C TOP VIEW CTRL1 CTRL2 CTRL3 VREF OVP VC COMP+ COMP– 1 2 3 4 5 6 7 8 16 15 14 13 12 11 10 9 17 GND VIN_SENSE SENSE GATE INTVCC EN/UVLO VIN DCM FB MSE PACKAGE 16-LEAD PLASTIC MSOP θJA = 50°C/W, θJC = 10°C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LT3798EMSE#PBF LT3798EMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 125°C LT3798IMSE#PBF LT3798IMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 125°C LT3798HMSE#PBF LT3798HMSE#TRPBF 3798 16-Lead Plastic MSOP –40°C to 150°C LT3798MPMSE#PBF LT3798MPMSE#TRPBF 3798 16-Lead Plastic MSOP –55°C to 150°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted. PARAMETER CONDITIONS MIN Input Voltage Range VIN Quiescent Current TYP 10 VEN/UVLO = 0.2V VEN/UVLO = 1.5V, Not Switching 45 60 70 MAX UNITS 38 V 70 µA µA VIN Quiescent Current, INTVCC Overdriven VINTVCC = 11V 60 µA VIN Shunt Regulator Voltage I = 1mA 40 V 8 mA VIN Shunt Regulator Current Limit INTVCC Quiescent Current VEN/UVLO = 0.2V VEN/UVLO = 1.5V, Not Switching EN/UVLO Pin Threshold EN/UVLO Pin Voltage Rising EN/UVLO Pin Hysteresis Current EN/UVLO=1V VIN_SENSE Threshold Turn Off VREF Voltage 0µA Load 200µA Load CTRL1/CTRL2/CTRL3 Pin Bias Current CTLR1/CTRL2/CTRL3 = 1V l 12.5 1.8 15.5 2.2 17.5 2.7 1.21 1.25 1.29 V 8 10 12 µA 1.97 1.95 2.0 1.98 27 l l µA mA µA 2.03 2.03 V V ±30 nA 3798fa 2 LT3798 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 18V, INTVCC = 11V, unless otherwise noted. PARAMETER SENSE Current Limit Threshold CONDITIONS VIN_SENSE = 150µA MIN TYP MAX UNITS 96 102 107 mV Minimum SENSE Current Limit Threshold VIN_SENSE = 34µA 14 mV Minimum SENSE Current Limit Threshold VIN_SENSE = 21µA 4 mV SENSE Input Bias Current Current Out of Pin, SENSE = 0V 15 µA FB Voltage l 1.22 1.25 1.28 V 0.01 0.03 %/V 4.25 4.4 FB voltage Line Regulation 10V < VIN < 35V FB Pin Bias Current (Note 3), FB = 1V FB Error Amplifier Voltage Gain ΔVVC/ΔVFB, CTRL1=1V, CTRL2=2V, CTRL3=2V 180 V/V FB Error Amplifier Transconductance ΔI = 5µA 170 UMHOS 4.05 µA Current Error Amplifier Voltage Gain ∆VCOMP+/∆VCOMP–, CTRL1 = 1V, CTRL2 = 2V, CTRL3 = 2V 100 V/V Current Error Amplifier Transconductance ∆I = 5µA 50 UMHOS Current Loop Voltage Gain ΔVCTRL/ΔVSENSE,1000pF Cap from COMP+ to COMP– 21 V/V DCM Current Turn-On Threshold Current Out of Pin 80 µA Maximum Oscillator Frequency COMP+ = 0.95V, VIN_SENSE = 150µA 150 kHz Minimum Oscillator Frequency COMP+ = 0V, VFB <VOVP 4 kHz Minimum Oscillator Frequency COMP+ = 0V, VFB >VOVP 0.5 kHz 20 kHz Backup Oscillator Frequency Linear Regulator INTVCC Regulation Voltage No Load 9.8 10 10.4 V 500 900 mV Dropout (VIN-INTVCC) IINTVCC = –10mA, VIN = 10V Current Limit Below Undervoltage Threshold 12 25 mA Current Limit Above Undervoltage Threshold 80 120 mA Gate Driver tr GATE Driver Output Rise Time CL = 3300pF, 10% to 90% 18 ns tf GATE Driver Output Fall Time CL = 3300pF, 90% to 10% 18 ns GATE Output Low (VOL) GATE Output High (VOH) Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3798E is guaranteed to meet specified performance from 0°C to 125°C junction temperature. Specification over the –40°C and 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The 0.01 INTVCC50mV V V LT3798I is guaranteed to meet specified performance from –40°C to 125°C operating junction temperature range. The LT3798H is guaranteed to meet performance specifications over the –40°C to 150°C operating junction temperature range. The LT3798MP is guaranteed to meet performance specifications over the –55°C to 150°C operating junction temperature range. High junction temperatures degrade operating lifetimes. Operating lifetime is derated for junction temperatures greater than 125°C. Note 3: Current flows out of the FB pin. 3798fa 3 LT3798 TYPICAL PERFORMANCE CHARACTERISTICS EN/UVLO Threshold vs Temperature EN/UVLO Hysteresis Current vs Temperature RISING 1.26 1.24 FALLING 1.22 1.2 –50 –25 0 11.5 11 60 50 40 30 10.5 20 10 0 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 0 VREF vs Temperature 2.05 SENSE Current Limit Threshold vs Temperature VREF vs VIN 120 2.04 CURRENT LIMIT (mA) VIN = 24V WITH NO LOAD 2.000 1.975 VIN = 24V WITH 200µA LOAD VREF (V) 2.02 2.025 2.01 NO LOAD 2 1.99 200µA LOAD 1.98 1.950 1.97 1.925 1.96 1.900 –50 –25 1.95 0 25 50 75 100 125 150 TEMPERATURE (°C) 10 20 15 25 VIN (V) 30 35 40 Backup Oscillator Frequency vs Temperature 25 3798 G06a 20 VFB < VOVP 2 0 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) 3798 G06 FREQUENCY (kHz) FREQUENCY (kHz) 25 50 75 100 125 150 TEMPERATURE (°C) 0 3798 G05 3 1 0 MIN ILIM VIN_SENSE = 34µA MIN ILIM VIN_SENSE = 21µA 4 195 120 –50 –25 40 0 –50 –25 5 145 60 Minimum Oscillator Frequency vs Temperature 220 170 80 20 3798 G05 Maximum Oscillator Frequency vs Temperature MAX ILIM 100 2.03 2.050 25 50 75 100 125 150 TEMPERATURE (°C) 3798 G03 3798 G02 2.075 VREF (V) VIN = 12V 70 3798 G01 FREQUENCY (kHz) VIN = 24V 80 10 –50 –25 25 50 75 100 125 150 TEMPERATURE (°C) VIN IQ vs Temperature 90 IQ (µA) 1.28 EN/UVLO (V) 100 12 EN/UVLO HYSTERESIS CURRENT (µA) 1.3 2.100 TA = 25°C, unless otherwise noted. 10 5 VOVP > VFB 0 15 25 50 75 100 125 150 TEMPERATURE (°C) 3799 G07 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) 3798 G07a 3798fa 4 LT3798 TYPICAL PERFORMANCE CHARACTERISTICS 10.5 INTVCC vs Temperature 10.2 NO LOAD 10mA LOAD 25mA LOAD TA = 25°C, unless otherwise noted. INTVCC vs VIN 42 10 VIN SHUNT VOLTAGE (V) INTVCC (V) INTVCC (V) 9.8 9.6 9.4 9.2 0 9 25 50 75 100 125 150 TEMPERATURE (°C) 5 10 25 20 VIN (V) 30 35 9 8 7 6 25 50 75 100 125 150 TEMPERATURE (°C) 3798 G11 25 1.8 1.6 VOUT vs Temperature PAGE 17 SCHEMATIC: UNIVERSAL 24.5 1.4 1.2 1 0.8 VAC = 120V 24 VAC = 220V 0.6 23.5 0.4 0.2 0 0 120 60 40 80 100 20 SENSE CURRENT LIMIT THRESHOLD (mV) 23 –50 –25 3798 G12 0 25 50 75 100 125 150 TEMPERATURE (°C) 3798 G012a Output Current vs Input Voltage 1.10 PAGE 17 SCHEMATIC: UNIVERSAL OUTPUT CURRENT (A) VOUT (V) 25 50 75 100 125 150 TEMPERATURE (°C) 3798 G10 2 24.4 24.2 24 23.8 23.6 0 3798 G09 Output Voltage vs Input Voltage 24.6 39 –50 –25 40 VOUT (V) LEAKAGE INDUCTANCE BLANKING TIME (µs) 10 SHUNT CURRENT (mA) 15 Leakage Inductance Blanking Time vs SENSE Current Limit Threshold Maximum Shunt Current vs Temperature 0 40 39.5 3798 G08 5 –50 –25 41 40.5 9.75 9.5 –50 –25 ISHUNT = 1mA 41.5 10.25 10 VIN Shunt Voltage vs Temperature 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3798 G13 1.05 PAGE 17 SCHEMATIC: UNIVERSAL VOUT = 22V 1.00 0.95 0.90 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3798 G14 3798fa 5 LT3798 TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, unless otherwise noted. Power Factor vs Input Voltage Efficiency vs Input Voltage 100 1.00 0.99 90 0.97 EFFICIENCY (%) POWER FACTOR 0.98 PAGE 17 SCHEMATIC: UNIVERSAL 0.96 0.95 0.94 0.93 80 70 0.92 0.91 PAGE 17 SCHEMATIC: UNIVERSAL 0.90 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3798 G15 60 90 110 130 150 170 190 210 230 250 270 VIN (VAC) 3798 G16 PIN FUNCTIONS CTRL1, CTRL2, CTRL3 (Pin 1, Pin 2, Pin 3): Current Output Adjustment Pins. These pins control the output current. The lowest value out of the three CTRL inputs is compared to negative input of the operational amplifier. VREF (Pin 4): Voltage Reference Output Pin. Typically 2V. This pin drives a resistor divider for the CTRL pin, either for analog dimming or for temperature limit/compensation of output load. Can supply up to 200µA. OVP (Pin 5): Overvoltage Protection. This pin accepts a DC voltage to compare to the sample and hold’s voltage output information. When output voltage information is above the OVP, the part divides the minimum switching frequency by 8, around 500Hz. This protects devices connected to the output. This also allows the part to operate with very little power consumption with no load to meet energy star requirements. VC (Pin 6): Compensation Pin for Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator. A 100pF capacitor in parallel helps eliminate noise. COMP+, COMP– (Pin 7, Pin 8): Compensation Pins for Internal Error Amplifier. Connect a capacitor between these two pins to compensate the internal feedback loop. FB (Pin 9): Voltage Loop Feedback Pin. FB is used to regulate the output voltage by sampling the third winding. If the converter is used in current mode, the FB pin will normally be at a voltage level lower than 1.25V, and will reach the steady state of 1.25V if it detects an open output condition. DCM (Pin 10): Discontinuous Conduction Mode Detection Pin. Connect a capacitor and resistor in series with this pin to the third winding. VIN (Pin 11): Input Voltage. This pin supplies current to the internal start-up circuitry and to the INTVCC LDO. This pin must be locally bypassed with a capacitor. A 42V shunt regulator is internally connected to this pin. EN/UVLO (Pin 12): Enable/Undervoltage Lockout. A resistor divider connected to VIN is tied to this pin to program the minimum input voltage at which the LT3798 will turn on. When below 1.25V, the part will draw 60µA with most of the internal circuitry disabled and a 10µA hysteresis current will be pulled out of the EN/UVLO pin. When above 1.25V, the part will be enabled and begin to switch and the 10µA hysteresis current is turned off. INTVCC (Pin 13): Regulated Supply for Internal Loads and GATE Driver. Supplied from VIN and regulates to 10V (typical). INTVCC must be bypassed with a 4.7µF capacitor placed close to the pin. 3798fa 6 LT3798 PIN FUNCTIONS GATE (Pin 14): N-Channel FET Gate Driver Output. Switches between INTVCC and GND. Driven to GND during shutdown state and stays high during low voltage states. VIN_SENSE (Pin 16): Line Voltage Sense Pin. The pin is used for sensing the AC line voltage to perform power factor correction. Connect a resistor in series with the line voltage to this pin. If no PFC is needed, connect this pin to INTVCC with a 25k resistor. SENSE (Pin 15): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor, RSENSE, in the source of the NFET. The negative terminal of the current sense resistor should be connected to the GND plane close to the IC. GND (Exposed Pad Pin 17): Ground. The exposed pad of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. The exposed pad must be soldered to the circuit board for proper operation. BLOCK DIAGRAM D2 • R4 R1 R13 L1C C1 R5 R15 EN/UVLO + A2 + – – VIN_SENSE R9 COMP+ FB ONE SHOT COMP– – A7 + 600mV A9 A3 – A5 + CTRL2 S R C5 S GATE DRIVER SENSE A4 A6 MULTIPLIER VOUT – Q MASTER LATCH MINIMUM C7 R11 + A1 – SW1 CTRL1 VOUT + INTVCC R7 CURRENT COMPARATOR R12 1M C6 D1 L1B N:1 VIN 1.22V S&H OVP T1 L1A R14 STARTUP INTERNAL REG VREF C2 C3 DCM R8 VRECTIFIED R3 GND LOW OUTPUT CURRENT OSCILLATOR R6 CTRL3 S&H FB VC R10 1.22V + A8 – 3798 BD C4 3798fa 7 LT3798 OPERATION The LT3798 is a current mode switching controller IC designed specifically for generating a constant current/ constant voltage supply in an isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage and current on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with an opto-isolator. The LT3798 uses a novel method of using the external MOSFETs peak current information from the sense resistor to calculate the output current of a flyback converter without the need of an opto-coupler. Active power factor correction is becoming a requirement for offline power supplies and the power levels are decreasing. A power factor of one is achieved if the current drawn is proportional to the input voltage. The LT3798 modulates the peak current limit with a scaled version of the input voltage. This technique can provide power factors of 0.97 or greater. The Block Diagram shows an overall view of the system. The external components are in a flyback topology configuration. The third winding senses the output voltage and also supplies power to the part in steady-state operation. The VIN pin supplies power to an internal LDO that generates 10V at the INTVCC pin. The novel control circuitry consists of two error amplifiers, a minimum circuit, a multiplier, a transmission gate, a current comparator, a low output current oscillator and a master latch, which will be explained in the following sections. The part also features a sample-and-hold to sample the output voltage from the third winding. A comparator is used to detect discontinuous conduction mode (DCM) with a cap connected to the third winding. The part features a 1.9A gate driver. The LT3798 is designed for both off-line and DC applications. The EN/UVLO and a resistor divider can be configured for a micropower hysteretic start-up. In the Block Diagram, R3 is used to stand off the high voltage supply voltage. The internal LDO starts to supply current to the INTVCC when VIN is above 2.5V. The VIN and INTVCC capacitors are charged by the current from R3. When VIN exceeds the turn-on threshold and INTVCC is in regulation at 10V, the part begins to switch. The VIN hysteresis is set by the EN/ UVLO resistor divider. The third winding provides power to VIN when its voltage is higher than the VIN voltage. A voltage shunt is provided for fault protection and can sink 8mA of current when VIN is over 40V. During a typical cycle, the gate driver turns the external MOSFET on and a current flows through the primary winding. This current increases at a rate proportional to the input voltage and inversely proportional to the magnetizing inductance of the transformer. The control loop determines the maximum current and the current comparator turns the switch off when the current level is reached. When the switch turns off, the energy in the core of the transformer flows out the secondary winding through the output diode, D1. This current decreases at a rate proportional to the output voltage. When the current decreases to zero, the output diode turns off and voltage across the secondary winding starts to oscillate from the parasitic capacitance and the magnetizing inductance of the transformer. Since all windings have the same voltage across them, the third winding rings too. The capacitor connected to the DCM pin, C1, trips the comparator A2, which serves as a dv/dt detector, when the ringing occurs. This timing information is used to calculate the output current and will be described below. The dv/dt detector waits for the ringing waveform to reach its minimum value and then the switch turns back on. This switching behavior is similar to zero volt switching and minimizes the amount of energy lost when the switch is turned back on and improves efficiency as much as 5%. Since this part operates on the edge of continuous conduction mode and discontinuous conduction mode, the operating mode is called critical conduction mode (or boundary conduction mode). Primary Side Control Loops The LT3798 achieves constant current/constant voltage operation by using two separate error amplifiers. These two amplifiers are then fed to a circuit that outputs the lower voltage of the two, shown as the "minimum" block in the Block Diagram. This voltage is converted to a current before being fed into the multiplier. 3798fa 8 LT3798 OPERATION Primary Side Current Control Loop The CTRL1/CTRL2/CTRL3 pins control the output current of the flyback controller. To simplify the loop, let’s assume the VIN_SENSE pin is held at a constant voltage above 1V eliminating the multiplier from the control loop. The error amplifier, A5, is configured as integrator with the external capacitor C6. The COMP+ node voltage is converted to a current into the multiplier with the V/I converter, A6. Since A7’s output is constant, the output of the multiplier is proportional to A6 and can be ignored. The output of the multiplier controls the peak current with its connection to the current comparator, A1. The output of the multiplier is also connected to the transmission gate, SW1, and to a 1M resistor. The transmission gate, SW1, turns on when the secondary current flows to the output capacitor. This is called the flyback period when the output diode D1 is on. The current through the 1M resistor gets integrated by A5. The lowest CTRL input is equal to the negative input of A5 in steady state. A current output regulator normally uses a sense resistor in series with the output current and uses a feedback loop to control the peak current of the switching converter. In this isolated case, the output current information is not available so instead the LT3798 calculates it using the information available on the primary side of the transformer. The output current may be calculated by taking the average of the output diode current. As shown in Figure 1, the diode current is a triangle waveform with a base of the flyback time and a height of the peak secondary winding current. In a flyback topology, the secondary winding current is N times the primary winding current, where NPS is the primary to secondary winding ratio. Instead of taking the area of the triangle, let’s think of it as a pulse width modulation (PWM) waveform. During the flyback time, the average current is half the peak secondary winding current and zero during the rest of the cycle. The equation to express the output current is: IOUT = 0.5 • IPK • NPS • D where D is equal to the percentage of the cycle that the flyback time represents. The LT3798 has access to the primary winding current, the input to the current comparator, and when the flyback time starts and ends. Now the output current can be calculated by averaging a PWM SECONDARY DIODE CURRENT IPK(sec) SWITCH WAVEFORM TFLYBACK TPERIOD 3798 F01 Figure 1. Secondary Diode Current and Switch Waveforms waveform with a height of the current limit and a duty cycle of the flyback time over the entire cycle. In the feedback loop described above, the input to the integrator is such a waveform. The integrator adjusts the peak current until calculated output current equals the control voltage. If the calculated output current is low compared to the control pin, the error amplifier increases the voltage on the COMP+ node thus increasing the current comparator input. Primary Side Voltage Control The output voltage is available through the third winding on the primary side. A resistor divider attenuates the output voltage for the voltage error amplifier. A sample-and-hold circuit samples the attenuated output voltage and feeds it to the error amplifier. The output of the error amplifier is the V pin. This node needs a capacitor to compensate the output voltage control loop. Power Factor Correction When the VIN_SENSE voltage is connected to a resistor divider of the supply voltage, the current limit is proportional to the supply voltage. The minimum of the two error amplifier outputs is multiplied with the VIN_SENSE pin voltage. If the LT3798 is configured with a fast control loop, slower changes from the VIN_SENSE pin would not interfere with the current limit or the output current. The COMP+ pin would adjust to the changes of the VIN_SENSE. The only way for the multiplier to function is to set the control loop to be an order of magnitude slower than the fundamental frequency of the VIN_SENSE signal. In an offline case, the 3798fa 9 LT3798 OPERATION fundamental frequency of the supply voltage is 120Hz so the control loop unity gain frequency needs to be set less than approximately 12Hz. Without a large amount of energy storage on the secondary side, the output current will be affected by the supply voltage changes, but the DC component of the output current will be accurate. For DC input or non-PFC AC input applications, connect a 25k resistor from VIN_SENSE to INTVCC instead of the AC line voltage. VIN R1 EN/UVLO R2 LT3798 GND 3798 F02 Figure 2. Undervoltage Lockout (UVLO) Startup The LT3798 uses a hysteretic start-up to operate from high offline voltages. A resistor connected to the supply voltage protects the part from high voltages. This resistor is connected to the VIN pin on the part and bypassed with a capacitor. When the resistor charges the VIN pin to a turn-on voltage set with the EN/UVLO resistor divider and the INTVCC pin is at its regulation point, the part begins to switch. The resistor cannot provide power for the part in steady state, but relies on the capacitor to start-up the part, then the third winding begins to provide power to the VIN pin along with the resistor. An internal voltage clamp is attached to the VIN pin to prevent the resistor current from allowing VIN to go above the absolute maximum voltage of the pin. The internal clamp is set at 40V and is capable of 8mA(typical) of current at room temperature. Programming Output Voltage Setting the VIN Turn-On and Turn-Off Voltages The temperature coefficient of the diode's forward drop needs to be the opposite of the term, (R4 • ITC)/NST. By taking the partial derivative with respect to temperature, the value of R4 is found to be the following: A large voltage difference between the VIN turn-on voltage and the VIN turn-off voltage is preferred to allow time for the third winding to power the part. The EN/UVLO sets these two voltages. The pin has a 10µA current sink when the pins voltage is below 1.25V and 0µA when above 1.25V. The VIN pin connects to a resistor divider as shown in Figure 2. The UVLO threshold for VIN rising is: 1.25V • (R1+ R2) + 10µA • R1 R2 The UVLO Threshold for VIN Falling is : VIN(UVLO,RISING) = VIN(UVLO,FALLING) 1.25V • (R1+ R2) = R2 The output voltage is set using a resistor divider from the third winding to the FB pin. From the Block Diagram, the resistors R4 and R5 form a resistor divider from the third winding. The FB also has an internal current source that compensates for the diode drop. This current source causes an offset in the output voltage that needs to be accounted for when setting the output voltage. The output voltage equation is: VOUT = VBG (R4+R5)/(NST • R5)–(VF + (R4 • ITC)/NST) where VBG is the internal reference voltage, NST is the winding ratio between the secondary winding and the third winding, VF is the forward drop of the output rectifying diode, and ITC is the internal current source for the FB pin. R4 = NST(1/(δITC /δT)(δVF /δT)) δITC /δT = 12.4nA/°C ITC = 4.25µA where δITC /δT is the partial derivative of the ITC current source, and δVF /δT is the partial derivative of the forward drop of the output rectifying diode. With R4 set with the above equation, the resistor value for R5 is found using the following: R5 = (VBG • R4)/(NST(VOUT+VF)+R4 • ITC-VBG) 3798fa 10 LT3798 OPERATION Programming Output Current The maximum output current depends on the supply voltage and the output voltage in a flyback topology. With the VIN_SENSE pin connected to 100µA current source and a DC supply voltage, the maximum output current is determined at the minimum supply voltage, and the maximum output voltage using the following equation: NPS IOUT(MAX) = 2•(1– D)• 42 • R SENSE where VOUT • NPS D= VOUT • NPS + VIN The maximum control voltage to achieve this maximum output current is 2V • (1-D). It is suggested to operate at 95% of these values to give margin for the part’s tolerances. When designing for power factor correction, the output current waveform is going to have a half sine wave squared shape and will no longer be able to provide the above currents. By taking the integral of a sine wave squared over half a cycle, the average output current is found to be half the value of the peak output current. In this case, the recommended maximum average output current is as follows: NPS IOUT(MAX) = 2•(1−D) • • 47.5% 42 • R SENSE where D= VOUT • NPS VOUT • NPS + VIN The maximum control voltage to achieve this maximum output current is (1-D) • 47.5%. For control voltages below the maximum, the output current is equal to the following equation: NPS IOUT = CTRL • 42 • R SENSE The VREF pin supplies a 2V reference voltage to be used with the control pins. To set an output current, a resistor divider is used from the 2V reference to one of the control pins. The following equation sets the output current with a resistor divider: 2NPS R1= R2 – 1 42 •IOUT • RSENSE where R1 is the resistor connected to the VREF pin and the CTRL pin and R2 is the resistor connected to the CTRL pin and ground. Setting VIN_SENSE Resistor The VIN_SENSE resistor sets the current feeding the internal multiplier that modulates the current limit for power factor correction. At the maximum line voltage, VMAX, the current is set to 360µA. Under this condition, the resistor value is equal to (VMAX/360µA). For DC input or non-PFC AC input applications, connect a 25k resistor from VIN_SENSE to INTVCC instead of the AC line voltage. Critical Conduction Mode Operation Critical conduction mode is a variable frequency switching scheme that always returns the secondary current to zero with every cycle. The LT3798 relies on boundary mode and discontinuous mode to calculate the critical current because the sensing scheme assumes the secondary current returns to zero with every cycle. The DCM pin uses a fast current input comparator in combination with a small capacitor to detect dv/dt on the third winding. To eliminate false tripping due to leakage inductance ringing, a blanking time of between 600ns and 2μs is applied after the switch turns off, depending on the current limit shown in the Leakage Inductance Blanking Time vs SENSE Current Limit Threshold curve in the Typical Performance Characteristics section. The detector looks for 80μA of current through the DCM pin due to falling voltage on the third winding when the secondary diode turns off. This detection is important since the output current is calculated using this comparator’s output. This is not the optimal time to turn the switch on because the switch voltage is still close to VIN + VOUT • NPS and would waste all the energy stored in the parasitic capacitance on the switch node. Discontinuous ringing begins when the secondary current reaches zero and the energy in the parasitic capacitance on the switch node transfers 3798fa 11 LT3798 OPERATION to the input capacitor. This is a second-order network composed of the parasitic capacitance on the switch node and the magnetizing inductance of the primary winding of the transformer. The minimum voltage of the switch node during this discontinuous ring is VIN – VOUT • NPS. The LT3798 turns the switch back on at this time, during the discontinuous switch waveform, by sensing when the slope of the switch waveform goes from negative to positive using the dv/dt detector. This switching technique may increase efficiency by 5%. Sense Resistor Selection The resistor, RSENSE, between the source of the external N-channel MOSFET and GND should be selected to provide an adequate switch current to drive the application without exceeding the current limit threshold. For applications without power factor correction, select a resistor according to: RSENSE = 2(1– D)NPS • 95% IOUT • 42 where D= VOUT • NPS VOUT • NPS + VIN For applications with power factor correction, select a resistor according to: RSENSE = 2(1– D)NPS • 47.5% IOUT • 42 where D= VOUT • NPS VOUT • NPS + VIN Minimum Current Limit The LT3798 features a minimum current limit of approximately 18% of the peak current limit. This is necessary when operating in critical conduction mode since low current limits would increase the operating frequency to a very high frequency. The output voltage sensing circuitry needs a minimum amount of flyback waveform time to sense the output voltage on the third winding. The time needed is 350ns. The minimum current limit allows the use of smaller transformers since the magnetizing primary inductance does not need to be as high to allow proper time to sample the output voltage information. To help improve crossover distortion of the line input current, a second minimum current limit of 6% becomes active when the VIN_SENSE current is lower than 27µA. Since the off-time becomes very short with this lower minimum current limit, the sample-and-hold is deactivated. Universal Input The LT3798 operates over the universal input voltage range of 90VAC to 265VAC. In the Typical Performance Characteristics section, the Output Voltage vs VIN and the Output Current vs VIN graphs, show the output voltage and output current line regulation for the first application picture in the Typical Applications section. Selecting Winding Turns Ratio Boundary mode operation gives a lot of freedom in selecting the turns ratio of the transformer. We suggest to keep the duty cycle low, lower NPS, at the maximum input voltage since the duty cycle will increase when the AC waveform decreases to zero volts. A higher NPS increases the output current while keeping the primary current limit constant. Although this seems to be a good idea, it comes at the expense of a higher RMS current for the secondary-side diode which might not be desirable because of the primary side MOSFET’s superior performance as a switch. A higher NPS does reduce the voltage stress on the secondary-side diode while increasing the voltage stress on the primaryside MOSFET. If switching frequency at full output load is kept constant, the amount of energy delivered per cycle by the transformer also stays constant regardless of the NPS. Therefore, the size of the transformer remains the same at practical NPS’s. Adjusting the turns ratio is a good way to find an optimal MOSFET and diode for a given application. 3798fa 12 LT3798 OPERATION Switch Voltage Clamp Requirement Leakage inductance of an offline transformer is high due to the extra isolation requirement. The leakage inductance energy is not coupled to the secondary but goes into the drain node of the MOSFET. This is problematic since 400V and higher rated MOSFETs cannot always handle this energy by avalanching. Therefore the MOSFET needs protection. A transient voltage suppressor (TVS) and diode are recommended for all offline application and connected, as shown in Figure 3. The TVS device needs a reverse breakdown voltage greater than (VOUT + VF) • NPS where VOUT is the output voltage of the flyback converter, VF is the secondary diode forward voltage, and NPS is the turns ratio. An RCD clamp can be used in place of the TVS clamp. VSUPPLY VSUPPLY period, as well. Similarly, initial values can be estimated using stated switch capacitance and transformer leakage inductance. Once the value of the drain node capacitance and inductance is known, a series resistor can be added to the snubber capacitance to dissipate power and critically dampen the ringing. The equation for deriving the optimal series resistance using the observed periods (tPERIOD, and tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is below, and the resultant waveforms are shown in Figure 4. CPAR = LPAR = CSNUBBER tPERIOD(SNUBBED) 2 –1 tPERIOD tPERIOD2 CPAR • 4π2 RSNUBBER = GATE LPAR CPAR GATE 90 80 70 3798 F03 In addition to clamping the spike, in some designs where short circuit protection is desired, it will be necessary to decrease the amount of ringing by using an RC snubber. Leakage inductance ringing is at its worst during a short circuit condition, and can keep the converter from cycling on and off by peak charging the bias capacitor. On/off cycling is desired to keep power dissipation down in the output diode. Alternatively, a heat sink can be used to manage diode temperature. The recommended approach for designing an RC snubber is to measure the period of the ringing at the MOSFET drain when the MOSFET turns off without the snubber and then add capacitance—starting with something in the range of 100pF—until the period of the ringing is 1.5 to 2 times longer. The change in period will determine the value of the parasitic capacitance, from which the parasitic inductance can be determined from the initial VDRAIN (V) Figure 3. TVS & RCD Switch Voltage Clamps 60 50 40 30 NO SNUBBER WITH SNUBBER CAPACITOR WITH RESISTOR AND CAPACITOR 20 10 0 0 0.05 0.10 0.15 0.20 TIME (µs) 0.25 0.30 3798 F04 Figure 4. Observed Waveforms at MOSFET Drain when Iteratively Implementing an RC Snubber Note that energy absorbed by a snubber will be converted to heat and will not be delivered to the load. In high voltage or high current applications, the snubber may need to be sized for thermal dissipation. To determine the power dissipated in the snubber resistor from capacitive losses, measure the drain voltage immediately before the MOSFET turns on and use the following equation relating that 3798fa 13 LT3798 OPERATION voltage and the MOSFET switching frequency to determine the expected power dissipation: PSNUBBER = fSW • CSNUBBER • VDRAIN2/2 Decreasing the value of the capacitor will reduce the dissipated power in the snubber at the expense of increased peak voltage on the MOSFET drain, while increasing the value of the capacitance will decrease the overshoot. Transformer Design Considerations Transformer specification and design is a critical part of successfully applying the LT3798. In addition to the usual list of caveats dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. Since the current on the secondary side of the transformer is inferred by the current sampled on the primary, the transformer turns ratio must be tightly controlled to ensure a consistent output current. A tolerance of ±5% in turns ratio from transformer to transformer could result in a variation of more than ±5% in output regulation. Fortunately, most magnetic component manufacturers are capable of guaranteeing a turns ratio tolerance of 1% or better. Linear Technology has worked with several leading magnetic component manufacturers to produce predesigned flyback transformers for use with the LT3798. Table 1 shows the details of several of these transformers. Loop Compensation The voltage feedback loop is a traditional GM error amplifier. The loop cross-over frequency is set much lower than twice the line frequency for PFC to work properly. The current output feedback loop is an integrator configuration with the compensation capacitor between the negative input and output of the operational amplifier. This is a one-pole system therefore a zero is not needed in the compensation. For offline applications with PFC, the crossover should be set an order of magnitude lower than the line frequency of 120Hz or 100Hz. In a typical application, the compensation capacitor is 0.1μF. In non-PFC applications, the crossover frequency may be increased to improve transient performance. The desired crossover frequency needs to be set an order of magnitude below the switching frequency for optimal performance. Table 1. Predesigned Transformers—Typical Specifications, Unless Otherwise Noted TRANSFORMER SIZE PART NUMBER (L × W × H) JA4429 21.1mm × 21.1mm × 17.3mm LPRI (µH) NPSA (NP:NS:NA) RPRI (mΩ) RSEC (mΩ) MANUFACTURER TARGET APPLICATION (VOUT /IOUT) 400 1:0.24:0.24 252 126 Coilcraft 22V/1A 7508110210 15.75mm × 15mm × 18.5mm 2000 6.67:1:1.67 5100 165 Würth Elektronik 10V/0.4A 750813002 15.75mm × 15mm × 18.5mm 2000 20:1.0:5.0 6100 25 Würth Elektronik 3.8V/1.1A 750811330 43.2mm × 39.6mm × 30.5mm 300 6:1.0:1.0 150 25 Würth Elektronik 18V/5A 750813144 16.5mm × 18mm × 18mm 600 4:1:0.71 2400 420 Würth Elektronik 28V/0.5A 750813134 16.5mm × 18mm × 18mm 600 8:1:1.28 1850 105 Würth Elektronik 14V/1A 750811291 31mm × 31mm × 25mm 400 1:1:0.24 550 1230 Würth Elektronik 85V/0.4A 750813390 43.18mm × 39.6mm × 30.48mm 100 1:1:0.22 150 688 Würth Elektronik 90V/1A 750811290 31mm × 31mm × 25mm 460 1:1:0.17 600 560 Würth Elektronik 125V/0.32A X-11181-002 23.5mm × 21.4mm × 9.5mm 500 72:16:10 1000 80 Premo 30V/0.5A 750811248 31mm × 31mm × 25mm 300 4:1.0:1.0 280 25 Würth Elektronik 24V/2A RLLT-1001 25mm × 22.2mm × 16mm 820 16:1.0:4.0 1150 10 Renco 5V/4A 750312872 43.2mm × 39.6mm × 30.5mm 14 1:1:0.8 11 11 Würth Elektronik 28V/4A 3798fa 14 LT3798 OPERATION MOSFET and Diode Selection Power Factor Correction/Harmonic Content With a strong 1.9A gate driver, the LT3798 can effectively drive most high voltage MOSFETs. A low Qg MOSFET is recommended to maximize efficiency. In most applications, the RDS(ON) should be chosen to limit the temperature rise of the MOSFET. The drain of the MOSFET is stressed to VOUT • NPS + VIN during the time the MOSFET is off and the secondary diode is conducting current. But in most applications, the leakage inductance voltage spike exceeds this voltage. The voltage of this stress is determined by the switch voltage clamp. Always check the switch waveform with an oscilloscope to make sure the leakage inductance voltage spike is below the breakdown voltage of the MOSFET. A transient voltage suppressor and diode are slower than the leakage inductance voltage spike, therefore causing a higher voltage than calculated. The LT3798 attains high power factor and low harmonic content by making the peak current of the main power switch proportional to the line voltage by using an internal multiplier. A power factor of >0.97 is easily attainable for most applications by following the design equations in this data sheet. With proper design, LT3798 applications can easily meet most harmonic standards. The secondary diode stress may be as much as VOUT + 2 • VIN/NPS due to the anode of the diode ringing with the secondary leakage inductance. An RC snubber in parallel with the diode eliminates this ringing, so that the reverse voltage stress is limited to VOUT + VIN/NPS. With a high NPS and output current greater than 3A, the IRMS through the diode can become very high and a low forward drop Schottky is recommended. Discontinuous Mode Detection The discontinuous mode detector uses AC-coupling to detect the ringing on the third winding. A 22pF capacitor with a 30k resistor in series is recommended in most designs. Depending on the amount of leakage inductance ringing, an additional current may be needed to prevent false tripping from the leakage inductance ringing. A resistor from INTVCC to the DCM pin adds this current. Up to an additional 100μA of current may be needed in some cases. The DCM pin is roughly 0.7V, therefore the resistor value is selected using the following equation: R= Operation Under Light Output Loads The LT3798 detects output overvoltage conditions by looking at the voltage on the third winding. The third winding voltage is proportional to the output voltage when the main power switch is off and the secondary diode is conducting current. Sensing the output voltage requires delivering power to the output. When the output current is very low, this periodic delivery of output current can exceed the load current. The OVP pin sets the output overvoltage threshold. When the output of the sample-and-hold is above this voltage, the minimum switching frequency is divided by 8 as shown in Figure 5. This OVP threshold needs to be set above 1.35V and should be set out of the way of output voltage transients. The output clamp point is set with the following formula: VOUT = VOVP(R4 + R5)/(NST • R5)–(VF + (R4•ITC)/NST) The VOVP pin voltage may be provided by a resistor divider from the VREF pin. This frequency division greatly reduces the output current delivered to the output but a Zener or resistor is required to dissipate the remaining output current. The Zener diode’s voltage needs to be 5% higher than the output voltage set by the resistor divider connected to the FB pin. Multiple Zener diodes in series may be needed for higher output power applications to keep the Zener’s temperature within the specification. 10V – 0.7V I where I is equal to the additional current into the DCM pin. 3798fa 15 LT3798 OPERATION Protection from Shorted Output Conditions Usage with DC Input Voltage During a shorted output condition as shown in Figure 6, the LT3798 operates at the minimum operating frequency. In normal operation, the third winding provides power to the IC, but the third winding voltage is zero during a shorted condition. This causes the part’s VIN UVLO to shutdown switching. The part starts switching again when VIN has reached its turn-on voltage. The LT3798 is flexible enough to operate well from low voltage to very high voltage DC input voltage applications. When the supply voltage is less than 40V, the startup resistor is not needed and the part's VIN can be connected directly to the supply voltage. The startup sequence for voltages higher than 40V is the same as what is described for high voltage offline supply voltages. The loop compensation component values can be chosen to provide faster loop response since the LT3798 does not have to provide PFC for the slow 50Hz/60Hz AC input voltage. For DC input applications, connect a 25k resistor from VIN_SENSE to INTVCC. V3RD WINDING 20V/DIV VOUT 10V/DIV IOUT 1A/DIV 1ms/DIV 3798 F05 Figure 5. Switching Waveforms When Output Open-Circuits or at Very Light Load Conditions VIN 20V/DIV V3RD WINDING 50V/DIV IPRI 1A/DIV 100ms/DIV 3798 F06 Figure 6. Switching Waveforms When Output Short-Circuits 3798fa 16 LT3798 TYPICAL APPLICATIONS Universal Input 24W PFC Bus Converter L2 800µH L1 33mH BR1 C1 0.068µF 90V TO 265V AC C2 0.1µF R3 499k R4 499k R5 1M R7 100k R17 D2 20Ω R8 100k C4 C5 4.7pF 10µF D3 4:1:1 R13 2k VIN DCM VIN_SENSE FB R14 90.9k Z1 R15 4.99k EN/UVLO R6 95.3k GATE VREF BR1: DIODES, INC. HD06 C8: VISHAY 440LD22-R D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: DIODES INC. BAV20W D4: CENTRAL SEMICONDUCTOR CMR1U-02M M1: FAIRCHILD FDPF15N65 T1: COILCRAFT JA4429-AL Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ5937B R9 40.2k CTRL3 SENSE CTRL2 INTVCC CTRL1 R12 221k R10 16.5k COMP C3 2.2µF + 24V 1A + C10 560µF ×2 D1 M1 Z2 RS 0.05Ω C8 2.2nF C9 4.7µF GND OVP VC C6 22pF R16 20Ω LT3798 R11 100k D4 COMP– “Y1 CAP” C7, 0.1µF 3798 TA02 Universal Input 48W PFC Application L1 1mH L2 27mH BR1 C1 0.1µF 90V TO 265V AC C2 0.22µF R3 499k R7 100k R4 499k R8 100k R17 D2 47Ω D3 R5 2.4M VIN EN/UVLO R13 33k BR1: DIODES, INC. HD06 C8: VISHAY 440LD22-R C11: MURATA GRM32ER7YA106KA12L D1: CENTRAL SEMICONDUCTOR CMR1U-06M D2, D3: DIODES INC. BAV20W D4: DIODES INC. SBR20A200CTB M1: INFINEON IPB60R165CP T1: WÜRTH ELEKTRONIK 750811248 Z1: FAIRCHILD SMBJ170A Z2: CENTRAL SEMICONDUCTOR CMZ5937B C6 22pF R15 5.49k D4 Z1 R9 40.2k CTRL3 GATE CTRL2 SENSE CTRL1 INTVCC R10 31.6k GND COMP+ VC C3 1µF COMP– C7, 0.1µF R16 20Ω 1000µF ×2 M1 C9 4.7µF 24V 2A + C10 D1 OVP R12 221k R14 100k LT3798 VREF R11 100k 4:1:1 DCM FB VIN_SENSE R6 301k C4 22pF C5 10µF C11 10µF ×2 Z2 RS 0.03Ω C8 2.2nF “Y1 CAP” 3798 TA03 3798fa 17 LT3798 PACKAGE DESCRIPTION Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings. MSE Package 16-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1667 Rev E) BOTTOM VIEW OF EXPOSED PAD OPTION 2.845 ± 0.102 (.112 ± .004) 5.23 (.206) MIN 2.845 ± 0.102 (.112 ± .004) 0.889 ± 0.127 (.035 ± .005) 8 1 1.651 ± 0.102 (.065 ± .004) 1.651 ± 0.102 3.20 – 3.45 (.065 ± .004) (.126 – .136) 0.305 ± 0.038 (.0120 ± .0015) TYP 16 0.50 (.0197) BSC 4.039 ± 0.102 (.159 ± .004) (NOTE 3) RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 0.35 REF 0.12 REF DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY 9 NO MEASUREMENT PURPOSE 0.280 ± 0.076 (.011 ± .003) REF 16151413121110 9 DETAIL “A” 0° – 6° TYP 3.00 ± 0.102 (.118 ± .004) (NOTE 4) 4.90 ± 0.152 (.193 ± .006) GAUGE PLANE 0.53 ± 0.152 (.021 ± .006) DETAIL “A” 1.10 (.043) MAX 0.18 (.007) SEATING PLANE 0.17 – 0.27 (.007 – .011) TYP 1234567 8 0.50 NOTE: (.0197) 1. DIMENSIONS IN MILLIMETER/(INCH) BSC 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE. 0.86 (.034) REF 0.1016 ± 0.0508 (.004 ± .002) MSOP (MSE16) 0911 REV E 3798fa 18 LT3798 REVISION HISTORY REV DATE DESCRIPTION PAGE NUMBER A 12/12 Added H- and MP-grade parts 2, 3 3798fa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LT3798 TYPICAL APPLICATION 112W Wide DC Input Industrial Power Supply VIN 20V TO 60V R17 D2 20Ω R7 6.8k C2 10µF TO INTVCC R8 6.8k R4 24k R5 402k C4 15pF C5 10µF D3 R13 15k VIN DCM FB VIN_SENSE EN/UVLO R6 51.1k 1:1:0.8 R14 100k R15 5.9k LT3798 D4 C6 22pF Z1 Z3 D1 D5 C10 10µF ×4 28V 4A VREF R11 100k R9 40.2k CTRL3 GATE CTRL2 SENSE CTRL1 INTVCC OVP R12 221k R10 34.8k VC M1 C9 4.7µF GND COMP+ C3 0.1µF R3 16.2k COMP– RS 0.004Ω C8 2.2nF “Y1 CAP” C7, 22nF C2: TDK C5750X7S2A106M C8: VISHAY 440LD22-R C10: MURATA GRM32ER7YA106KA12L D1: DIODES INC. DFLS1150 D2, D3: DIODES INC. BAV20W D4: ON SEMICONDUCTOR MBR20200CT D5: DIODES INC. DFLS2100 Z2 3798 TA04 M1: T1: Z1: Z2: Z3: FAIRCHILD FDP2532 WÜRTH ELEKTRONIK 750312872 DIODES INC. SMCJ60A CENTRAL SEMICONDUCTOR CMZ59398 FAIRCHILD SMBJ170A RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT3799/LT3799-1 Offline Isolated Flyback LED Controller with Active PFC No Opto-Coupler Required, TRIAC Dimmable, VIN and VOUT Limited Only by External Components, MSOP-16E LT3748 100V Isolated Flyback Controller 5V ≤ VIN ≤ 100V, No-Opto Flyback , MSOP-16 with High Voltage Spacing LT3573/LT3574/LT3575 40V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 1.25A/0.65A/2.5A Switch LT3511/LT3512 100V Isolated Flyback Converters Monolithic No-Opto Flybacks with Integrated 240mA/420mA Switch LT3757/LT3758 40V/100V Flyback/Boost Controllers Universal Controllers with Small Package and Powerful Gate Drive LT3957/LT3958 40V/100V Flyback/Boost Converters Monolithic with Integrated 5A/3.3A Switch LTC3803/LTC3803-3/LTC3803-5 200kHz/300kHz Flyback Controllers VIN and VOUT Limited Only by External Components LTC3805/LTC3805-5 VIN and VOUT Limited Only by External Components Adjustable Frequency Flyback Controllers 3798fa 20 Linear Technology Corporation LT 1212 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2012