Audio Circuits Using the NE5532/4

AND8177/D
Audio Circuits Using the
NE5532/4
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APPLICATION NOTE
Audio Circuits Using the NE5532/34
The following will explain some of ON Semiconductors
low noise op amps and show their use in some audio
applications.
preamp, input buffer, 5--band equalizer, and mixer.
Although the circuit design is not new, its performance using
the 5532 has been improved.
The RIAA preamp section is a standard compensation
configuration with low frequency boost provided by the
Magnetic cartridge and the RC network in the op amp
feedback loop. Cartridge loading is accomplished via R1.
47 kΩ was chosen as a typical value, and may differ from
cartridge to cartridge.
The Equalizer section consists of an input buffer, 5 active
variable band pass/notch (depending on R9’s setting) filters,
and an output summing amplifier. The input buffer is a
standard unity gain design providing impedance matching
between the preamplifier and the equalizer section. Because
the 5532 is internally--compensated, no external
compensation is required. The 5--band active filter section is
actually five individual active filters with the same feedback
design for all five. The main difference in all five stages is
the values of C5 and C6, which are responsible for setting the
center frequency of each stage. Linear pots are
recommended for R9. To simplify use of this circuit, a
component value table is provided, which lists center
frequencies and their associated capacitor values. Notice
that C5 equals 10 × C6 and the value of R8 and R10 are
related to R9 by a factor of 10 as well. The values listed in
the table are common and easily found standard values.
Description
The 5532 is a dual high--performance low noise
operational amplifier. Compared to most of the standard
operational amplifiers, such as the 1458, it shows better
noise performance, improved output drive capability and
considerably higher small--signal and power bandwidths.
This makes the device especially suitable for application
in high quality and professional audio equipment,
instrumentation and control circuits, and telephone channel
amplifiers. The op amp is internally--compensated for gains
equal to one. If very low noise is of prime importance, it is
recommended that the 5532A version be used which has
guaranteed noise voltage specifications.
APPLICATIONS
ON Semiconductors 5532 High--Performance Op Amp is
an ideal amplifier for use in high quality and professional
audio equipment which requires low noise and low
distortion.
The circuit (Figure 1) included in this application note has
been assembled on a PC board, and tested with actual audio
input devices (Tuner and Turntable). It consists of a
Recording Industry Association of America (RIAA)
C5
RIAA
C1
3
2
R1
RIAA Out
+
1/2 5532
1
Equ In
R5
--
5
6
+
1/2 5532
-R5
R2
R3
C2
C3
7
R7
R7
R8
R9 C6
R9
2
--
1/2 5532
3 +
R11
1
R10
+
C7
6
--
1/2 5532
5 +
7
FLAT
TO VOL./
BAL AMP
EQUALIZE
R4
C4
REPEAT ABOVE CIRCUIT
FOR DESIRE NO. OF
STAGES.
R12
Figure 1. RIAA -- Equalizer Schematic
© Semiconductor Components Industries, LLC, 2005
November, 2005 -- Rev. 0
1
Publication Order Number:
AND8177/D
AND8177/D
RIAA Equalization Audio Preamplifier Using
NE5532A
With the onset of new recording techniques with
sophisticated playback equipment, a new breed of low noise
operational amplifiers was developed to complement the
state--of--the--art in audio reproduction. The first ultra--low
noise op amp introduced by ON Semiconductors was called
the NE5534A. This is a single operational amplifier with
sub--audible tones, they become quite objectionable because
the speakers try to reproduce these tones. This causes
non--linearities when the actual recorded material is
amplified and converted to sound waves.
The RIAA has proposed a change in its standard playback
response curve in order to alleviate some of the problems
that were previously discussed. The changes occur primarily
at the low frequency range with a slight modification to the
high frequency range (See Figure 2). Note that the response
peak for the bass section of the playback curve now occurs
at 31.5 Hz and begins to roll off below that frequency. The
roll--off occurs by introducing a fourth RC network with a
7950 ms time constant to the three existing networks that
make up the equalization circuit. The high end of the
equalization curve is extended to 20 kHz, because
recordings at these frequencies are achievable on many
current discs.
less than 4 nV∕ Hz input noise voltage. The NE5534A is
internally--compensated at a gain of three. This device has
been used in many audio preamp and equalizer (active filter)
applications since its introduction.
Many of the amplifiers that are being designed today are
DC--coupled. This means that very low frequencies
(2--15 Hz) are being amplified. These low frequencies are
common to turntables because of rumble and tone arm
resonances. Since the amplifiers can reproduce these
--25
OLD RIAA
--20
--15
--10
NEW RIAA
--5
0
(dB) 5
10
15
20
25
30
1
10
100
(Hz)
1K
10K
100K
Figure 2. Proposed RIAA Playback Equalization
--15 V
.1 mF
.27 mF
+
3
INPUT
8
47 KΩ
NE5532A
TO LOAD
1
4
2
---15 V
.1 mF
49.9 KΩ
49.9 Ω
.056 mF
4.99 KΩ
47 mF
.015 mF
NOTE:
All resistors are 1% metal film.
Figure 3. RIAA Phonograph Preamplifier Using the NE5532A
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AND8177/D
COMPONENT VALUES FOR FIGURE 1
R8 = 25 kΩ
R7 = 2.4 kΩ R9 = 240 kΩ
R8 = 50 kΩ
R7 = 5.1 kΩ R9 = 510 kΩ
R8 = 100 kΩ
R7 = 10 kΩ R9 = 1 megΩ
fO
C5
C6
fO
C5
C6
fO
C5
C6
23 Hz
1 mF
0.1 mF
25 Hz
0.47 mF
0.047 mF
12 Hz
0.47 mF
0.047 mF
50 Hz
0.47 mF
0.047 mF
36 Hz
0.33 mF
0.033 mF
18 Hz
0.33 mF
0.033 mF
72 Hz
0.33 mF
0.033 mF
54 Hz
0.22 mF
0.022 mF
27 Hz
0.22 mF
0.022 mF
108 Hz
0.22 mF
0.022 mF
79 Hz
0.15 mF
0.015 mF
39 Hz
0.15 mF
0.015 mF
158 Hz
0.15 mF
0.015 mF
119 Hz
0.1 mF
0.01 mF
59 Hz
0.1 mF
0.01 mF
238 Hz
0.1 mF
0.01 mF
145 Hz
0.082 mF
0.0082 mF
72 Hz
0.082 mF
0.0082 mF
290 Hz
0.082 mF
0.0082 mF
175 Hz
0.068 mF
0.0068 mF
87 Hz
0.068 mF
0.0068 mF
350 Hz
0.068 mF
0.0068 mF
212 Hz
0.056 mF
0.0056 mF
106 Hz
0.056 mF
0.0056 mF
425 Hz
0.056 mF
0.0056 mF
253 Hz
0.047 mF
0.0047 mF
126 Hz
0.047 mF
0.0047 mF
506 Hz
0.047 mF
0.0047 mF
360 Hz
0.033 mF
0.0033 mF
180 Hz
0.033 mF
0.0033 mF
721 Hz
0.033 mF
0.0033 mF
541 Hz
0.022 mF
0.0022 mF
270 Hz
0.022 mF
0.0022 mF
1082 Hz
0.022 mF
0.0022 mF
794 Hz
0.015 mF
0.0015 mF
397 Hz
0.015 mF
0.0015 mF
1588 Hz
0.015 mF
0.0015 mF
1191 Hz
0.01 mF
0.001 mF
595 Hz
0.01 mF
0.001 mF
2382 Hz
0.01 mF
0.001 mF
1452 Hz
0.0082 mF
820 pF
726 Hz
0.0082 mF
820 pF
2904 Hz
0.0082 mF
820 pF
1751 Hz
0.0068 mF
680 pF
875 Hz
0.0068 mF
680 pF
3502 Hz
0.0068 mF
680 pF
2126 Hz
0.0056 mF
560 pF
1063 Hz
0.0056 mF
560 pF
4253 Hz
0.0056 mF
560 pF
2534 Hz
0.0047 mF
470 pF
1267 Hz
0.0047 mF
470 pF
5068 Hz
0.0047 mF
470 pF
3609 Hz
0.0033 mF
330 pF
1804 Hz
0.0033 mF
330 pF
7218 Hz
0.0033 mF
330 pF
5413 Hz
0.0022 mF
220 pF
2706 Hz
0.0022 mF
220 pF
10827 Hz
0.0022 mF
220 pF
7940 Hz
0.0015 mF
150 pF
3970 Hz
0.0015 mF
150 pF
15880 Hz
0.0015 mF
150 pF
11910 Hz
0.001 mF
100 pF
5955 Hz
0.001 mF
100 pF
23820 Hz
0.001 mF
100 pF
14524 Hz
820 pF
82 pF
7262 Hz
820 pF
82 pF
17514 Hz
680 pF
68 pF
8757 Hz
680 pF
68 pF
21267 Hz
560 pF
56 pF
10633 Hz
560 pF
56 pF
12670 Hz
470 pF
47 pF
18045 Hz
330 pF
33 pF
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AND8177/D
NE5534 Description
The 5534 is a single high--performance low noise
operational amplifier. Compared to other operational
amplifiers, such as TL083, they show better noise
performance, improved output drive capability and
considerably higher small--signal and power bandwidths.
This makes the devices especially suitable for application
in high quality and professional audio equipment,
instrumentation and control circuits, and telephone channel
amplifiers.
The op amps are internally--compensated for gain equal
to, or higher than, three. The frequency response can be
optimized with an external compensation capacitor for
various applications (unity gain amplifier, capacitive load,
slew rate, low overshoot, etc.) If very low noise is of prime
importance, it is recommended that the 5534A version be
used which has guaranteed noise specifications.
R1 C1 circuit deleted, the device slew rate falls to
approximately 7 V/ ms. The input waveform will reach 2
V/25 0V/ ms or 8 ns, while the output will have changed (8
× 10--3) only 56 mV. The differential input signal is then (VIN
-- VO) RI/RI + RF or approximately 1 V.
The diode limiter will definitely be active and output
distortion will occur; therefore, VIN < 1 V as indicated.
Next, a sine wave input is used with a similar circuit.
The slew rate of the input waveform now depends on
frequency and the exact expression is:
dv = 2ω cos ωt
dt
The upper limit before slew rate distortion occurs for
small--signal (VIN < 100 mV) conditions is found by setting
the slew rate to 7 V/ ms. That is:
7 x 10 6 V∕ms = 2ω cos ωt
at ωt = 0
APPLICATIONS
6
ω LIMIT = 7 x 10 = 3.5 x 10 6 rad∕s
2
Diode Protection of Input
The input leads of the device are protected from
differential transients above ±0.6 V by internal
back--to--back diodes (Figure 4). Their presence imposes
certain limitations on the amplifier dynamic characteristics
related to closed--loop gain and slew rate.
6
f LIMIT = 3.5 x 10 ≈ 560 kHz
2π
dV/dt
+2
---2V
VIN = 2 Sin ωt
1 KΩ
+
22 pF
1 KΩ
+
NE
5534
Figure 4.
Consider the unity gain follower as an example.
Assume a signal input square wave with dV/dt of
250 V/ ms and 2 V peak amplitude as shown (Figures 5
and 6). If a 22 pF compensation capacitor is inserted and the
Figure 5.
RF
5
Rt
2V
R1
0
∆t1
--Vin
C1
22 pF
CC
-2
8
NE
5534
3
+
0
--VO
6
∆t2
Figure 6.
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AND8177/D
External Compensation Network Improves Bandwidth
RF
By using an external lead--lag network, the follower
circuit slew rate and small--signal bandwidth can be
increased. This may be useful in situations where a
closed--loop gain less than 3 to 5 is indicated. A number of
examples are shown in subsequent figures. The principle
benefit of using the network approach is that the full slew
rate and bandwidth of the device is retained, while
impulse--related parameters such as damping and phase
margin are controlled by choosing the appropriate circuit
constants. For example, consider the following
configuration (Figure 7):
1 KΩ
Ri
LAG
NETWORK
-R
NE5534
C
+
Figure 7.
GAIN
PHASE
0
90
45
θ --90o
LAG NETWORKS
0
0
0.1
1.0
10
50
--180o
0
0.1
MHz
Figure 8.
10
50
Figure 9.
The major problem to be overcome is poor phase margin
leading to instability.
By choosing the lag network break frequency one decade
below the unity gain crossover frequency (30--50 MHz), the
phase and gain margin are improved (see Figures 8 and 9).
An appropriate value for R is 270 Ω. Setting the lag network
break frequency at 5 MHz, C may be calculated:
C=
1.0
MHz
2
-6
3
VIN
VOUT
8
+
NOTES
5
C1 = CC(1)
C1
CC = 22 pF for NE5533/34
CC = 22 pF [See graph under typical performance characteristics]
1
2 π ⋅ 270 ⋅ 5 x 10 6
Figure 10. Unity Gain Non--Inverting Configuration
= 118 pF
RF
Rules and Examples
Compensation Using Pins 5 and 8 (Limited Bandwidth
and Slew Rate)
VIN
A single--pole and zero inserted in the transfer function
will give an added 45° of phase margin, depending on the
network values.
RIN
2
-6
3
8
+
VOUT
5
C1
Calculating the Lead--Lag Network
C1 =
Figure 11. Unity Gain Inverting Configuration
R
1
Let R 1 = IN
10
2 π F1 R1
where
F 1 = 1 (UGBW)
10
UGBW = 30 MHz
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AND8177/D
External Compensation for Wide--Band
Voltage--Follower
CF
Shunt Capacitance Compensation
RF
CF =
or
CF ≈
1
, F F ≈ 30MHz
2π F F R F
RIN
VIN
C DIST
-C1
VOUT
A CL
CDIST ≈ Distributed Capacitance ≈ 2 -- 3 pF
Many audio circuits involve carefully--tailored frequency
responses. Pre--emphasis is used in all recording mediums to
reduce noise and produce flat frequency response. The most
often used de--emphasis curves for broadcast and home
entertainment systems are shown in Figures 13 through 17
on the following page. Operational amplifiers are well
suited to these applications because of their high gain and
easily--tailored frequency response.
R1
+
NOTE:
Input diodes limit differential to <0.5V
Figure 12. External Compensation for Wideband
Voltage Follower
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AND8177/D
30
20
5
0
--5
--10
TIME CONSTANTS
3150 ms
50 ms
30
RELATIVE GAIN (dB)
10
TURN OVER FREQUENCIES
50 Hz, 3180 Hz
35
TIME CONSTANTS
3150 ms
318 ms
75 ms
15
25
20
15
10
--15
5
--20
0
--25
--30
10
100
1K
10K
FREQUENCY (Hz)
10
100K
Figure 13. RIAA Equalization
1K
10K
FREQUENCY (Hz)
100K
TURN OVER FREQUENCIES
50 Hz, 1326 Hz
35
RELATIVE GAIN (dB)
100
Figure 14. NAB Standard Playback 71/2 IPS
40
TIME CONSTANTS
3150 ms
125 ms
30
25
20
15
10
5
0
10
100
1K
10K
100K
FREQUENCY (Hz)
Figure 15. 3.75 IPS Tape Equalization
25
20
5
TURN OVER FREQUENCY 1 kHz
--5
RELATIVE GAIN (dB)
10
5
0
--5
--10
--10
--15
--20
--25
--15
--30
--20
--25
10
TURN OVER FREQUENCY 2122 CPS
TIME CONSTANT 75 ms
0
15
RELATIVE GAIN (dB)
RELATIVE GAIN 9dB)
40
TURN OVER FREQUENCIES
50 Hz, 500 Hz, 2122 Hz
25
100
1K
10K
FREQUENCY (Hz)
--35
100K
10
100
1K
10K
100K
FREQUENCY (Hz)
Figure 16. Base Treble Curve
Figure 17. Standard FM Broadcast Equalization
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AND8177/D
RIAA PREAMP USING THE NE5534
The preamplifier for phono equalization is shown in
Figure 18 with the theoretical and actual circuit response.
Low frequency boost is provided by the inductance of the
magnetic cartridge with the RC network providing the necessary
break points to approximate the theoretical RIAA curve.
--15 V
0.22 mF
+
INPUT
RSL
1.1 MΩ
OUTPUT
NE5534
--
1.1 KΩ
--15 V
100 KΩ
20 mF
750 pF
NOTES:
*Select to provide specified transducer loading.
Output Noise ≥0.8mVRMS (with input shorted)
1 MΩ
RIAA
NAB
0.0033 mF
1.1 MΩ
0.003 mF
16 KΩ
a.
70
70
BODE PLOT
GAIN — dB
GAIN — dB
50
ACTUAL
RESPONSE
50
40
BODE PLOT
60
60
30
30
20
20
10
10
0
101
ACTUAL
RESPONSE
40
0
102
103
104
105
101
102
103
104
105
FREQUENCY (Hz)
FREQUENCY (Hz)
c. Bode Plot of NAB Equalization and the Response
Realized in the Actual Circuit Using the 531.
b. Bode Plot of RIAA Equalization and the Response
Realized in an Actual Circuit Using the 531.
Figure 18. Preamplifier -- RIAA/NAB Compensation
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AND8177/D
RUMBLE FILTER
Following the amplifier stage, rumble and scratch filters
are often used to improve overall quality. Such a filter
designed with op amps uses the 2--pole Butterworth
approach and features switchable break points. With the
circuit of Figure 19, any degree of filtering from fairly sharp
to none at all is switch--selectable.
1
20 KΩ
10 KΩ
0.1 mF
-NE5534
0.1 mF
2
--
6.8 KΩ 3
4
+
100 Ω
0.0022 mF NE5534
+
1
220 kΩ
75 kΩ
2
47 kΩ
3
27 kΩ
20 KΩ
10 KΩ
4
0.0056 mF
6.8 KΩ
39 kΩ
20 KΩ
22 kΩ
10 KΩ
13 kΩ
6.8 KΩ
RUMBLE
POSITION FREQ.
1
FLAT
2
30 MHz
3
50 Hz
4
80 Hz
SCRATCH
POSITION FREQ.
1
5 kHz
2
10 MHz
3
15 Hz
4
FLAT
330 pF
6.8 KΩ
Figure 19. Rumble/Scratch Filter
TONE CONTROL
Tone control of audio systems involves altering the flat
response in order to attain more low frequencies or more
high ones, dependent upon listener preference. The circuit
10 KΩ
100 KΩ
+140
10 KΩ
+30
0.033 mF
10 KΩ
+
OUTPUT
5V
PEAK TO PEAK
A
--
3.3 KΩ
0.033 mF
MAX
BASS
BOOST
MAX
TREBLE
BOOST
MAX
BASS
CUT
MAX
TREBLE
CUT
+20
V+
0.033 mF
GAIN (dB)
1 mF
INPUT
of Figure 20 provides 20 dB of bass or treble boost or cut as
set by the variable resistance. The actual response of the
circuit is shown also.
+10
0
--10
--20
0.033 mF
68 KΩ
--30
V--
100 KΩ
--40
NOTES:
1. Amplifier A may be a NE531 or 301. Frequency compensation, as for unity gain non--inverting amplifiers, must be used.
2. Turn--over frequency -- 1k Hz.
3. Base boost +20 dB, bass cut --20 dB, treble boost +19 dB at 20 Hz, treble cut --19 dB at 20 Hz.
10
Figure 20. Tone Control Circuit for Operational Amplifiers
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100
1,000
10,000
FREQUENCY (Hz)
100,000
AND8177/D
BALANCE AND LOUDNESS AMPLIFIER
Figure 21 shows a combination of balance and loudness
controls. Due to the non--linearity of the human hearing
system, the low frequencies must be boosted at low listening
levels. Balance, level, and loudness controls provide all the
listening controls to produce the desired music response.
100 KΩ
0.5 mF
A IN
LEVEL
100 KΩ
-5534
+
220 pF
-100 KΩ
4.7 KΩ
120 Ω
A OUT
OUT
BALANCE
26 KΩ
5534
+
LOUDNESS
IN
1.2 KΩ
4.7 KΩ
0.33 mF
100 KΩ
100 KΩ
0.5 mF
B IN
-100 KΩ
5534
+
220 pF
-5534
1290 Ω
100 KΩ
+
1.2 KΩ
0.33 mF
100 KΩ
Figure 21. Balance Amplifier with Loudness Control
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B OUT
AND8177/D
VOLTAGE AND CURRENT OFFSET ADJUSTMENTS
Many IC amplifiers include the necessary pin connections
to provide external offset adjustments. Many times, however,
it becomes necessary to select a device not possessing
external adjustments. Figures 22, 23, and 24 suggest some
possible arrangements for off--circuitry. The circuitry of
Figure 24 provides sufficient current into the input to cancel
the bias current requirement. Although more simplified
arrangements are possible, the addition of Q2 and Q3 provide
a fixed current level to Q1, thus, bias cancellation can be
provided without regard to input voltage level.
R3
+V
INPUT
R4
+
OUTPUT
NE5534
R5
50 KΩ
R1
100 KΩ
R1
200 KΩ
R5
50 KΩ
--
R3
R4
--
OUTPUT
NE5534
RANGE = V
R2
100 Ω
RR21
--V
R2
100 Ω
+
RANGE = V
RR21
GAIN = 1 +
R3
R4 + R2
INPUT
Figure 23. Universal Offset Null for Non--Inverting
Amplifiers
Figure 22. Universal Offset Null for Inverting
Amplifiers
BIAS CURRENT
COMPENSATION
V+
R3
R1
Q3
Q2
R2
Q1
-NE5534
+
VIN
VOUT
SELECT R2 AND R3 FOR
DESIRED CURRENT
V--
Figure 24. Bias Current Compensation
ChipFET is a trademark of Vishay Siliconix. POWERMITE is a registered trademark of and used under a license from Microsemi Corporation.
ON Semiconductor and
are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice
to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
“Typical” parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. SCILLC does not convey any license under its patent rights
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