AN-1318 APPLICATION NOTE One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106, U.S.A. • Tel: 781.329.4700 • Fax: 781.461.3113 • www.analog.com Differential Overvoltage Protection Circuits for Current Sense Amplifiers by Emmanuel Adrados and Paul Blanchard INTRODUCTION Most current sense amplifiers are capable of handling high common-mode voltages (CMVs) but not high differential input voltages. In certain applications, there are fault conditions wherein the differential input voltage at the shunt exceeds the specified maximum voltage of the amplifier. These conditions can cause damage to the amplifier. This application note introduces two basic overvoltage protection circuits for current sense amplifiers and discusses the effects of the circuits on device performance for two types of current sense architectures—a current sense amplifier (using the AD8210 as an example) and a difference amplifier (using the AD8418 as an example). VIN _ MAX − VRATED _ MAX R = 3 mA (1) where: VIN_MAX is the expected maximum differential voltage. VRATED_MAX is the maximum rated voltage (0.7 V). R is the total series resistance (R1 + R2). For example, if the expected maximum transient input voltage is 10 V, the equation is 10 V − 0.7 V R = 3 mA (2) OVERVOLTAGE PROTECTION CIRCUIT If R = 3.1 kΩ, then based on Equation 1, R1 and R2 = 1.55 kΩ. Figure 1 shows the basic connection for overvoltage protection of a current sense amplifier. When the differential input voltage exceeds the maximum rated value for a given amplifier, the amplifier may begin to pull current into the internal protection diodes. The additional series resistors, R1 and R2, prevent large current flow to the internal protection diodes if a large differential voltage signal is present between the input pins. These values for R1 and R2 are significant, relative to the input impedance of certain amplifiers, and can contribute a large error to the overall system performance. Keep the values of R1 and R2 as low as possible to minimize error contribution. One way to reduce the value of R1 and R2 is to add external protection diodes with higher current capabilities on the input pins, as shown in Figure 2. +5V +5V VS R1 VREF1 VS R2 GND 10Ω VCM GND D1 VREF2 D2 IN GND Figure 2. Overvoltage Protection Circuit with External Input Differential Protection Diodes 12582-001 GND VREF1 OUT R2 VREF2 VCM VS V1 OUT V1 IN 10Ω 12582-002 VS R1 Figure 1. Basic Overvoltage Protection Circuit Both the maximum rated voltage and the maximum input current tolerated by a protection circuit vary from device to device. As a general rule, limit the current passing through the internal differential protection diodes to 3 mA. Given this value, calculate the values of R1 and R2 using Equation 1. For example, when using the Digi-Key B0520LW-7-F Schottky diode, which can handle up to 500 mA of forward current, the value of R decreases to 20 Ω. Rev. 0 | Page 1 of 3 AN-1318 Application Note In addition to protection against overvoltage, R1 and R2 can also be used to form an electromagnetic interference (EMI) filter by adding capacitors, as shown in Figure 3. This configuration helps the circuit reject any high frequency interference from external sources such as mobile communications, electric motors, and utility power lines. Both the AD8210 and the AD8418 are evaluated to measure the impact of the additional series resistors on the gain error, CMRR, and offset voltage parameters of the devices. Gain Error The gain error of an amplifier is usually specified in the corresponding data sheet. Table 1 shows the calculated additional gain error and the actual gain error of the AD8210. +5V R1 10Ω C1 V1 VREF1 C3 R2 The AD8418 is tested with and without the protection circuit. Table 2 shows the calculated additional gain error and the actual gain error of the amplifier. VS OUT D1 C2 Common-Mode Rejection Ratio VREF2 D2 GND 10Ω 12582-003 VCM Figure 3. Overvoltage Protection Circuit with EMI Filters There are two types of bandwidths to consider for this type of EMI filter circuit—differential and common-mode. These bandwidths are determined using Equation 3 and Equation 4, respectively. Differential Filter Bandwidth (−3 dB) = 1 C1 × C2 2π × (R1 + R2 ) × + C3 C1 + C2 (3) Because current sense amplifiers are usually exposed to environments with high CMV, CMRR is one of the most important specifications. CMRR assesses the ability of a device to reject high CMVs and attain optimal accuracy and performance. It refers to a measure of change in output voltage when equal voltage is applied at the two input terminals of the amplifier. CMRR is defined as a ratio of the differential gain to the common-mode gain and is usually specified in decibels. Use Equation 5 and Equation 6 to find the CMRR values for both amplifiers. CMRR = Common-Mode Filter Bandwidth (−3 dB) = 1 2π × R1× C1 20∆VCM CMRR = 20 log (4) ∆VOUT TRADE-OFFS IN SYSTEM PERFORMANCE Adding series resistors to the input of the amplifier can degrade certain performance parameters. In some amplifiers, R1 and R2 appear in series with internal precision resistors. In other amplifiers, offset currents work with the resistors to create offset voltages. The parameters most likely to be affected are gain error, common-mode rejection ratio (CMRR), and offset voltage. The test setup used for evaluating gain error, CMRR, and offset voltage is shown in Figure 4. This setup uses the Agilent E3631A power supply for providing the 5 V single-supply to the device, the Yokogawa GS200 precision dc source for the differential input voltage signal, the HAMEG HMP4030 for setting the CMV, and the Agilent 3458A precision multimeter for measuring the output voltage of the current sense amplifiers. AGILENT E3631A +5V POWER SUPPLY R1 HAMEG HMP4030 The results indicate that the effect of the additional external series resistors on CMRR performance impacts the AD8418 but does not significantly affect the AD8210. AGILENT 3458A METER VREF1 OUT R2 10Ω D1 D2 (6) The CMRR measurement results for the AD8210 and the AD8418 current sense amplifiers are shown in Table 3 and Table 4, respectively. VS 10Ω (5) where: ADM is the differential gain of the AD8210 and the AD8418 (ADM = 20). ACM is the common-mode gain, ΔVOUT/ΔVCM. VREF2 GND 12582-004 YOKOGAWA GS200 PRECISION DC SOURCE 20 × ∆VCM ADM = ∆VOUT ACM Figure 4. Test Setup for Evaluating the Gain Error, CMRR, and Offset Voltage Rev. 0 | Page 2 of 3 Application Note AN-1318 Table 1. AD8210 Gain Error R1 (Ω) 0 10.2 R2 (Ω) 0 10.2 Additional Gain Error (%) 0 0.497 Actual Gain (V/V) 19.9781 19.88059 Actual Gain Error (%) −0.1095 −0.59705 Additional Gain Error (%) 0 0.013 Actual Gain (V/V) 19.99815 19.9955 Actual Gain Error (%) −0.00925 −0.0225 Table 2. AD8418 Gain Error R1 (Ω) 0 10.2 R2 (Ω) 0 10.2 Table 3. AD8210 CMRR Performance at a Gain of 20 R1 (Ω) 0 10.2 R2 (Ω) 0 10.2 CMV = 0 V and 4 V (dB) −92.77 −94.37 CMV = 4 V and 6 V (dB) −104.96 −107.99 CMV = 4 V and 65 V (dB) −121.49 −121.86 CMV = 6 V and 65 V (dB) −123.35 −123.10 Table 4. AD8418 CMRR Performance at a Gain of 20 R1 (Ω) 0 10.2 R2 (Ω) 0 10.2 CMV = 0 V and 35 V (dB) −127.72 −88.89 CMV = 35 V and 70 V (dB) −123.72 −104.35 CMV = 0 V and 70 V (dB) −138.39 −93.05 Table 5. AD8210 Additional Offset Voltage Due to Input Offset Current and External Impedances R1 (Ω) 0 10.2 R2 (Ω) 0 10.2 VOUT (mV) 5.598 5.938 Additional Offset Voltage (RTI) (μV) 17 17 Table 6. AD8418 Additional Offset Voltage Due to Input Offset Current and External Impedances R1 (Ω) 0 10.2 R2 (Ω) 0 10.2 VOUT (mV) −0.91 26.09 Additional Offset Voltage (RTI) (mV) 1.3 1.3 Offset Voltage When bias currents pass through the external resistors, they produce a voltage in series with the intrinsic offset voltage of the device. To compute this additional offset voltage, multiply the input offset current (IOS), the difference between the two input bias currents, by the external impedance present on the input pins, as shown in Equation 7. Offset Voltage = IOS × R (7) where: IOS is the input offset current. R is the additional external impedance. The increase in offset voltage based on the actual measurements from both the AD8210 and the AD8418 current sense amplifiers are shown in Table 5 and Table 6, respectively. The results show that the increase in offset voltage in the AD8418 is larger than the increase in offset voltage in the AD8210. This is caused by the considerably high input offset current of the AD8418, which is around 100 μA. Any additional impedances present on the input pins, together with the input offset current, reflect as added offset voltage. Therefore, it is recommended to use smaller values for the series resistors of the AD8418 to minimize the effect on offset voltage. CONCLUSION Implementing additional series resistors on the input pins is the simplest way to protect a current sense amplifier against overvoltage. The circuit gives the user extra headroom for the allowable input voltage but at the cost of extra components. The amount of impact on performance such as gain error, CMRR, and offset voltage is measurable and largely related to the magnitude of the total external resistors and the type of current sense amplifier used. For more information on overvoltage protection for robust amplifiers, see the Analog Dialogue article “Robust Amplifiers Provide Integrated Overvoltage Protection.” ©2015 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. AN12582-0-3/15(0) Rev. 0 | Page 3 of 3

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