Application Notes

AN11588
GreenChip TEA1832(L)TS fixed frequency flyback controller
Rev. 2 — 24 June 2015
Application note
Document information
Info
Content
Keywords
GreenChip, TEA1832(L)TS, SMPS, flyback, adapter, notebook, ultralight
notebook, LCD TV, LCD monitors
Abstract
The TEA1832(L)TS is a low-cost member of the GreenChip family in a
very small package. It is a fixed-frequency flyback controller, especially
suited for medium-power applications, such as ultralight notebook,
notebooks, printers, LCD TVs, and LCD monitors.
No-load power can be as low as 55 mW at 230 V (AC).
Delivery of 200 % peak power for short periods is possible.
AN11588
NXP Semiconductors
GreenChip TEA1832(L)TS fixed frequency flyback controller
Revision history
Rev
Date
Description
v.2
20150624
second issue
•
Modifications:
v.1
20141217
Text and graphics have been updated throughout the document.
first issue
Contact information
For more information, please visit: http://www.nxp.com
For sales office addresses, please send an email to: [email protected]
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1. General description
The TEA1832(L)TS is a fixed-frequency flyback controller that can be used for
Discontinuous Conduction Mode (DCM) and Continuous Conduction Mode (CCM).
Despite the very small TSOP6 package, it almost has the full functionality of the TEA1738
series.
1.1 Scope
This application note describes the functionality of the TEA1832(L)TS series.
Fixed-frequency flyback fundamentals and calculation of transformer and other large
signal parts are not dealt with in this document.
1.2 Features
1.2.1 Power features
• SMPS controller IC enabling low-cost applications
• Small low-cost TSOP6 package
• Fixed switching frequency with frequency jitter to reduce ElectroMagnetic Interference
(EMI)
• Frequency reduction with fixed minimum peak current to maintain high efficiency and
low output ripple at low-power output levels
• Peak power operation in CCM, increased peak current (1.4 times) and switching
frequency (2 times)
•
•
•
•
Slope compensation for CCM operation
Integrated soft start
High/low line compensation for constant overpower protection level
Wide VCC voltage range (10.5 V to 36 V)
1.2.2 Green features
• No-load power consumption < 55 mW at 230 V (AC) possible for 65 W applications.
• Very low supply current during start and restart (11 A typical) enabling the use of a
high-ohmic start-up resistor.
• Burst mode when the voltage on the VCC pin drops to the burst threshold level (just
above UnderVoltage LockOut (UVLO) level) to prevent a restart during prolonged no
switching. It enables using a small VCC capacitor and a high-ohmic start-up resistor.
• Low supply current during normal operation (0.58 mA typical).
1.2.3 Protection features
•
•
•
•
•
AN11588
Application note
Accurate OverVoltage Protection (OVP) via the ISENSE pin
OverVoltage Protection on the VCC supply pin
UnderVoltage LockOut (UVLO) on the VCC supply pin
Internal OverTemperature Protection (OTP)
External OverTemperature Protection (OTP) using 100 k NTC resistor
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• Overpower time-out: 27.5 ms (TEA1832TS); 160 ms (TEA1832LTS)
• Overpower time-out during output short circuit conditions, reducing the average input
power: 14.5 ms (TEA1832TS only)
• Restart timer after an overpower condition for low average input power at overload
and short circuit (TEA1832TS only)
• Output Short Circuit Protection (OSCP) to prevent transformer saturation during
start-up, overload events or short circuit events
• Brownin and brownout protection to avoid operation during mains undervoltage
conditions
• Maximum duty cycle protection for  > 90 %
1.3 Applications
The TEA1832(L)TS is intended for applications that require an efficient and cost-effective
power supply solution. It is especially suited for medium-power applications like:
• Ultrabooks and notebooks
• LCD TVs and LCD monitors
• Printers
1.4 Differences between the TEA1832 and the TEA1738 series
• Smaller package (TSOP6):
• The VINSENSE pin has been removed. The functionality has been combined on the
PROTECT pin with the external overtemperature protection.
• The OPTIMER pin has been removed. The functionality has been integrated:
– Internal overpower time-out: 27.5 ms (TEA1832TS); 160 ms (TEA1832LTS);
external timing components no longer required.
– Internal restart timer; extra external components for time constant no longer
required.
• Increased rating of the VCC clamp (1 mA instead of 730 A).
• Extra filtering on latched protections:
– Latched protection can only be triggered if a fault condition lasts at least four
consecutive switching cycles or measuring cycles (external OTP).
– Low-pass filtering provides immunity against high-frequency signals, for example,
from mobile phones.
• Fixed frequency operation with frequency jitter for simple EMI filtering (no frequency
reduction for lower power).
• Maximum duty cycle protection modified:
In the TEA1738 series, the maximum on-time protection is only active during peak
power conditions (Vctrl(Ipeak) > 400 mV). This restriction has been removed because of
the new frequency control implementation. The maximum duty cycle is increased from
80 % to 90 %.
• VCC range increased to 10.5 V to 36 V (was 12 V to 30 V)..
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• New features:
– Burst mode when the voltage on the VCC pin drops to below the burst threshold
level (just above UVLO) preventing restart during prolonged no switching. It
enables using a small VCC capacitor and a high-ohmic start-up resistor.
– Output Short Circuit Protection (OSCP) to prevent transformer saturation at
start-up, overload events, and short circuit events.
1.5 Differences between the TEA1832 and the TEA1733 series
• Smaller package (TSOP6):
• The VINSENSE pin has been removed. The functionality has been partly combined
on the PROTECT pin with the external OTP; the mains OVP function does not exist in
the TEA1832(L)TS anymore:
• The OPTIMER pin has been removed. The functionality has been integrated:
– Internal overpower time-out: 27.5 ms (TEA1832TS); 160 ms (TEA1832LTS);
external timing components no longer required
– Internal restart timer; extra external components for time constant no longer
required
• Internal OverVoltage Protection (OVP) added. If the VCC pin exceeds 36 V, the
latched protection mode is triggered
• Increased rating of the VCC clamp (1 mA instead of 240 A)
• Extra filtering on latched protections:
– Latched protection can only be triggered if a fault condition lasts at least four
consecutive switching cycles or measuring cycles (external OTP)
– Low-pass filtering provides immunity against high-frequency signals; for example,
from mobile phones
• Maximum duty cycle protection added; when 8 cycles  > 90 %, the restart protection
is activated
• Fixed frequency operation with frequency jitter for simple EMI filtering (no frequency
reduction for lower power, as with the TEA1733)
• Increased switching frequency during peak load allowing more output power with the
same core
• New features:
– Burst mode when the voltage on the VCC pin drops to below the burst threshold
level (just above UVLO) preventing restart during prolonged no switching. It
enables using a small VCC capacitor and a high-ohmic start-up resistor
– Output Short Circuit Protection (OSCP) to prevent transformer saturation at
start-up, overload events, and short circuit events
• Latch version (TEA1832LTS) only:
UnderVoltage LockOut (UVLO) has changed to latched protection. It ensures that a
shorted output always triggers the latched protection when VCC drops below Vth(UVLO)
before the overpower protection has a chance to respond.
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GreenChip TEA1832(L)TS fixed frequency flyback controller
1.6 Latch and safe restart version
The TEA1832(L)TS is available in a restart version and a latch version. The only
differences between the two versions are the overpower time-out and how the OverPower
Protection (OPP) and UnderVoltage LockOut (UVLO) events are handled:
• TEA1832TS: Overpower time-out = 27.5 ms, OPP, or UVLO event initiates safe
restart
• TEA1832LTS: Overpower time-out = 160 ms, OPP, or UVLO event sets the IC to the
latched off-state
See Section 3.4 for more detailed information about these protection features.
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Application note
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Rev. 2 — 24 June 2015
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AN11588
Application note
1.7 Application diagram
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TEA1832(L)TS application diagram - 65 W
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GreenChip TEA1832(L)TS fixed frequency flyback controller
Rev. 2 — 24 June 2015
All information provided in this document is subject to legal disclaimers.
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AN11588
NXP Semiconductors
GreenChip TEA1832(L)TS fixed frequency flyback controller
2. Pinning
2.1 Pinning diagram
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Pinning diagram
2.2 Pin description
Table 1.
Pin description
Pin
Pin name
number
Description
1
supply voltage
VCC
start-up
At mains switch-on, an external start-up circuit (usually a passive
resistor network) charges the capacitor connected to this pin.
When VCC exceeds Vstartup (= 22 V typical), the IC wakes up from
power-down mode and checks if all other conditions are met to start
switching.
undervoltage lockout
When the voltage on the pin drops below the 10.5 V (typical; the stop
voltage (Vth(UVLO))), the IC stops switching and restarts
(TEA1832TS) or latches (TEA1832LTS).
burst mode
When no protection is active and the voltage on the pin drops to
below 11.1 V (Vth(burst)), two strokes are initiated with minimal Ipk to
keep the VCC capacitor charged and prevent a restart.
latch reset and clamp
During latched protection, this pin is internally clamped to just above
the 4.5 V latch reset voltage (Vrst(latch)), enabling fast latch reset after
unplugging the mains.
internal overvoltage protection
An internal OVP sets the IC to latched off-state when the voltage on
pin VCC exceeds 36 V (typical) for four consecutive switching
cycles.
2
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GND
ground
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GreenChip TEA1832(L)TS fixed frequency flyback controller
Table 1.
Pin description
Pin
Pin name
number
Description
3
protection input
PROTECT
Two independent protection features are combined on this pin. Both
functions are alternatively activated for 500 s.
mains detection input
During mains detection, the PROTECT pin is clamped to 0.25 V
(no interference with OTP network due to used diode). It measures
the current from a high-ohmic resistor, connected to the mains
electrolytic capacitor. The measured current is used to realize the
brownin and brownout functions and the High/low line compensation
for the overpower protection.
external OverTemperature Protection (OTP)
For OTP, a diode in series with an NTC resistor is connected to
ground. During OTP detection, a current of 200 A flows out the pin
via a diode and NTC to the ground. The voltage is measured. When
it is four consecutive measuring cycles below Vdet(PROTECT)
(2 V typical), the OTP protection is activated.
4
CTRL
power control input
general
The voltage on the CTRL pin controls both the switching frequency
and the peak current.
input configuration
The input is internally connected to 5.4 V via a 26 k resistor.
range
The active range of pin CTRL is from 1.45 V (no load) to 4.5 V
(maximum peak load).
5
ISENSE
current sense input
general
This pin senses the primary coil current across an external resistor. It
compares this coil current to an internal control voltage (Vctrl(Ipeak))
which is proportional to the voltage on the CTRL pin. It switches off
the MOSFET when the level is reached.
propagation delay
The delay time from detecting the level to actually switching off the
driver is approximately 150 ns.
leading-edge blanking
During the first 325 ns of each switching cycle, the ISENSE input is
internally blanked to prevent that spikes, due to parasitic
capacitance, prematurely trigger the peak current comparator.
overpower protection
When the internal Ipeak control voltage (Vctrl(Ipeak)) exceeds 400 mV,
the overpower timer is started. When this condition lasts longer than
27.5 ms (TEA1832TS)/160 ms (TEA1832LTS), the IC triggers a long
restart (TEA1832TS) or enters the latched protection mode
(TEA1832LTS).
overcurrent protection
The internal control voltage is limited to 575 mV, which limits the
primary peak current and thus the input power.
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Table 1.
Pin description
Pin
Pin name
number
Description
Output Short Circuit Protection (OSCP)
To prevent transformer saturation in CCM during start-up, overload
events, or short circuit events, the switching frequency is lowered
when the CTRL voltage exceeds 2.85 V (above nominal power level)
and the measured level on the ISENSE pin is above 400 mV
(Vth(sense)opp) within 1 s after turn-on of the driver output.
high/low line compensation
A current, proportional to the measured current at the mains detect
(see the PROTECT pin) flows out of the ISENSE pin. When
connecting a resistor between the ISENSE pin and the Rsense
resistor, a proportional DC voltage is deducted from the Rsense
voltage generated by Ipeak.The decreased level keeps the OPP level
constant.
overvoltage protection output voltage
During the secondary stroke, the voltage on the auxiliary winding
(which is related to the secondary winding voltage) is sensed using a
diode and a resistor in series connected from the auxiliary winding to
the ISENSE pin. When the measured voltage surpasses 2.5 V for
four consecutive switching cycles, the latched protection is triggered.
soft start
A built-in soft start function slowly enables the primary peak current
to grow.
slope compensation
The amount of slope compensation (related to the ISENSE pin):
18 mV/s. The slope compensation is only active at duty cycles
higher than 45 %.
6
DRIVER
gate driver output for MOSFET
driver capability
The driver can source and sink 0.3 A at 2 V. It can sink 0.75 A at
10 V.
frequency modulation
The switching frequency is modulated over a range of 4 kHz at a
rate of 260 Hz to improve EMI behavior.
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3. Functional description
3.1 General
The TEA1832(L)TS has been designed for fixed-frequency flyback power supplies.
The TEA1832(L)TS uses peak current control. The output voltage is measured and
transferred back via an optocoupler to the CTRL pin.
This chapter describes how the controller works. See Section 4 for specific application
issues.
3.2 Start-up
3.2.1 Charging the VCC capacitor
To provide the start-up power, a resistor charges capacitor C11 (see Figure 1) on the VCC
pin. When VCC is below Vstartup (22 V typical), the IC current consumption is low
(11 A typical). When the capacitor is charged above Vstartup and all other conditions have
been met, the controller starts to switch. When the switching has started, the
TEA1832(L)TS is supplied by the auxiliary winding.
Connect the resistor in front of the bridge rectifier for fast latch reset1.
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1.
VCC pin
The only way to reset the latched protection is to bring the VCC pin below 4.5 V. During latched protection, the supply current is
 15 A. If the start-up resistor is connected after the bridge rectifier, the bulk capacitor continues to feed it for a long time after
unplugging the mains.
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A low-cost and efficient implementation for the start-up circuit is using two resistors in
series with a low voltage diode to L and N. The resistors also discharge the X-capacitor
(CX1) after the mains has been unplugged (see Figure 1). See Section 5 for more
information about the start-up circuit.
3.2.2 Start-up conditions
When the VCC pin reaches Vstartup (22 V typical), the controller wakes up from
power-down mode and checks the PROTECT pin. The PROTECT pin cycles between
mains voltage detection and external OverTemperature Protection (OTP). If during mains
voltage detection the brownin level is not met, switching is not started. The IC waits until
the brownin level is reached. If during OverTemperature Detection (ODT) the level on the
external NTC is too low, switching is not started. The IC waits until the voltage on the
PROTECT pin surpasses the Vdet(PROTECT) level (2 V typical). Due to the waking up, the
supply current increases. When switching does not start, VCC drops to below Vth(UVLO).
The IC enters power-down mode. The start-up circuit charges the VCC capacitor and the
cycle repeats itself.
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3.2.3 Soft start
Only when both brownin and the OTP level are correct, the internal soft start is initiated.
During soft start, the Vctrl(Ipeak) level (level at the ISENSE pin where the MOSFET is
switched off) increases from 0 mV to 575 mV within 3.6 ms. The increase causes the
peak current to increase gradually. The switching frequency is at the maximum (130 kHz).
The purpose of the soft start feature is to avoid audible noise at start-up. Increasing the
peak current instantly from 0 to the maximum is audible.
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3.2.4 Clamp
The 5.4 V clamp on the VCC pin is only active during the latched off-state (see Figure 3).
The purpose of this clamp is to keep the VCC pin just above the 4.5 V latch reset level,
ensuring a fast latch reset after unplugging the mains.
3.3 Power control
3.3.1 General
The CTRL pin controls the amount of output power by changing both the peak current and
the switching frequency (see Figure 5).
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CTRL, ISENSE, and DRIVER pins
3.3.2 Input biasing
An internal 26 k resistor connected to 5.4 V enables direct connection of an optocoupler
transistor. External components to convert the output current of the optocoupler to the
control voltage are not required. The relationship between the current and the voltage on
the CTRL pin can be calculated with Equation 1 (see Figure 6).
3
V CTRL = 5.4 V – 26  10  I O  CTRL 
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GreenChip TEA1832(L)TS fixed frequency flyback controller
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VCTRL as a function of IO(CTRL)
3.3.3 Peak current control
The CTRL pin controls the primary peak current. Figure 7 shows the relationship between
the voltage and the peak current on the CTRL pin.
The DRIVER output is switched on by each oscillator pulse. The voltage on the CTRL pin
controls the oscillator frequency. It is switched off when the primary peak current
measured on the ISENSE pin exceeds the peak current set by the CTRL pin or if the duty
cycle exceeds 90 %.
3.3.4 Frequency control
The voltage on the CTRL pin controls the switching frequency. The frequency curve
(see Figure 7) can be split into three areas:
• Peak power
At peak power, the switching frequency is increased to 130 kHz to enable a higher
output power from the same core. It also increases the switching losses but is
irrelevant during temporary peak loads. For the maximum benefit of the frequency
increase, the supply must operate (mainly) in DCM (in CCM, the frequency increase
does not have much influence).
Peak power can only be delivered during the 27.5 ms (TEA1832TS)/160 ms
(TEA1832LTS) overpower time-out.
• High and medium power
At high and medium power, the switching frequency is fixed to 65 kHz. Only the peak
current is controlled.
• Low power
The peak current is not reduced below 22 % of its maximum value to ensure efficient
operation at low output power. Instead, to reduce the output power, the switching
frequency is reduced. The frequency now enters the audible spectrum but does not
become audible because of the low peak current. This part of the frequency curve is
also referred to as Voltage Controlled Oscillator (VCO) mode.
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GreenChip TEA1832(L)TS fixed frequency flyback controller
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Vctrl(Ipeak) and fsw as a function of VCTRL
It is important to use the entire CTRL pin input range. If the chosen current sense resistor
value is too low, only the lower part of the control curve is used. The frequency reduction
already starts at a relatively high peak current which can result in audible noise.
3.3.5 Burst mode
When the control voltage (VCTRL) is pulled below 1.45 V (typical), the IC is not switching. It
waits until the VCTRL exceeds this minimum level before starting the next cycle. During this
period of no switching, the supply current of the TEA1832(L)TS discharges the VCC
capacitor. When the voltage on the VCC pin drops to below Vth(burst) (11.1 V typical), two
strokes are asserted with minimal Ipk to recharge the VCC capacitor. The assertion avoids
that the voltage on the VCC pin drops to below UVLO level during a longer off-time.
Longer off-times occur when the output switches from peak load to no-load. The output
voltage shows an overshoot and the system holds switching until the output voltage drops
to below the regulation level while there is no load at the output.
The burst mode enables the use of a smaller VCC capacitor, shortening the start-up time
or decreasing the no-load power by using higher value start-up resistors.
The burst mode is only intended to help with load changes. It is not intended for use under
normal no-load conditions. For a minimum no-load power, the system must be in Vout
regulation, not in VCC regulation. The auxiliary winding voltage and the size of the VCC
capacitor must be chosen so that this condition is met.
3.3.6 Switch-off delay
The primary peak current does not immediately stop when it passes the threshold level
because there is some internal and external delay. During this delay, the primary current
still continues to grow. The exact increase depends on the delay, the primary inductance
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and the voltage on the main electrolytic capacitor. The OPP compensation largely
compensates this dependency on the bulk voltage. Due to this switch-off delay, fine-tuning
of the Rsense resistor value may be required to obtain the correct output power level.
The switch-off delay can be split into three delays:
• External filter delay
Resistor R13 (see Figure 5) and the parasitic capacitance of the track (2 pF to 10 pF)
create a delay. For minimal disturbance and capacitance, place R13 as close as
possible to the ISENSE pin. The delay approximately equals R  C (13 ns to 68 ns).
• Propagation delay
The internal delay from passing the threshold level on the ISENSE pin to the actual
switching off the DRIVER pin is approximately 150 ns.
• MOSFET switch-off delay
The MOSFET does not immediately switch off when the DRIVER pin switches off.
The MOSFET switch-off delay is defined as the delay from the moment the voltage on
the DRIVER pin starts to drop until the drain of the MOSFET reaches the bulk voltage.
It must be measured in the application.
3.3.7 Leading-Edge Blanking (LEB)
The ISENSE input is internally blanked for the first 325 ns of each switching cycle. The
blanking prevents that spikes caused by parasitic capacitance (gate-source and
drain-source capacitance of the MOSFET and the parasitic capacitance of the
transformer) trigger the peak current comparator prematurely.
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Leading-edge blanking
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3.4 Protections
3.4.1 Overview
Depending on which protection is triggered and on the version of the IC, the protection
causes a safe restart or latches the converter to an off-state. Table 2 shows an overview
of the protection features.
Table 2.
Protection handling TEA1832(L)TS
Protection
TEA1832TS
TEA1832LTS
Comment
internal VCC OVP
latch
latch
four consecutive switching
cycles
UVLO
restart
latch
maximum duty cycle
restart
restart
eight consecutive switching
cycles
internal OTP
latch
latch
four consecutive switching
cycles
brownin/brownout
restart
restart
external OTP
latch
latch
four consecutive measuring
cycles
external OVP
latch
latch
four consecutive switching
cycles
OPP
slow restart
latch
TEA1832TS:
27.5 ms overpower time-out
(14.7 ms in case of OSCP)
TEA1832LTS:
160 ms overpower time-out
(also in case of OSCP)
OCP
cycle-by-cycle
cycle-by-cycle
OSCP
cycle-by-cycle
cycle-by-cycle
See Section 3.4.2 and Section 3.4.3 for explanation of safe restart and latched off-state.
3.4.2 Protection handling: Restart
3.4.2.1
Regular restart (short)
If one of the protections triggers a restart, the TEA1832(L)TS immediately stops switching
and the supply current of the IC quickly discharges the VCC capacitor. When VCC drops
below the undervoltage lockout level, the IC enters power-down mode. The supply current
drops to approximately 11 A and a normal start-up sequence follows.
3.4.2.2
OPP restart (slow restart, not in TEA1832LTS)
If the OPP triggers a restart, the regular restart delay is insufficient to keep the average
input power below an acceptable level (usually 5 W) in a continuous overload. The
TEA1832TS first carries out a regular restart sequence, but instead of starting up when
VCC reaches 22 V, it does not wake up from power-down mode. Instead it discharges the
VCC capacitor to 10.5 V again and lets the start-up circuit charge the capacitor. It
continues charging and discharging between Vstartup and Vth(UVLO) three times before
starting up (see Figure 4).
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The rising slope of this saw tooth depends on the mains voltage, the start-up circuit and
the VCC capacitor. At low input voltage the restart time increases. An internal current
source (2.5 mA) determines the falling slope of the saw tooth.
3.4.3 Protection handling: Latch
3.4.3.1
Latched off-state
When one of the protection features triggers the latched off-state, the IC immediately
stops switching and enters power-down mode. It clamps the VCC pin to 5.4 V, which is
just above the reset level (4.5 V).
3.4.3.2
Resetting a latched protection
To reset a latched protection, the voltage on the VCC pin must drop to below 4.5 V. A
"power-cycle" of the mains must be done. Unplug the mains, wait a moment and then
reconnect the mains.
If a latched protection is triggered, the VCC pin is automatically clamped to a voltage just
above the reset level. When the mains is unplugged, the start-up current stops and the
11 A supply current to the TEA1832(L)TS discharges the VCC capacitor. Because the
capacitor only is required to discharge from 5.4 V to 4.5 V, it resets quickly.
When capacitor CVCC = 2.2 F, the discharge time is 0.3 s. In practice, the start-up current
does not always immediately stop charging the VCC capacitor after unplugging the mains
because the X-cap can still be charged for about one second.
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Latched protections
3.4.4 Internal VCC OVP
An internal overvoltage protection sets the IC to latched-off state when the voltage on the
VCC pin exceeds 36 V for four consecutive switching cycles (all TEA1832 versions). The
internal OVP measures on the falling edge of the DRIVER signal.
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3.4.5 UnderVoltage LockOut (UVLO)
3.4.5.1
Restart version (TEA1832TS)
When during normal operation the voltage on the VCC pin drops to below the
undervoltage lockout threshold (Vth(UVLO) = 10.5 V typical), the IC stops switching and
enters power-down mode. The start-up circuit charges the VCC capacitor and a normal
start-up sequence follows.
3.4.5.2
Latch version (TEA1832LTS)
When during normal operation VCC drops below the undervoltage lockout threshold, the
IC is set to the latched protection mode. It ensures that a shorted output always triggers
the latched protection mode, including if VCC drops to below Vth(UVLO) before the OPP has
a chance to respond.
3.4.6 Maximum duty cycle protection
The main purpose of the maximum duty cycle protection is to ensure a well-defined
response to mains supply dips.
If the peak current measured by the ISENSE pin does not reach the Vctrl(Ipeak) level set by
the CTRL pin within a duty cycle of 90 %, the maximum duty cycle limitation ends the
driver pulse. If the maximum duty cycle is exceeded during eight consecutive switching
cycles, the maximum duty cycle protection triggers a restart.
In low-power mode, the switching frequency can become very low. Limiting the duty cycle
to 90 % is insufficient to prevent a long on-time. In low-power mode, the maximum duty
cycle limitation changes to maximum on-time limitation. The maximum on-time is 14 s.
3.4.7 Internal OverTemperature Protection (OTP)
When the temperature in the chip exceeds 140 C, the internal OTP sets the controller to
the latched off-state (in all TEA1832 versions).
3.4.8 Brownin/brownout and external OTP protection (PROTECT pin)
The PROTECT pin cycles between mains detection measurement used for
brownin/brownout and high/low line compensation and external OTP. Each measurement
lasts 500 s.
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3.4.8.1
Brownin
Figure 16 shows the circuit.
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The components, related to the external OTP, are grayed out but present for reference.
Fig 10. PROTECT pin: Mains detection function
The mains detection function determines the bulk voltage on the mains electrolytic
capacitor by measuring the current through a 20 M resistor connected to the PROTECT
pin. During mains detection measurement, the PROTECT pin is clamped to 0.25 V
ensuring that diode D6 blocks the influence of the connected external OTP circuit. The
measured mains detection current is stored for reference.
To allow the start-up of the IC when VCC reaches Vstartup (22 V typical), the mains
detection current must exceed 5.7 A. It correlates to a bulk voltage of
5.7 A  20 M = 114 V (DC). Because there is no load yet, the bulk voltage equals the
peak voltage of the mains minus 2 diodes, which yield for the AC voltage:
 114 + 1.4   2 = 82 V .
3.4.8.2
Brownout
The mains detection current is also used to realize the brownout function. When the mains
detect current drops to below 5 A for 32 ms, switching is stopped and a restart is made
for both the latched and the non-latched versions.
Each time the mains detection current exceeds 5 A, the 32 ms counter is reset. The
counter-reset ensures that only the peak of the mains is measured and the brownout level
is independent of the load (see Figure 11).
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Fig 11. Bulk voltage versus brownout detection
At 90 V (AC), the valley of the mains ripple drops to below 5 A. However, each peak
exceeding 5 A resets the 32 ms counter. Only when the peak of the mains detection
current remains < 5 A, the protection is triggered.
The brownout level of 5 A corresponds with a bulk voltage of
5 A  20 M = 100 V  DC  . Because this level corresponds with the peak of the AC
input voltage, we get:  100 + 1.4   2 = 72 V  AC 
3.4.8.3
High/low line compensation
In fixed-frequency DCM, peak current limitation can also act as overpower protection
because the maximum output power is independent of the input voltage. In
fixed-frequency CCM, the maximum amount of power that can be transferred to the output
not only depends on the primary peak current. It also depends on the duty cycle and
therefore on the input voltage.
To obtain the same overpower protection level over the mains input range, high/low line
compensation adjusts the peak current level as a function of the mains input voltage.
The high/low line compensation is realized by forcing a current Iopc (overpower correction)
out of the ISENSE pin (see Figure 12).
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Fig 12. High/low line compensation ISENSE pin
The current Iopc is related to the measured mains detection current. It causes a voltage
drop over resistor Ropc between the ISENSE pin and Rsense. The higher Iopc, the earlier
the required Vctrl(Ipeak) is reached at the ISENSE pin.
Figure 13 shows the relationship between the mains detection current and Iopc.
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Fig 13. Relationship between Idet(mains) and Iopc
The compensation starts from Idet(mains) = 6.24 A which corresponds with 125 V (DC)
bulk voltage. For 365 V bulk voltage, Idet(mains) = 18.24 A and Iopc = 6 A. For
Idet(mains) > 6.24 A: I opc = 0.50   I det  mains  – 6.24  A .
The value of resistor Ropc determines the level of compensation. With the Ropc value used
in the schematic (6.8 k; see Figure 12), the correction of the voltage on the ISENSE pin
is 41 mV at 365 V (DC) bulk voltage (264 V (AC)), which equals about 10 % of the
400 mV Vth(sense)opp.
At low output power (Vctrl(Ipeak) < 350 mV), the high/low line compensation is switched off.
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3.4.8.4
External OverTemperature Protection (OTP)
The external OverTemperature measurement is combined with the mains detection
function on the PROTECT pin (see Section 3.4.8). The PROTECT pin cycles between
mains detect (500 s) and external OverTemperature measurement (500 s). Figure 14
show the circuit. The components for mains detection are grayed out but present for
reference.
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Fig 14. External OTP circuit
During temperature measurement, a 200 A current flows out of the PROTECT pin
through the connected diode in series with an NTC to ground. The additional fixed series
resistor can be used to tune the temperature trigger point of the detection. The latched
protection is triggered when, during the 500 s measurement, the voltage on the
PROTECT pin remains below 2.0 V for four consecutive measurement cycles (see
Figure 15). At trigger point, the voltage over the NTC plus optional series resistor equals
2.0 V minus the forward voltage of the diode: V NTC + V Rseries = 2.0 – 0.55 = 1.45 V .
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Fig 15. Voltage on the PROTECT pin
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The mains detection measurement does not disturb the temperature measurement. The
current through the 20 M resistor, measured during mains detection, is stored and
subtracted from the 200 A current during the OTP measurement.
Two remarks regarding diode D6:
• The forward voltage of diode D6 is temperature-dependent. However, it does not
influence the detection accuracy. It behaves as an additional temperature sensor with
a negative temperature coefficient of 2 mV/K.
• To minimize the error of the mains detect current measurement, diode D6 must be a
low leakage type. The NXP Semiconductors BAS416 is suited for this purpose.
3.4.8.5
Clamp
An internal clamp prevents that the voltage level at the pin exceeds 4.1 V (typical) when a
high ohmic NTC is used. At start-up, when the pin is not active yet, the divider of the
mains detection resistor and the NTC can cause a voltage level that is too high at high
mains when a high ohmic NTC is used. The clamp voltage is specified at a 200 A input
current (the exact voltage depends on the current). In power-down mode, the clamp
voltage drops to approximately 2.0 V.
3.4.9 External output OverVoltage Protection (OVP)
The purpose of the overvoltage protection is to protect the devices connected to the
output. It also protects the supply itself against output voltages that are too high, for
example, when the voltage feedback loop is disturbed.
The output voltage is measured via the auxiliary winding using a diode in series with a
resistor connected to the ISENSE pin (see Figure 16).
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Fig 16. External output OVP circuit
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Resistors Rovp, Ropc, and Rsense form a resistive divider. The voltage on the ISENSE pin is
measured during the secondary stroke. When the voltage exceeds 2.5 V for four
consecutive switching cycles, the controller is set to the latched-off state.
The first 2 s of the secondary stroke are discarded to avoid incorrect level measurement
due to ringing (see Figure 17).
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Fig 17. Voltage on the ISENSE pin
The Vout OVP measurement can be combined with the peak current measurement
because the branch with resistor Rovp is inactive during the primary stroke. The branch
with resistor Rovp is inactive during the primary stroke because of the added diode and the
negative voltage on the auxiliary winding.
At the start of the primary stroke, a negative spike is present on the ISENSE pin. The
reverse recovery of the diode causes the spike. The internal ESD diode of the ISENSE
pin, which can handle 10 mA, clamps the spike.
3.4.10 OverPower Protection (OPP)
The maximum nominal output power is reached when the internal control voltage
Vctrl(Ipeak) exceeds the overpower level (400 mV typical) while switching at 65 kHz
(see Figure 18).
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Fig 18. Overpower protection and compensation
When Vctrl(Ipeak) exceeds 400 mV, the OPP timer is started. When the maximum nominal
output power is continuously exceeded for the overpower time-out duration (see Table 3),
the OverPower Protection (OPP) is activated. The controller immediately stops switching
and performs a slow restart (TEA1832TS) or enters the latched-off state (TEA1832LTS).
Table 3.
Overpower time-out
Vout level
TEA1832TS
TEA1832LTS
Vout > 0.5 Vout(ovp)[1]
27.5 ms
160 ms
Vout < 0.5 Vout(ovp)[1]
14.5 ms
160 ms
[1]
Vout(ovp) is the output voltage at which the external OVP triggers.
During a slow restart, the IC cycles the voltage on the VCC pin three times between
Vth(UVLO) and Vstartup before switching begins. It reduces the input power for continuous
overload due to the lower repetition rate of the switching cycles (see Section 3.4.2.2).
When Vctrl(Ipeak) drops to below 400 mV before the OPP timer reaches the overpower
time-out, the timer is immediately reset. The reset enables (repeated) temporary
overloads (see Figure 19).
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Fig 19. Overpower time-out
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The OPP time-out is reduced during heavy overload or short circuit events, where the
output voltage drops to below half the OVP level of Vout (TEA1832TS only). Reducing the
OPP time-out limits the input power during slow restart in continuous overload condition.
Details are described in Section 3.4.13.
The OPP time-out is realized by dividing the internal jitter clock. Because an occurrence of
an OPP condition is not synchronized with the internal jitter clock, the OPP time-out varies
with one clock cycle of the internal jitter clock (3.8 ms).
The TEA1832(L)TS incorporates a built-in high/low line compensation to ensure that the
OPP level is constant over the full mains input range. This feature is described in
Section 3.4.8.
3.4.10.1
Slow restart delay timing calculation (TEA1832TS only)
When the OPP triggers a safe restart procedure, the TEA1832TS immediately stops
switching, discharges VCC to below Vth(UVLO) and enters power-down mode.
The start-up circuit charges the VCC capacitor to Vstartup but instead of starting up, the
VCC capacitor is discharged again. To obtain a longer restart time, the
charging/discharging is repeated for three cycles.
The restart time of one restart cycle consists of two periods:
• Discharging the VCC capacitor by an internal current source (2.5 mA). The discharge
time is independent of the mains voltage2.
• Charging the capacitor from Vth(UVLO) to Vstartup by the external start-up circuit of
typically 81 A at 264 V (AC).
The discharge current is much higher than the charge current. So the mains voltage and
the start-up circuit mainly determine the restart time.
The discharge time is approximately:
C VCC   V startup – V th  UVLO  
2.3 F   22 V – 10.5 V 
t dch = ------------------------------------------------------------------------- = -------------------------------------------------------------- = 10 ms
I CC
2.5 mA
(2)
Where:
• CVCC is the total capacitance on the VCC pin
• ICC(restart) is the current discharged by the internal current source
Remark: The first discharge time differs from the one calculated in Equation 2. The
starting voltage can be higher or lower (depending on the load) and the discharge current
source is not yet switched on. The operating supply current (= 0.58 mA) discharges the
VCC capacitor.
The charge time is pending on the value of the start-up resistors. Worst case (shortest
charge time) occurs for the highest mains voltage.
The charge current at 264 V (AC) and Rstartup = 2.4 M becomes
2.
The first discharge differs somewhat from the following ones. The starting voltage can be lower or higher (depending on the load)
and the discharge current source is not yet switched on. The operating supply current (= 0.58 mA) discharges The VCC capacitor.
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I ch
V startup + V th  UVLO 
2
---  2  V mains – -----------------------------------------------
2
= ---------------------------------------------------------------------------------------------- – I CC  startup 
R1
(3)
22 V + 10.5 V
0.64  1.41  264 V  AC  – ---------------------------------2
= -------------------------------------------------------------------------------------------------------- – 11 A = 81 A
2.4 M
Where:
• Vmains is the RMS input voltage
The charge time becomes:
C VCC   V startup – V th  UVLO  
2.3 F   22 V – 10.5 V 
t ch = ------------------------------------------------------------------------- = -------------------------------------------------------------- = 0.32 s
I ch
81 A
(4)
Where:
• Ich is the charge current into the VCC capacitor (see Equation 3)
The total restart delay for the start-up circuit with two resistors and diodes approximately
equals:
t restart = 3   t dch + t ch  = 3   0.01 s + 0.32 s  = 0.99 s at V mains = 264 V
3.4.10.2
(5)
Ratio OPP time-out/restart delay
In a continuous overload, the supply keeps switching on and off (only valid for the
non-latched version). At high mains, the ratio of the on-time and off-time is approximately
1:31 for the 27.5 ms OPP timer. It is sufficient to keep the average input power during a
continuous overload below 5 W in most applications.
The average input power during a continuous overload at maximum peak power is:
P o  max 
t opp
27.5 ms
130 W
P i  AV  =  -------------------------------   ------------------ =  --------------------------------------------  ---------------- = 4.15 W
 t opp + t restart   
 27.5 ms + 930 ms
0.9
(6)
at V mains = 264 V  AC 
Where:
• Pi(AV) is the average input power during an overload
• topp is the overpower protection time-out time, the time from exceeding the overpower
threshold to triggering the protection. This time is fixed to 27.5 ms in the IC.
• trestart is the restart delay, the time from triggering the protection until the next start-up
attempt
• Po(max) is the maximum peak output power; the maximum power that the supply can
deliver during the overpower protection time-out time.
For a short circuit at the output, the OSCP protection shortens the maximum OPP time to
14.5 ms. The delivered output power is also reduced due to the lower output voltage and
the lower switching frequency when OSCP is triggered. The shortening of the OPP time
leads to a reduction of the Pi(AV) by a factor 4.
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During an output short Pi(AV)  1.05 W at 264 V (AC).
In practice, measurements < 1 W at 264 V (AC) were done in a 65 W demo board.
The restart delay depends on the input voltage, the start-up circuit and the VCC capacitor.
The restart delay can be changed by changing the values of the start-up circuit or the VCC
capacitor. However, changing the values of the start-up circuit of the VCC capacitor also
influences the start-up time.
The input power at continuous overload can be decreased in the following way:
1. Increase the restart delay by increasing the start-up resistors or capacitor CVCC.
2. Decrease the maximum output power by increasing the current sense resistor
(if possible).
3.4.11 OverCurrent Protection (OCP)
To prevent saturation of the transformer and so currents in the MOSFET that are too high,
a cycle-by-cycle primary inductor current limitation is built in.
When the voltage on the ISENSE pin exceeds 575 mV, the current switching cycle is
immediately ended. When the OCP limits the peak current, the output voltage can no
longer be maintained. The converter continues to switch until the OPP is triggered or until
VCC has dropped to below Vth(UVLO) (10.5 V typical).
3.4.12 Temporary peak power
It is possible to deliver a peak power of 200 % for a short time. To keep the output voltage
within regulation and prevent that protections are triggered, the following conditions must
be met:
• The voltage on the ISENSE pin must remain < 575 mV (to prevent that OCP is
triggered and limits the power)
• The duration must be shorter than the OPP time-out (to avoid the triggering of OPP)
To deliver the power, the switching frequency increases from 65 kHz to 130 kHz. The
voltage on the ISENSE pin increases from 400 mV to 575 mV.
The actual peak power which can be delivered depends on the input voltage, the size of
the mains electrolytic capacitor, and the duration of the peak power. The high/low line
compensation for the input voltage is also active during peak power. However, the
correction is optimized for OPP. For low input voltage, the mains electrolytic capacitor and
the duration determine the maximum peak power. Figure 20 shows the peak power of a
65 W converter.
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DDD
3SHDN
:
9PDLQV9506
Fig 20. Peak power as a function of mains voltage
The duration of the peak power is 2 ms. The repetition time is 50 ms. The peak power is
defined as the level where Vout just starts dipping (just triggering OCP).
3.4.13 Output Short Circuit Protection (OSCP)
The OSCP prevents saturation of the transformer during continuous overload or shorted
output. Transformer saturation can cause runaway of the input power and the peak
current. The OSCP lowers the switching frequency and shortens the duration of the OPP
timer from 27.5 ms to 14.5 ms (typical).
To clarify the phenomenon, a simplified circuit is used. In the simplified circuit, a
transformer with a turn ratio of 1:1 is used because then the start value and the peak
value of the primary and secondary currents are the same (see Figure 21).
Q
,SULP
,SULP
,VHF
9VHF
/3
5ORDG
,VHF
FORVHG
Q5ORDG
VZLWFK
,SULP
VZLWFK
Q9VHF
/3
VZLWFK
,SULP
,VHF
,VHFQ
,VHF
,SULP
,VHFQ
,SULP
RSHQ
FORVHG
RSHQ
VZLWFK
,SULP
,VHFQ
,SULP
,VHFQ
FORVHG
RSHQ
FORVHG
RSHQ
DDD
Fig 21. Simplified flyback circuit
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The currents are drawn for CCM. The primary current starts with an offset equal to the
value of Isec/n at the moment the primary stroke is initiated. The primary current increases
until the drive is switched off. The secondary current Isec/n starts with the value of the
primary current at switch-off and decreases until the next primary stroke is initiated. The
transfer of the transformer winding ratio from n:1 to 1:1 means that the secondary current
is divided by the turn ratio n. The output voltage is multiplied with the turns ratio n.
The simplified flyback circuit is used to explain the saturation/runaway problem. When the
circuit is in balance, the current increase during the primary stroke equals the current
decrease during secondary stroke. If the output is short-circuited, VO (and also nVO)
becomes low. The decay of the secondary current Isec/n also becomes low because it is
proportional to VO (Idecrease = (n  VO / Lp)  ts.
In CCM, the switching time is fixed. When the current increase during the primary stroke
exceeds the decrease during the secondary stroke, the DC offset part of the current
increases at every stroke. This effect is called runaway, because it is not possible to reach
a balanced state (see Figure 22).
,SULP
,VHFQ
VZLWFK
,SULP
FORVHG
,VHFQ
RSHQ
,SULP
FORVHG
,VHFQ
RSHQ
,SULP
FORVHG
,VHFQ
RSHQ
,SULP
FORVHG
,VHFQ
RSHQ
DDD
Fig 22. Runaway effect
Because the maximum flux density B of the core is proportional to the primary current
(B = Lp  Ip / Np  Ae), the core flux density also increases. The transformer finally
saturates and Lp reduces dramatically. The primary current can increase to very high
levels.
To avoid runaway, the current decrease during the secondary stroke must be enlarged.
The only available option is to enlarge the secondary stroke time ts by lowering the
switching frequency. The increased secondary stroke time allows the secondary current to
decrease to zero before the next primary stroke is initiated. It also reduces the DC offset
at the start of the primary stroke to zero which stops the runaway. The OSCP initiates the
lower switching frequency when activated. (see Figure 23).
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,SULP
,VHFQ
,VHFEXLOWGRZQWR]HUR
—V
26&3LQFUHDVHVWSHULRGWR—V
—V
26&3LQFUHDVHVWSHULRGWR—V
9,6(16(
9FWUO,SHDN
—V
26&3GHWHFWHG
DDD
Fig 23. OSCP cycle-by-cycle protection
To detect if the risk of runaway is present and the OSCP must be activated, the voltage
level at the ISENSE pin is monitored. When the voltage, reflecting the primary peak
current, is too high within 1 s after switch-on of the drive, the OSCP is activated. The
exact conditions are:
• The Vctrl(Ipeak) level must be > 400 mV (OPP condition). The typical control voltage
(Vctrl) on the CTRL pin > 2.85 V.
• The measured voltage level on the ISENSE pin must be higher than the Vctrl(Ipeak)
level after 1 s. Depending on the voltage on the CTRL pin, VCTRL(Ipeak) ranges from
400 mV to 575 mV.
When these conditions are met, the next stroke is delayed with a factor 4 (31 s instead of
7.7 s). This delay is repeated as long as the above conditions exist. If the voltage on the
ISENSE pin is < Vctrl(Ipeak) after 1 s or Vctrl(Ipeak) drops to below the OPP condition,
normal switching is resumed.
Because the runaway problem only happens when the output voltage is far below its
nominal value, the OSCP is only enabled if the output voltage is lower than half the Vout
OVP value. The Vout level is detected in a similar way as the Vout OVP (see Figure 24).
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%$6:
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Fig 24.
Detection 0.5 Vout OVP level
To lower the average input power further during a short circuit event, the OPP timer is
reduced from 27.5 ms to 14.7 ms when OSCP is enabled (TEA1832TS only).
The OSCP prevents overstress during start-up. Because the output capacitors are
discharged, they behave as a short circuit until the output voltage starts to increase after
the first strokes.
3.5 Slope compensation
The TEA1832(L)TS has built-in slope compensation to prevent subharmonic oscillation in
CCM at duty cycles above 50 %. The slope compensation is internally added to the CTRL
input signal (see Figure 25). The slope compensation is 18 mV/s referred to the ISENSE
pin. The slope compensation is only active on duty cycles higher than 45 %.
oscillator
slope
compensation
0
45
75 100
t / T (%)
019aaa170
Fig 25. Slope compensation waveforms
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3.6 Driver
The driver circuit has a current sourcing capability of 300 mA typical and a current sink
capability of 750 mA typical. These capabilities permit a fast turn-on and turn-off of the
power MOSFET for efficient operation. See Figure 5 for the DRIVER pin control.
3.7 Frequency modulation
The switching frequency and its harmonics are responsible for a large part of the
conducted EMI problems. Modulation of the switching frequency spreads all frequency
peaks that are related to the switching frequency over wider bands, significantly
decreasing the so-called "average measurement". See Figure 5 for the location of
oscillator and frequency modulation.
The oscillator is continuously modulated at a rate of 260 Hz and a range of 6 %
(4 kHz at 65 kHz). For example, the third harmonic of 65 kHz is spread over a frequency
band of 3  8 = 24 kHz.
IVZN+]
7 +]
W
DDD
Fig 26. Frequency modulation
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4. Application (pin-by-pin)
4.1 VCC pin
4.1.1 Start-up circuit
A low-cost start-up circuit that caters for the X-cap discharge is the two-resistor-+-diode
start-up circuit. The added diode improves the start-up time. See Section 5 for a full
explanation of the start-up circuit.
4.1.2 VCC capacitor
The VCC capacitor must be as small as possible to make the start-up time (and the latch
reset time) as short as possible.
First of all, the value of the capacitor must be sufficient to supply the TEA1832(L)TS until
the auxiliary winding takes over. It depends on the soft start time, the load on the output
and the values of the secondary capacitors.
Other factors also determine the minimum value of the capacitor. Some worst case tests
to determine the minimum value of the VCC capacitor are:
• No-load operation
The supply runs at low frequency so there is a long interval between two consecutive
charge pulses from the auxiliary winding. To prevent that the IC runs on VCC
regulation instead of Vout regulation, the lowest level of VCC must remain 2 V above
Vth(burst).
• Transient from full load to no-load
A transient from full load to no-load can cause a small overshoot on the output
voltage. It can take a long time for the output capacitor to discharge to the level at
which the supply starts to switch again because of the absence of any external load.
During this time, the auxiliary winding does not charge the VCC capacitor. Because
the implemented burst mode which asserts two strokes with minimum Ipk to recharge
the VCC capacitor when VCC drops to below Vth(burst) overcomes this problem, this
condition puts no requirement on the value of the VCC capacitor.
• Low ESR type
The VCC capacitor must be a low Equivalent Series Resistance (ESR) type. Under
low load or no load conditions, the switching frequency can be very low, for example,
250 Hz. The auxiliary voltage, which can have a duty cycle of only 0.1 %, must charge
the VCC capacitor. It is only present for a few s per cycle and the time between two
cycles can be a few ms. So, the VCC capacitor charge current at no load is roughly
equal to the current consumption of the IC multiplied by 1000: 1000  0.5 mA = 0.5 A .
Use a low ESR capacitor because the relatively high ESR of a regular electrolytic
capacitor limits the charge current and can cause the VCC to drop to below Vth(burst). If
VCC drops to below Vth(burst), the IC runs on VCC regulation instead of Vout regulation
increasing the no-load input power.
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4.1.3 Auxiliary winding or take-over winding
The optimal voltage of the auxiliary winding is approximately 20 V:
• Not much lower because during a no-load operation it has to be able to keep the VCC
above the Vth(burst). To retain some margin for disturbances and component spread,
keep a margin of 2 V above the maximum Vth(burst) value.
• Not much higher because at maximum load the VCC must not trigger the OVP.
The series inductor L1 serves two purposes:
• To ensure that the VCC capacitor is charged evenly at very low switching frequencies
during no-load. The high recharge current of the VCC capacitor can lead to loop
disturbance and irregular switching in the no-load condition. L1 smoothes the charge
current of the VCC capacitor which stabilizes the behavior.
• To prevent that the ringing at the start of the secondary stroke is rectified. Ringing can
charge the VCC capacitor to above the OVP level.
L1 is not always required. It depends on the coupling between the windings. Sometimes a
resistor of approximately 4.7  is good enough.
4.1.4 Maximum start-up current
The maximum current that can be supplied by the start-up circuit must not exceed 1 mA.
The start-up circuit must not be able to deliver more than 1 mA at maximum mains
voltage. During latched protection, VCC is clamped to 5.4 V. The clamp can sink at least
1 mA over the entire operating temperature range. Above 1 mA, the clamp holding VCC
below its maximum rating cannot be guaranteed3. The minimum Rstartup value is 470 k.
If a charge current higher than 1 mA is required, a 33 V Zener diode from the VCC pin to
the GND pin can be used to protect the IC. In that case, the internal OVP cannot work
anymore because the Zener keeps the VCC below 33 V. An external OVP at a slightly
lower voltage, for example 30 V, must replace the internal OVP.
4.1.5 PCB layout
The VCC pin must not be treated as a small signal pin because the relatively high current
to charge the gate of the MOSFET also passes through this pin. Keep the loop from the
positive terminal of the VCC capacitor to the VCC pin, via the DRIVER pin, to the gate of
the MOSFET, and via the source of the MOSFET back to the negative terminal of the VCC
capacitor, as small as possible.
4.2 CTRL pin
A capacitor of 1 nF on the CTRL pin is a good value for most applications.
If the capacitor is much larger, it responds too slowly to load transients.
If the capacitor is much smaller, the control loop can become unstable and sensitive to
disturbances.
3.
Above a certain current, the clamp behaves like a current source. The voltage increases and the current remains constant.
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In the layout, treat both the ground and the collector connections of the optocoupler to the
CTRL pin as small signal tracks. To minimize disturbance, avoid additional currents
through these tracks.
4.3 DRIVER pin
The source and sink currents on the DRIVER pin can be high. The source current also
rushes through pin VCC and the sink current through the GND pin. The PCB layout
requires extra attention to avoid large loops.
4.4 ISENSE pin
4.4.1 General application recommendations
Place Rovp and Ropc (see Figure 24) close to the ISENSE pin. It helps to filter out
disturbances.
Do not place a capacitor directly on the ISENSE pin. The voltage during the secondary
stroke can be as high as 2.5 V. However, the voltage during the primary stroke is below
500 mV. So, at the end of the secondary stroke, the voltage on the ISENSE pin must drop
quickly. A capacitor slows this discharge down. It also causes a delay in the peak current
measurement (much more than in an application with TEA1733/TEA1738 because Rocp is
not bypassed by a capacitor). Extra delay is unwanted because it has the opposite effect
as the high/low line compensation. Place Ropc and Rovp close to the IC to keep the
parasitic capacitance on the ISENSE pin low.
4.4.2 Configuration order
The ISENSE pin combines three functions on one pin. To avoid unnecessary iterations,
these functions must be configured in the right order.
1. At low mains, adjust RISENSE to obtain the required OPP level.
2. At high mains, adjust Ropc to obtain the required amount of high/low line
compensation
3. To obtain the required OVP level, adjust Rovp
Remark: Changing brownin/brownout resistor also changes high/low line compensation.
So, when changing brownin/brownout resistor, the high/low line compensation must also
be changed.
4.4.3 Configuring the current sense resistor
Calculate the maximum primary peak current before the correct value of the current sense
resistor is calculated (Equation 7 or Equation 8).
In DCM:
I peak  max  =
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2  P o  max 
--------------------------  L  f sw
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In CCM:
V i  NV o
P o  max  V i + NV o
1
I peak  max  = ------------------  ---------------------- + -------------------------  ---------------------
V i  NV o 2  L  f sw V i + NV o
(8)
Where:
• Ipeak(max) is the maximum continuous primary peak current, the peak current at which
the OPP triggers
• Po(max) is the maximum continuous output power, the output power at which the OPP
triggers
•  is the expected efficiency of the flyback at maximum output power
• Vi is the minimum rectified mains voltage (= 2  the minimum mains voltage) at
which the supply must be able to deliver the maximum continuous output power4.
• N is the winding ratio of the transformer
• Vo is the output voltage
• fsw is the switching frequency, in this case 65 kHz
Now the (maximum) current sense resistor value can be calculated with Equation 9:
V th  sense opp
400 mV
R ISENSE = ----------------------------- = ----------------------I peak  max 
I peak  max 
(9)
Where:
• Vth(sense)opp is the overpower protection threshold voltage on the ISENSE pin
• Ipeak(max) is the maximum continuous primary peak current, the peak current at which
the OPP triggers
Another way to determine the correct value for the sense resistor is by trial and error:
1. Connect a load to the output.
2. Apply the minimum mains voltage at which the supply must be able to deliver the
maximum continuous output power.
3. Increase the load to maximum nominal power. If the OPP kicks in before, decrease
the value of the Rsense resistor until the nominal power can be delivered. If OPP does
not kick in, increase the value of the Rsense resistor until the OPP is triggered.
4. Finally, decrease the value of the Rsense resistor by about 5 % to keep some margin
for normal operation.
The sense resistor must be accurate (1 %) and have a low inductance. Splitting the sense
resistance into two or three parallel resistors not only makes it easier to implement the
desired resistance and power rating but it also helps to reduce the inductance.
4.
The peak current is larger during the valley of the mains ripple. During normal operation, the OPP threshold can already be
exceeded during the valleys of the ripple on the bulk capacitor voltage. However, it does not trigger the OPP. As long as the
threshold is not exceeded during the tops of the bulk capacitor ripple voltage, the OPP time-out counter is reset each 8.33 ms or
10 ms. It never reaches the 27.5 ms (TEA1832TS)/160 ms (TEA1832LTS) OPP time-out. The OPP can only trigger if the OPP
threshold is exceeded throughout the entire mains cycle.
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4.4.4 Calculating the maximum temporary peak output power
The maximum instantaneous primary peak current can now be calculated with
Equation 10:
V ctrl  Ipeak max
575 mV
I peak  M   max  = ---------------------------------- = -------------------R ISENSE
R ISENSE
(10)
Where:
• Vctrl(Ipeak)max is the maximum peak control voltage on the ISENSE pin
Now the maximum temporary peak output power can be calculated5.
In DCM:
1
2
P o  M   max  =   ---  L   I peak  M   max    f sw
2
(11)
In CCM:
V i  NV o
V i  NV o
P o  M   max  =   ----------------------   I peak  M   max  – ----------------------------------------------------------

V i + NV o
2  L  f sw   V i + NV o 
(12)
Where:
•  is the expected efficiency of the flyback at maximum output power
• Ipeak(M)(max) is the maximum instantaneous primary peak current, limited by the OCP
• fsw is the switching frequency, in this case 130 kHz, the "peak power" area of the
frequency curve, see Figure 7
• Vi is the value of the rectified mains voltage during the valley of the ripple
This value is the maximum temporary peak output power at which the output voltage
remains intact.
If the temporary peak output power is not high enough, it can only be increased by
decreasing the current sense resistor value. Decreasing the current sense resistor value
also increases the maximum continuous output power.
4.4.5 Tuning the OPP compensation (Ropc)
The value of Ropc determines the amount of overpower compensation
(see Section 3.4.8.3 and Figure 12). The ISENSE pin generates a current Iopc from 0 A
to 6 A. This current depends on Vbulk of the mains capacitor. The voltage drop over Ropc
reduces the actual peak level over resistor Rsense at switch-off.
The amount of compensation required depends on the value of the current sense resistor.
So the current sense resistor must be chosen first.
Steps for trimming the overpower compensation:
1. Start with a 6.8 k for resistor Ropc.
5.
Calculating the maximum temporary output power is complicated because it depends on the mains ripple on the bulk capacitor,
which itself depends on the output power.
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2. Apply the minimum mains voltage at which the supply must be able to operate.
3. Measure the maximum output power by slowly increasing the load until the OPP is
triggered.
4. Apply the maximum mains voltage at which the supply must be able to operate.
5. Measure the maximum output power by slowly increasing the load until the OPP is
triggered.
6. Increase the Ropc value if the maximum output power at high input voltage is higher
than the maximum output power at low input voltage (not sufficiently compensated).
Decrease the Ropc value if the maximum output power at high input voltage is lower
than the maximum output power at low input voltage (overcompensated).
7. Repeat steps 2 to 6 until the maximum output power at high and low input voltage is
equal.
Remarks:
• The output power as a function of the input voltage is not a linear function. When the
maximum output power has been tuned to be equal for both the absolute minimum
input voltage (90 V (AC)) and the absolute maximum input voltage (264 V (AC)), the
actual maximum output power is slightly higher between these limits.
• The external overvoltage protection also depends on Ropc. So, after changing Ropc,
adjusting Rovp is also required (see Section 4.4.6)
4.4.6 Calculating the effect of the OPP compensation
The resulting peak current reduction of the OPP compensation can be calculated with
Equation 13:
 I opc  R opc 
I peak = -----------------------------R sense
(13)
Where:
• Ipeak is the peak current reduction
• Rsense is the value of the current sense resistor (R11 in Figure 1)
Section 4.4.3 describes how to calculate the peak current and the resulting output power
without input voltage compensation. Ipeak must be subtracted from the peak current
before calculating the maximum output power. This subtraction is required to calculate the
output power with input voltage compensation.
4.4.7 Disabling the overpower protection
Unlike in the TEA1733 series or the TEA1738 series, the overpower protection in the
TEA1832(L)TS cannot be disabled because it is fully integrated in the IC. If the internal
OPP is not required because, for example, a secondary IC handles the overpower
protection, the internal OPP level can be shifted to a higher power by lowering the current
sense resistor. Do not decrease the Rsense value too much because:
• Controlling a relatively high power using only a fraction of the CTRL range results in a
very high gain and can cause instability.
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• Skipping part of the peak current reduction means that it is not sufficiently reduced
before entering the VCO mode. It can cause audible noise. Normally, the peak current
is first reduced by a factor 3 before the frequency enters the audible spectrum.
4.4.8 Calculating the OVP
The resistor for adjusting the OVP can be calculated with Equation 14:
R ovp
N aux
 ----------   V out  ovp  + V F  sec   – V F  aux  
 N sec

= R ocp   ----------------------------------------------------------------------------------------- – 1
V ovp  ISENSE 




(14)
Where:
•
•
•
•
•
Vout(ovp) is the output voltage at which the OVP should be triggered
VF(sec) is the forward voltage of the secondary diode
VF(aux) is the forward voltage of Dovp
See Figure 24 for Rovp and Ropc
Vovp(ISENSE) is the OVP detection level of the ISENSE pin (= 2.5 V)
Example:
•
•
•
•
Ropc = 6.8 k
Vout(ovp) = 24 V
VF(sec) = VF(aux) = 0.6 V
Naux = Nsec = 8 turns
The result: R ovp
AN11588
Application note
8 turns
 ------------------   24 V + 0.6 V  – 0.6 V 
8
turns

= 6.8 k  ------------------------------------------------------------------------------ – 1 = 58.5 k

2.5 V


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4.5 PROTECT pin
4.5.1 Mains detection
To measure Vbulk of the mains electrolytic capacitor, connect a high-ohmic resistor
(20 M) from the positive terminal of C1 to the pin. The voltage at C1 ranges up to
375 V (DC). Because the maximum voltage over 1206 resistors is 200 V (DC), it is best to
place two resistors of 10 M in series. To avoid disturbance pickup, place the resistors as
close as possible to the PROTECT pin. During the mains detection measurement, the
PROTECT pin is clamped to 260 mV (typical).
To minimize the error in the measured mains detection current, select a low reverse
leakage type6 for diode D6. The NXP BAS416 is suited for this purpose.
4.5.2 External OverTemperature Protection (OTP)
The external OTP function is designed for use with a 100 k NTC. The NTC is connected
to ground and via a series resistor + a low leakage diode to the PROTECT pin. To filter
disturbance a 1 nF capacitor is connected from the cathode of the diode to ground. During
the measurement, 200 A current flows out of the pin. OTP is detected when the level on
the PROTECT pin during the measurement remains below 2 V (typical). The voltage over
the NTC + series resistor is then 1.45 V, assuming a 0.55 V drop over the diode in series.
This voltage/current equals a resistor value of 7.25 k. To tune the OTP point, the value
of the series resistor can be adapted. When Rseries is zero, the trigger point is around
87 C. For Rseries = 1.8 k, it is around 95 C. For Rseries = 3.3 k, it reaches 105 C. All
values given are typical and for indication only.
The NTC is often placed near the hottest component on the PCB, which is not always
close to the IC. The long PCB track from NTC to the PROTECT pin can act as an antenna
and pick up disturbances. A filter against such disturbances can be constructed without
additional components by placing the series resistor (which is often required anyway)
between the NTC and the PROTECT pin and as close as possible to the pin.
4.6 GND pin
Do not treat the GND pin as a small signal ground because the DRIVER discharge current
also flows through this pin.
6.
Because the diode is used in a forward direction, low reverse leakage may not seem an important parameter for a diode in this
application. However, a low leakage diode is preferred here because of its higher forward voltage at low forward currents. To
ensure that during the brownin/brownout measurement the entire measurement current flows into the PROTECT pin and not into
the diode, the forward current must be negligible at 0.26 V forward voltage and at high temperatures.
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5. Start-up circuit
5.1 Start-up circuit with two resistors and two diodes
Figure 27(a) shows a simple circuit with good performance which not only caters for
start-up but also for X-capacitor discharge.
Figure 27(b) shows the circuit shown in Figure 27(a) but drawn to show more clearly how
the VCC capacitor is charged.
/
%'D
%'G
%'E
%'F
&;
/
&
5
1
%'G
%'D
5
5
'
&&
&
&;
'
'
'
%'F
%'E
9&&
5
&&
1
DDD
a. Start-up circuit using diodes in series
DDD
b. Simplified representation
Fig 27. Start-up circuits
Once the bulk capacitor C1 is fully charged, the diodes of the bridge stop conducting.
During the positive half of the mains, current flows from the L terminal through resistor R1,
diode D1, capacitor C11 + C7, and bridge rectifier BD1b to the N terminal, charging
C11 + C7. Diode D2 prevents that R2 discharges C11 + C7 during the positive half. During
the negative half of the mains, resistor R2, diode D2, capacitor C11 + C7, and bridge
rectifier BD1a form the current charging path. Diode D1 prevents discharge by R1.
The charge current into the VCC capacitor approximately can be calculated with
Equation 15:
I ch
 --2-  2  V

mains – V CC

= ------------------------------------------------------------- – I CC  startup 
R1
(15)
Where:
• Vmains is the effective mains voltage
• ICC(startup) is the supply current of the IC in power-down mode (11 A)
When VCC = 11 V, the average over the charge period (0 V to Vstartup), the charge current
for R1 = R2 = 2.4 M at 90 V (AC) equals:
I ch
AN11588
Application note
 --2-  2  90 V – 11 V


–6
–6
= ---------------------------------------------------------- – 11 A = 29  10 – 11  10 = 18 A
2.4 M
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The diodes do not block the X-capacitor discharge path. The discharge of the X-capacitor
takes place via resistor R1 or R2 through the series diode to the VCC pin. From the VCC
pin, there are several paths to ground. Even when the IC is in power-down mode a clamp
on the VCC pin is active. From ground, it can find its return path to the X-capacitor through
one of the bridge diodes.
For discharging the X-capacitor, the RC time constant must be < 1 s. For an X-capacitor of
220 nF, it means a maximum value for resistors R1 and R2 of 4.5 M. For 330 nF, it
means a maximum 3 M. Table 4 contains for some start-up resistors and their
performance in a 65 W application.
Table 4.
Start-up resistors and their performance
Rstartup (M)
tstartup (s)
Pnoload (mW) at
230 V (AC)
90 V (AC)
115 V (AC)
1
0.72
0.52
80
1.5
1.13
0.80
64
2
1.57
1.12
59
2.4
1.99
1.34
53
2.7
2.37
1.57
52
3
2.77
1.77
51
Selecting the best value for the start-up resistor is a balance between start-up time and
no-load input power, taking into account the maximum allowed value for discharging the
X-capacitor.
When cost is premium and start-up time or no-load power is less important, it is possible
to omit the two diodes D1 and D2. The circuit still works fine. However, while one start-up
resistor is providing the charge current, the other is loading the VCC capacitor and draining
off part of the charge current. It has a negative effect on the start-up time. For 90 V (AC),
the start-up time increases about 20 %, for 115 V (AC) about 10 %.
Example: Ich = 18 A at Rstartup = 2.4 M (see Figure 27). Idch = 4.6 A (take the average
value of VCC). The available charge current for the VCC capacitor decreases with 25 %.
The voltage rating of most 1206 resistors is only 200 V. For 230 V (AC) applications, the
voltage across the start-up resistors can become nearly 400 V. If SMD type resistors are
used, resistors R1 and R2 must each be split into two resistors in series.
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5.2 Measuring start-up time
Capacitance across the bridge diodes changes the wave shape of the voltage before the
bridge rectifier regarding the primary ground. It can significantly decrease the start-up
time. Depending on the capacitance of the mains supply to ground, connecting the ground
clip of an oscilloscope to the primary ground of the flyback converter can add a few nF
across the bridge diodes.
Make sure that the board has no capacitive coupling to primary ground, so the correct
worst case start-up time is measured:
• To detect mains switch-on, use a current probe in the mains input cable.
• The same current probe in the mains input cable can also be used to detect when the
supply starts switching. The time, from the moment the supply starts to switch until it
reaches 90 % of the output voltage, is only a few milliseconds. It can be ignored
regarding the total start-up time. If it is required to measure the output voltage with an
oscilloscope, remove the Y-cap so that there is no capacitive coupling to primary
ground.
• Use a resistor load instead of an electronic load. Remove the Y-cap if electronic load
must be used.
Also important when measuring the start-up time:
• Make sure that the VCC capacitor is entirely discharged before starting a
measurement.
• Do not connect a probe or multimeter to the VCC. Even a 10 M impedance
influences the measurement.
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6. Layout recommendations
The component numbers used in this section refer to the schematic diagram
(see Figure 1).
6.1 Input section
• Keep the mains tracks (L and N) low ohmic and close to each other to avoid loops.
• Position common-mode chokes away from the power section (MOSFET and
transformer) and from each other to prevent magnetic coupling to any of the other
components.
• Keep tracks from the bridge rectifier to capacitor C1 low ohmic and close to each
other.
6.2 Power section
• The connection from the negative terminal of the bridge rectifier to the current sense
resistor R11 must be routed via capacitor C1.
• The connection from the positive terminal of the bridge rectifier to the transformer
must be routed via capacitor C1.
• Keep the cross section of the loop from capacitor C1 via the transformer, MOSFET
Q1 and current sense resistor R11 back to capacitor C1 as small as possible.
• Place capacitor C2 close to capacitor C1.
• Place peak clamp circuit consisting of resistors R9 and R10, capacitor C3 and diode
D1 close to the transformer and away from the TEA1832(L)TS.
• If MOSFET Q1 has a metal tab, insulate it from the heat sink. Connect the heat sink to
the primary power ground.
6.3 Auxiliary winding
• Place rectifier diode D3, coil L1, VCC capacitor C11, and the network, consisting of
resistor R18 and capacitor C12, close to the auxiliary winding.
• The connection of the ground of the auxiliary winding to the central signal ground
point must go via capacitor C11. To avoid the noise in this ground causing noise in the
PROTECT pin, the ISENSE pin, etc., use a separate track.
• Connect the central signal ground with a low-ohmic track to the central power ground
(capacitor C1).
• Keep the cross section of the loop from the auxiliary winding (via diode D3 and coil
L1) to VCC capacitor C11 and back to the auxiliary winding as small as possible. The
same applies for the network R18/C12.
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6.4 Flyback controller
• Place the TEA1832(L)TS away from the transformer and MOSFET Q1.
• Keep the connection from current sense resistor R11 to the TEA1832(L)TS close to
the ground track.
• Place VCC decoupling capacitor C7 close to the VCC pin.
• The connection from the VCC pin to VCC capacitor C11 must be routed via VCC
decoupling capacitor C7.
• The connection from the GND pin to the central signal ground must be routed via VCC
decoupling capacitor C7.
• Place resistors R13 and R13a close to the ISENSE pin.
• Place resistors R4 and R5 close to the PROTECT pin.
• Place diode D4a, capacitor C10, and resistor R17 close to the PROTECT pin as well.
The other terminal of capacitor C10 must have a short connection to the GND pin.
• Place capacitor C9 close to the CTRL pin.
6.5 Mains insulation
• Keep at least 6 mm distance between the copper tracks of the primary and the
secondary side.
• Place Y-cap CY1 close to the transformer.
6.6 Secondary side
• Heat sink secondary diode D5:
Connect the metal tab (which is internally connected to the cathode) directly to the
heat sink. Connect the heat sink to the positive output track. If the diode has no metal
tab, the heat sink can be connected to ground or to the positive output track.
• Keep the cross section of the loop from the transformer via diode D5 and capacitors
C13 and C14 back to the transformer as small as possible. Keep output tracks close
to each other.
• Use a separate signal ground for resistor R24 and shunt regulator U3. Connect the
signal ground from resistor R24 and shunt regulator U3 via capacitor C19 to the
power ground at capacitors C13 and C14.
• Place capacitor C19 close to resistors R20 and R23.
• The connection of resistors R20 and R23 to the positive output voltage must go via
capacitor C19 to capacitors C13 and C14.
• Place the shunt regulator U3 and surrounding components away from transformer.
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7. Abbreviations
Table 5.
Abbreviations
Acronym
Description
CCM
Continuous Conduction Mode
CDM
Charged Device Model
DCM
Discontinuous Conduction Mode
EMI
ElectroMagnetic Interference
LEB
Leading-Edge Blanking
NTC
Negative Temperature Coefficient
OCP
OverCurrent Protection
OPP
OverPower Protection
OSCP
Output Short Circuit Protection
OTP
OverTemperature Protection
OVP
OverVoltage Protection
PFC
Power Factor Corrector
SMPS
Switched Mode Power Supply
UVLO
UnderVoltage LockOut
VCO
Voltage Controlled Oscillator
8. References
AN11588
Application note
[1]
TEA1832TS data sheet — GreenChip SMPS control IC
[2]
TEA1832LTS data sheet — GreenChip SMPS control IC
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9. Legal information
9.1
Definitions
Draft — The document is a draft version only. The content is still under
internal review and subject to formal approval, which may result in
modifications or additions. NXP Semiconductors does not give any
representations or warranties as to the accuracy or completeness of
information included herein and shall have no liability for the consequences of
use of such information.
9.2
NXP Semiconductors does not accept any liability related to any default,
damage, costs or problem which is based on any weakness or default in the
customer’s applications or products, or the application or use by customer’s
third party customer(s). Customer is responsible for doing all necessary
testing for the customer’s applications and products using NXP
Semiconductors products in order to avoid a default of the applications and
the products or of the application or use by customer’s third party
customer(s). NXP does not accept any liability in this respect.
Export control — This document as well as the item(s) described herein
may be subject to export control regulations. Export might require a prior
authorization from competent authorities.
Disclaimers
Limited warranty and liability — Information in this document is believed to
be accurate and reliable. However, NXP Semiconductors does not give any
representations or warranties, expressed or implied, as to the accuracy or
completeness of such information and shall have no liability for the
consequences of use of such information. NXP Semiconductors takes no
responsibility for the content in this document if provided by an information
source outside of NXP Semiconductors.
In no event shall NXP Semiconductors be liable for any indirect, incidental,
punitive, special or consequential damages (including - without limitation - lost
profits, lost savings, business interruption, costs related to the removal or
replacement of any products or rework charges) whether or not such
damages are based on tort (including negligence), warranty, breach of
contract or any other legal theory.
Notwithstanding any damages that customer might incur for any reason
whatsoever, NXP Semiconductors’ aggregate and cumulative liability towards
customer for the products described herein shall be limited in accordance
with the Terms and conditions of commercial sale of NXP Semiconductors.
Right to make changes — NXP Semiconductors reserves the right to make
changes to information published in this document, including without
limitation specifications and product descriptions, at any time and without
notice. This document supersedes and replaces all information supplied prior
to the publication hereof.
Suitability for use — NXP Semiconductors products are not designed,
authorized or warranted to be suitable for use in life support, life-critical or
safety-critical systems or equipment, nor in applications where failure or
malfunction of an NXP Semiconductors product can reasonably be expected
to result in personal injury, death or severe property or environmental
damage. NXP Semiconductors and its suppliers accept no liability for
inclusion and/or use of NXP Semiconductors products in such equipment or
applications and therefore such inclusion and/or use is at the customer’s own
risk.
Applications — Applications that are described herein for any of these
products are for illustrative purposes only. NXP Semiconductors makes no
representation or warranty that such applications will be suitable for the
specified use without further testing or modification.
Customers are responsible for the design and operation of their applications
and products using NXP Semiconductors products, and NXP Semiconductors
accepts no liability for any assistance with applications or customer product
design. It is customer’s sole responsibility to determine whether the NXP
Semiconductors product is suitable and fit for the customer’s applications and
products planned, as well as for the planned application and use of
customer’s third party customer(s). Customers should provide appropriate
design and operating safeguards to minimize the risks associated with their
applications and products.
Evaluation products — This product is provided on an “as is” and “with all
faults” basis for evaluation purposes only. NXP Semiconductors, its affiliates
and their suppliers expressly disclaim all warranties, whether express, implied
or statutory, including but not limited to the implied warranties of
non-infringement, merchantability and fitness for a particular purpose. The
entire risk as to the quality, or arising out of the use or performance, of this
product remains with customer.
In no event shall NXP Semiconductors, its affiliates or their suppliers be liable
to customer for any special, indirect, consequential, punitive or incidental
damages (including without limitation damages for loss of business, business
interruption, loss of use, loss of data or information, and the like) arising out
the use of or inability to use the product, whether or not based on tort
(including negligence), strict liability, breach of contract, breach of warranty or
any other theory, even if advised of the possibility of such damages.
Notwithstanding any damages that customer might incur for any reason
whatsoever (including without limitation, all damages referenced above and
all direct or general damages), the entire liability of NXP Semiconductors, its
affiliates and their suppliers and customer’s exclusive remedy for all of the
foregoing shall be limited to actual damages incurred by customer based on
reasonable reliance up to the greater of the amount actually paid by customer
for the product or five dollars (US$5.00). The foregoing limitations, exclusions
and disclaimers shall apply to the maximum extent permitted by applicable
law, even if any remedy fails of its essential purpose.
Safety of high-voltage evaluation products — The non-insulated high
voltages that are present when operating this product, constitute a risk of
electric shock, personal injury, death and/or ignition of fire. This product is
intended for evaluation purposes only. It shall be operated in a designated
test area by personnel that is qualified according to local requirements and
labor laws to work with non-insulated mains voltages and high-voltage
circuits.
The product does not comply with IEC 60950 based national or regional
safety standards. NXP Semiconductors does not accept any liability for
damages incurred due to inappropriate use of this product or related to
non-insulated high voltages. Any use of this product is at customer’s own risk
and liability. The customer shall fully indemnify and hold harmless NXP
Semiconductors from any liability, damages and claims resulting from the use
of the product.
Translations — A non-English (translated) version of a document is for
reference only. The English version shall prevail in case of any discrepancy
between the translated and English versions.
9.3
Trademarks
Notice: All referenced brands, product names, service names and trademarks
are the property of their respective owners.
GreenChip — is a trademark of NXP Semiconductors N.V.
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10. Contents
1
1.1
1.2
1.2.1
1.2.2
1.2.3
1.3
1.4
1.5
1.6
1.7
2
2.1
2.2
3
3.1
3.2
3.2.1
3.2.2
3.2.3
3.2.4
3.3
3.3.1
3.3.2
3.3.3
3.3.4
3.3.5
3.3.6
3.3.7
3.4
3.4.1
3.4.2
3.4.2.1
3.4.2.2
3.4.3
3.4.3.1
3.4.3.2
3.4.4
3.4.5
3.4.5.1
3.4.5.2
3.4.6
3.4.7
3.4.8
General description . . . . . . . . . . . . . . . . . . . . . . 3
Scope . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Features . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
Power features . . . . . . . . . . . . . . . . . . . . . . . . . 3
Green features . . . . . . . . . . . . . . . . . . . . . . . . . 3
Protection features . . . . . . . . . . . . . . . . . . . . . . 3
Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Differences between the TEA1832 and the
TEA1738 series . . . . . . . . . . . . . . . . . . . . . . . . 4
Differences between the TEA1832 and the
TEA1733 series . . . . . . . . . . . . . . . . . . . . . . . . 5
Latch and safe restart version . . . . . . . . . . . . . 6
Application diagram . . . . . . . . . . . . . . . . . . . . . 7
Pinning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Pinning diagram . . . . . . . . . . . . . . . . . . . . . . . . 8
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . 8
Functional description . . . . . . . . . . . . . . . . . . 11
General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Start-up . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Charging the VCC capacitor . . . . . . . . . . . . . . 11
Start-up conditions . . . . . . . . . . . . . . . . . . . . . 12
Soft start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Power control . . . . . . . . . . . . . . . . . . . . . . . . . 13
General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Input biasing . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Peak current control . . . . . . . . . . . . . . . . . . . . 14
Frequency control . . . . . . . . . . . . . . . . . . . . . . 14
Burst mode . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Switch-off delay . . . . . . . . . . . . . . . . . . . . . . . 15
Leading-Edge Blanking (LEB) . . . . . . . . . . . . 16
Protections . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Overview. . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Protection handling: Restart . . . . . . . . . . . . . . 17
Regular restart (short). . . . . . . . . . . . . . . . . . . 17
OPP restart (slow restart, not in TEA1832LTS) 17
Protection handling: Latch . . . . . . . . . . . . . . . 18
Latched off-state . . . . . . . . . . . . . . . . . . . . . . . 18
Resetting a latched protection . . . . . . . . . . . . 18
Internal VCC OVP . . . . . . . . . . . . . . . . . . . . . . 18
UnderVoltage LockOut (UVLO) . . . . . . . . . . . 19
Restart version (TEA1832TS). . . . . . . . . . . . . 19
Latch version (TEA1832LTS) . . . . . . . . . . . . . 19
Maximum duty cycle protection . . . . . . . . . . . 19
Internal OverTemperature Protection (OTP). . 19
Brownin/brownout and external OTP
protection (PROTECT pin) . . . . . . . . . . . . . . . 19
3.4.8.1
3.4.8.2
3.4.8.3
3.4.8.4
3.4.8.5
3.4.9
Brownin . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Brownout . . . . . . . . . . . . . . . . . . . . . . . . . . . .
High/low line compensation . . . . . . . . . . . . . .
External OverTemperature Protection (OTP) .
Clamp. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
External output OverVoltage Protection
(OVP) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
3.4.10
OverPower Protection (OPP). . . . . . . . . . . . .
3.4.10.1 Slow restart delay timing calculation
(TEA1832TS only) . . . . . . . . . . . . . . . . . . . . .
3.4.10.2 Ratio OPP time-out/restart delay . . . . . . . . . .
3.4.11
OverCurrent Protection (OCP). . . . . . . . . . . .
3.4.12
Temporary peak power . . . . . . . . . . . . . . . . .
3.4.13
Output Short Circuit Protection (OSCP). . . . .
3.5
Slope compensation . . . . . . . . . . . . . . . . . . .
3.6
Driver . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
3.7
Frequency modulation . . . . . . . . . . . . . . . . . .
4
Application (pin-by-pin) . . . . . . . . . . . . . . . . .
4.1
VCC pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
4.1.1
Start-up circuit . . . . . . . . . . . . . . . . . . . . . . . .
4.1.2
VCC capacitor . . . . . . . . . . . . . . . . . . . . . . . .
4.1.3
Auxiliary winding or take-over winding . . . . . .
4.1.4
Maximum start-up current . . . . . . . . . . . . . . .
4.1.5
PCB layout . . . . . . . . . . . . . . . . . . . . . . . . . . .
4.2
CTRL pin . . . . . . . . . . . . . . . . . . . . . . . . . . . .
4.3
DRIVER pin . . . . . . . . . . . . . . . . . . . . . . . . . .
4.4
ISENSE pin . . . . . . . . . . . . . . . . . . . . . . . . . .
4.4.1
General application recommendations . . . . .
4.4.2
Configuration order . . . . . . . . . . . . . . . . . . . .
4.4.3
Configuring the current sense resistor . . . . . .
4.4.4
Calculating the maximum temporary peak
output power . . . . . . . . . . . . . . . . . . . . . . . . .
4.4.5
Tuning the OPP compensation (Ropc) . . . . . .
4.4.6
Calculating the effect of the OPP
compensation. . . . . . . . . . . . . . . . . . . . . . . . .
4.4.7
Disabling the overpower protection . . . . . . . .
4.4.8
Calculating the OVP. . . . . . . . . . . . . . . . . . . .
4.5
PROTECT pin . . . . . . . . . . . . . . . . . . . . . . . .
4.5.1
Mains detection . . . . . . . . . . . . . . . . . . . . . . .
4.5.2
External OverTemperature Protection (OTP) .
4.6
GND pin . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
5
Start-up circuit. . . . . . . . . . . . . . . . . . . . . . . . .
5.1
Start-up circuit with two resistors and two
diodes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
5.2
Measuring start-up time . . . . . . . . . . . . . . . . .
6
Layout recommendations . . . . . . . . . . . . . . .
6.1
Input section. . . . . . . . . . . . . . . . . . . . . . . . . .
20
20
21
23
24
24
25
27
28
29
29
30
33
34
34
35
35
35
35
36
36
36
36
37
37
37
37
37
39
39
40
40
41
42
42
42
42
43
43
45
46
46
continued >>
AN11588
Application note
All information provided in this document is subject to legal disclaimers.
Rev. 2 — 24 June 2015
© NXP Semiconductors N.V. 2015. All rights reserved.
50 of 51
AN11588
NXP Semiconductors
GreenChip TEA1832(L)TS fixed frequency flyback controller
6.2
6.3
6.4
6.5
6.6
7
8
9
9.1
9.2
9.3
10
Power section . . . . . . . . . . . . . . . . . . . . . . . . .
Auxiliary winding . . . . . . . . . . . . . . . . . . . . . . .
Flyback controller . . . . . . . . . . . . . . . . . . . . . .
Mains insulation . . . . . . . . . . . . . . . . . . . . . . .
Secondary side . . . . . . . . . . . . . . . . . . . . . . . .
Abbreviations . . . . . . . . . . . . . . . . . . . . . . . . . .
References . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Legal information. . . . . . . . . . . . . . . . . . . . . . .
Definitions . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Disclaimers . . . . . . . . . . . . . . . . . . . . . . . . . . .
Trademarks. . . . . . . . . . . . . . . . . . . . . . . . . . .
Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
46
46
47
47
47
48
48
49
49
49
49
50
Please be aware that important notices concerning this document and the product(s)
described herein, have been included in section ‘Legal information’.
© NXP Semiconductors N.V. 2015.
All rights reserved.
For more information, please visit: http://www.nxp.com
For sales office addresses, please send an email to: [email protected]
Date of release: 24 June 2015
Document identifier: AN11588
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