Application Notes

AN10966
UBA2024B CFL ballast up to 120 V (AC) without voltage
doubler
Rev. 1 — 13 December 2010
Application note
Document information
Info
Content
Keywords
UBA2024B, CFL, integrated half-bridge driver with integrated switches,
lighting
Abstract
Application note for NXP Semiconductors UBA2024B CFL driver running
on 100 V (AC) to 120 V (AC) mains without a voltage doubler circuit
AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Revision history
Rev
Date
Description
v.1
20101213
First issue
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
1. Introduction
This application note describes the design process of a CFL ballast for mains voltages
from 100 V (AC) to 120 V (AC) and should therefore be considered as an addition to
Application note AN10713: 18 W CFL lamp design using UBA2024 application
development tool and application examples; see Ref. 1.
An application development tool is available to simplify lamp design and calculation of the
resonance circuit. It can also generate a bill of materials needed to build the application.
This application development tool is only available on the CD-Rom that comes with the
UBA2024B development box and is optimized for designing UBA2024B applications.
The UBA2024 is a family of integrated half-bridge power IC's designed for use in an
integrated/sealed Compact Fluorescent Lamp (CFL) with lamp powers of up to 26 W.
Typical input voltages are 100 V (AC) to 127 V (AC) and 220 V (AC) to 240 V (AC). The
term lamp is used throughout this publication meaning both burner and electronic ballast.
The UBA2024 includes both half-bridge power transistors with a level-shifter and drivers,
bootstrap circuitry, an internal power supply, a precision oscillator and a start-up frequency
sweep function for soft start and/or quasi-preheating. Due to the high level of integration,
only a few external components are needed in a lamp ballast with the UBA2024.
The UBA2024 family of integrated CFL ballast controller IC's have different RDS(on),
package and current ratings; see Table 1.
Table 1.
The UBA2024 family
Type number
Package
Parameters
Name
Description
Version
RDS(on)
ISAT
UBA2024P
DIP8
plastic dual in-line package; 8 leads (300 mil)
SOT97-1
9
900 mA
UBA2024T
SO14
plastic small outline package; 14 leads; body
width 3.9 mm
SOT108-1 9 
900 mA
UBA2024AP
DIP8
plastic dual in-line package; 8 leads (300 mil)
SOT97-1
UBA2024AT
SO14
plastic small outline package; 14 leads;
body width 3.9 mm
SOT108-1 6.4 
UBA2024BP
DIP8
plastic dual in-line package; 8 leads (300 mil)
SOT97-1
2
2500 mA
UBA2024BT
SO14
plastic small outline package; 14 leads;
body width 3.9 mm
SOT108-1 2 
2500 mA
Table 2.
6
1350 mA
1200 mA
UBA2024 application range
Type number
Lamp power[1] (W)
Mains voltage (AC)
Input configuration
UBA2024P
5 to 14
100 V to 127 V
voltage doubler
220 V to 240 V
standard
100 V to 127 V
voltage doubler
220 V to 240 V
standard
100 V to 127 V
standard
UBA2024T
UBA2024AP
15 to 18
UBA2024AT
UBA2024BP
5 to 26
UBA2024BT
[1]
Overall lamp power including driver circuit.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
1.1 UBA2024 family features
•
•
•
•
Integrated half-bridge power IC for CFL applications (both power and controller)
Accurate oscillator with adjustable frequency
Soft-start by frequency sweep down from start frequency
Quasi-preheat option (programmable sweep down timing)
1.2 System benefits
•
•
•
•
•
•
•
Allows for very compact integrated lamp ballast which fits a small shell
Low cost CFL applications due to low component count
Higher reliability due to low component count
Longer lamp life due to quasi-preheat
Easily applicable
Based on EZ-HV Silicon-On-Insulator (SOI) technology
UBA2024P, UBA2024AP, UBA2024T and UBA2024AT can withstand a maximum
voltage of 550 V
• UBA2024BP and UBA2024BT can withstand a maximum voltage of 250 V
1.3 UBA2024B benefits
The half-bridge power transistors of the UBA2024B have a lower Ron and allow higher
current through the power transistors. However, the breakdown voltage is limited and
therefore a UBA2024B cannot be used for mains voltages above 127 V (AC).
To achieve operation with a burner voltage of 80 V (RMS) and above from a 100 V (AC) to
120 V (AC) mains voltage, two topologies are commonly used as shown in Figure 1.
The first possibility is to use a "voltage doubler" circuit, that requires an extra electrolytic
capacitor. On top of that the half-bridge switches require a voltage rating equal to that
needed for a 230 V (AC) application.
BALLAST
AND
LAMP
120 V(AC)
voltage doubler
Fig 1.
BALLAST
AND
LAMP
120 V(AC)
bridge rectifier
019aaa827
Mains input configurations for 100 V (AC) to 120 V (AC)
Please refer to Ref. 1 “Application note AN10713” for more information about the voltage
doubler topology.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
The other possibility to drive burners at voltages above 80 V (RMS) is using resonant gain
from the LC-tank without a voltage doubler. This method is described in this application
note. The benefits are a lower voltage rating for the half-bridge switches and no
requirement for an extra electrolytic capacitor for the voltage doubler.
The reactive current will be higher when using resonant gain from the LC-tank. As this
current passes the integrated half-bridge switches in the IC, the half-bridge switches must
have a lower Ron to limit the power dissipation.
2. Circuit diagrams
U1
LFILT
D1
J1
1
HV
FS
RFUS
3
1
2
CON2
3
OUT
LR
4
8
RC
UBA2024BP
CFS
CBUS
2
VDD
ROSC
CFL
CHB1
D2
7
6
1
5
COSC
SW
CRS
D4
D3
CSW
CRP
CHB2
CDVDT PGND
2
4
RSW
CVDD
SGND
019aaa828
Fig 2.
Application diagram for the UBA2024BP
Figure 2 shows the typical circuit diagram of the UBA2024BP in a DIP8 package. Figure 3
shows a version with the UBA2024BT in an SO14 package.
U1
LFILT
HV
6
4
7
D1
J1
1
CFL
CHB1
D2
FS
RFUS
CFS
CBUS
2
1
CON2
2
3
LR
4
8
11
2
OUT
UBA2024BT
14
D3
3
5
CRS
D4
1
9
CRP
CHB2
CDVDT PGND
12
10
13
VDD
RC
ROSC
SW
SGND1
SGND2
COSC
SGND3
SGND4
SGND5
CSW
RSW
CVDD
SGND6
SGND7
019aaa829
Fig 3.
Application diagram for the UBA2024BT
The input circuit of the application comprises a Fusistor (RFUS), a diode rectifier bridge (D1
to D4), and a buffer capacitor (CBUS). LFILT suppresses the harmonic disturbances on the
mains supply from the half-bridge switching frequency.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
The controller IC is connected to timing components using the via pins RC for the
oscillator and SW for the frequency sweep during preheat. The output of the IC drives the
dV/dt capacitor CDVDT, the resonant tank and the burner, where CHB1 and CHB2 are used
for DC blocking. See Ref. 2 “Data sheet UBA2024” for a functional description of the IC.
U1
LFILT
HV
6
7
VDD
CHB1
LRS2
D1
J1
1
LRS1
LR
D2
RFUS
1
CBUS
2
2
3
ROSC
CFS
FS
3
8
RC
UBA2024BP
4
OUT
5
1
COSC
SW
CON2
CSW
D4
D3
CHB2
CFL
CRP
CDVDT PGND
4
2
RSW
CVDD
SGND
019aaa830
Fig 4.
Application diagram for the UBA2024BP with inductive preheating
An example using inductive preheating with a UBA2024BP is shown in Figure 4. In this
schematic, only one resonant capacitor CRP is needed. In this case, you can apply the
total resonant capacitance here. The design of a circuit with inductive preheat lies beyond
the scope of this document.
In Figure 2 and Figure 3 there are two resonant capacitors present, named CRP and CRS.
When the filament current (which in these two schematics is equal to the current through
capacitor CRS) is higher than the maximum allowed filament current (ILL), the total
resonant capacitance can be divided over both CRS and CRP. Part of the ILH (before CRP
was present) will now pass through CRP bringing ILL to the required value.
Figure 5 shows the flow of the lamp currents ILH (Lead High), ID (Discharge) and ILL (Lead
Low), where the discharge current is in fact the lamp current. The relationship between
these currents is as follows:
I LH =
2
I D + I LH
2
(1)
CRS
ILL
ID
ILH
Fig 5.
019aaa831
Lamp currents
So the total resonance capacitance is:
(2)
C RES = C RS + C RP
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Dividing the resonant capacitance CRES over CRS and CRP results in the specified filament
current and avoids decreased lifetime of the burner filaments due to enhanced
evaporation of the emissive material of the filaments and severe end-blackening of the
tube.
3. Modes of lamp power control
It should be understood that a resonant tank with a self-inductance L, a capacitance C and
a burner with an operating voltage Vlamp that is driven by a square wave voltage VHB (the
half-bridge output) with a given frequency f ( = 2f) will result in a determined output
power Pout. This is shown in Figure 6 and Equation 3.
L
VHB =
2
· VBUS
π
Vlamp
C
019aaa832
Fig 6.
Resonant tank with a burner driven by a square wave voltage
Equation 3 refers to Figure 6.
2
P out
V BUS
V HB 2
V lamp
V lamp
2
2
2 2
2
2
= ----------------   -------------  ------- –  1 –  LC  = -------------   ------------- –  1 –  LC 




L
V lamp 
L
V lamp
(3)
Resonant gain (Q > 1) is required when the burner operating voltage is higher than 2  
times the average bus voltage. Using resonant gain with a fixed frequency would give a
very high dependency of the lamp power on the frequency and other component values.
Therefore, the spread in lamp power given normal component tolerances would be too
high. An example of a transfer function with resonant gain running on a fixed frequency is
shown in Figure 7. The resulting variation in power is also shown for a frequency deviation
of  5 % of its nominal value.
The UBA2024B can be used in two modes of operation, fixed frequency and frequency
control by feedback. The choice of operating mode depends on the ratio between
operation voltage of the burner and the bus voltage (rectified mains). Fixed frequency
operation is applied when no resonant gain is required which is the case when:
Vlamp  VBUS  2  
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
019aaa833
35
Plamp
(W)
30
019aaa834
30
Plamp
(W)
nominal
nominal
25
20
20
15
10
10
5
0
0
30
40
50
60
30
f (kHz)
Fig 7.
40
50
60
f (kHz)
Resonant gain in a fixed frequency application
with a resonant gain tank
Fig 8.
Resonant gain in a feedback controlled
frequency application
A resonant tank driven close to its resonant frequency will operate in a similar way to a
current source. The lamp voltage will have a spread due to temperature, aging and
production. Therefore, in the case where a high gain is needed (as the lamp voltage is
high), it is desirable to operate the tank close to the resonant frequency as this gives the
smallest spread in power. This is shown in Figure 8, where a frequency deviation only
leads to small variations in power. Further details about this operating mode can be found
in Section 3.2.
3.1 Fixed frequency operation
Fixed frequency operation is well known and proven in the UBA2024(A). The half-bridge
switching frequency is determined by ROSC and COSC in Equation 4:
1
f osc ,HB = ----------------------------------------k  R OSC  C OSC
(4)
The oscillator constant k has a typical value of 1.1, see Ref. 2 “Data sheet UBA2024”. The
calculation tool calculates the inductor and capacitor values of the LC-tank in such a way
that the IC will not run into hard switching at normal operation. In this mode of operation,
practical values for ROSC range between 50 k and 400 k. Note that the lower the value
of ROSC, the higher the VDD output current is which increases the total package
dissipation. Practical values for COSC range between 100 pF and 1 nF. The recommended
value for COSC is 180 pF for 40 kHz to 50 kHz and 270 pF for 25 kHz to 30 kHz.
The oscillator start frequency is approximately 2.5 times the nominal frequency. It
gradually decreases, depending on the lamp type and temperature, until the nominal
operating frequency is reached.
The lamp inductor LR and lamp capacitors (CRS + CRP) gradually boost the lamp voltage
as the output frequency approaches the resonance frequency until it is sufficient to ignite
the lamp. The current in the resonance circuit flows through the filaments providing
quasi-preheating. The UBA2024 circuitry stops the frequency sweep at the resonance
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
frequency fres, if the lamp has not yet ignited (see Ref. 2 “Data sheet UBA2024” for
details). This ensures a maximum effort to ignite the lamp. The resonance frequency
depends on LR and the total capacitance CRS and CRP:
1
f res = -----------------------------------------------2 L R  C RS + C RP 
(5)
As the ignition frequency (fign) is higher than or equal to the resonance frequency, the
resonance frequency should be chosen to ensure the preferred ignition frequency totals:
1.6  fburn  fign  1.8  fburn.
3.2 Feedback controlled frequency operation
We advise the use of this topology when the burner operating voltage is higher than 2  
times the average bus voltage. The resonant tank needs to boost the voltage, therefore
the Q factor of the tank must be higher than 1 after the lamp ignites (Q > 1). The
expression for the output power of the resonant tank is shown in Equation 6:
2
V BUS
V lamp
2
2 2
2
P out = ----------------   -------------  ------- –  1 –  LC 
 V lamp  
L
(6)
where:
 = 2  f
(7)
The aim is to find inductor and capacitor values that generate the required output power
and at the same time set the IC to the matching frequency. A transfer function with
resonant gain will have a peak at a certain optimum frequency. An example of such a
transfer function is shown in Figure 9.
Another goal is to control the operating frequency of the IC so there is zero voltage
switching and the ballast is operating at the peak of the transfer function where the
calculated lamp power is delivered.
The frequency control is designed to enable the IC to operate close to the peak frequency
of the transfer function. This is beneficial because the slope of the transfer function is not
very steep at this point.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
019aaa835
25
Plamp
Pout
(W)
20
15
10
f(optimum)
5
0
20
40
60
80
100
f (kHz)
Fig 9.
Example of a transfer function for a resonant gain LC-tank loaded with an ignited
burner
3.2.1 Feedback controlled frequency using CDVDT
When an electronic ballast is running near capacitive mode at Zero Voltage Switching
(ZVS), the current through the LC-tank and lamp has a phase angle with respect to the
half-bridge voltage that is negative. In other words, the half-bridge voltage lags the coil
current. When the LC-tank and the lamp have an inductive character, the opposite is the
case and this means that the coil current lags the half-bridge voltage.
Close to the peak of the power transfer characteristic, the phase shift of the coil current
compared to the half-bridge voltage will be very low, as shown in Figure 10. We use the
coil current with a calculated CDVDT such that the UBA2024B will operate on the edge of
hard switching.
019aaa836
200
Vhb
(V)
0.6
IL
(A)
9.9641 ms, 223.994 mA
120
0.2
9.96546 ms, 30.5835 mA
0
−0.2
60
0
−20
9.944
9.952
9.960
9.968
−0.6
9.984
9.976
t (ms)
Fig 10. Half-bridge output voltage and coil current with CDVDT controlled frequency
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
The UBA2024 incorporates a feature originally intended for self-protection during ignition.
If the IC enters a hard switching condition, its internal self-protection circuitry will draw
charge from the CSW capacitor resulting in an increase in the switching frequency. The
frequency increase will reduce the hard switching to below 14 V and as a result the IC will
not overheat or be damaged by the switching losses due to charging and discharging
CDVDT. This internal self-protection circuitry together with a calculated CDVDT is used as a
feedback control loop to control the switching frequency.
Firstly the resonant tank and anticipated operating frequency are determined. In principle,
the operating frequency is a user input with an advised value of 40 kHz, but any frequency
between 20 kHz and 80 kHz could be given. The NXP Semiconductors tool then
calculates the LC tank for a specified operating frequency which is 2 kHz above the peak
in the resonant tank transfer function. Given this condition and Equation 6, an optimal
resonance capacitance and inductance is found.
It is necessary to ensure that the IC will run on this specified frequency and that it keeps
running on this frequency. If this is not a fixed frequency, it will vary a few kHz due to
component values and burner voltage (e.g. in case of a cold burner). This feedback
control loop is achieved by calculating the capacitance for CDVDT which is needed for a
coil current during the dead time after the trailing edge of the half-bridge output voltage
that equals:
I deadtime  t deadtime
C DVDT = -----------------------------------------------V bridge – 1
(8)
The value of CDVDT, that provides a frequency 2 kHz above the peak in the resonant tank
transfer function, depends on the components in the LC-tank and the properties of the
burner.
In the case of frequency control by CDVDT, both coil current and phase determine the
frequency. The dead time of the UBA2024B is fixed. A matching CDVDT capacitor value
can be calculated using the coil current values at the beginning and the end of the dead
time, enabling the UBA2024B to set the frequency to that where the required power is
delivered.
When hard switching occurs, there is still a voltage present over the load with a certain
polarity at the end of the dead time, as the coil current is still flowing to and from the load.
This will lead to extra losses in the half-bridge switches, but as stated earlier a protection
feature will prevent excessive hard switching, causing the IC to operate near hard
switching. The calculation tool calculates the CDVDT value needed to reach the hard
switch level allowed at the required operating frequency.
The CDVDT capacitor is charged and discharged by the inductive load during the dead
time. The coil current must not change polarity before the other half-bridge switch is
switched on. The half-bridge voltage and coil current can be seen in Figure 10.
3.2.2 Feedback controlled frequency using zero crossing of the coil current
Figure 11 illustrates when the tool returns a value for CDVDT that is relatively small. This is
the case when the LC-tank/burner combination already has a capacitive-like character
and does not need much additional capacitance to be running on the edge of hard
switching at the operation frequency.
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The coil current changes polarity during and at the end of the dead time, the half-bridge
output is charged to the allowed level of hard switching by the coil current flowing in the
opposite direction. The IC will protect itself against this kind of hard switching by
increasing the frequency. This is another way of hard switching caused by zero crossing
of the coil current during the non-overlap or dead time.
Frequency control at zero crossing of the coil current is also known as Zero Voltage
Switching (ZVS), a technique known to be used in discrete Colpitts self-oscillating
electronic ballasts.
This is not a preferred method of running on the edge of hard switching. The slopes of the
half-bridge voltage are very steep which may cause ElectroMagnetic Interference (EMI). A
second disadvantage is that high current flows through the body diodes of the half-bridge
switches which is disadvantageous for the efficiency. CDVDT could be increased to achieve
frequency control by CDVDT (waveform shown in Figure 11).
019aaa837
150
0.6
Vhb
(V)
IL
(A)
9.9641 ms, 206.827 mA
110
0.3
9.96546 ms, −59.1043 mA
70
0
−0.3
30
0
−10
9.944
9.952
9.960
9.968
−0.6
9.984
9.976
t (ms)
Fig 11. Half-bridge output voltage and coil current with zero crossing controlled
frequency
3.2.3 Losses due to hard switching
When the IC is working in feedback controlled frequency operation, it will operate on the
edge of hard switching which will lead to additional losses. Hard switching will not occur all
the time but as a function of the bus voltage ripple.
This is indicated in Figure 12.
The CSW capacitor is charged when the IC is not hard switching and discharged during
hard switching. The feedback system will then balance itself. As a result the hard
switching will only occur for approximately 25 % of the time. The hard switching voltage
has a maximum level of 14 V and the average power losses due to hard switching can be
calculated using Equation 9:
t hsw
2
P hsw = -------------  f burn  C DVDT  V hsw
TV
(9)
BUS
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
019aaa838
184
VBUS
(V) 176
168
VBUS(max)
160
152
144
hard switching
area
VBUS(min)
136
thsw
128
120
period of VBUS
112
104
96
82
84
86
88
90
92
94
96
98
100
t (ms)
Fig 12. Hard switching occurrence
4. Preheating
In this section the preheat methodologies are explained for both a feedback controlled
frequency application and a fixed frequency application. The starting frequency is set for
both topologies and consequently the time needed to reach ignition frequency. The
circuitry connected to pin SW has therefore changed compared to the default fixed
frequency application as shown in Data sheet UBA2024. The new schematic is shown in
Figure 13.
U1
LFILT
HV
6
7
VDD
R10
D1
J1
1
CFL
CHB1
D2
FS
RFUS
1
CON2
2
3
4
8
UBA2024B
CFS
CBUS
2
3
LR
OUT
CRP
CDVDT PGND
5
1
SW
D3
CHB2
COSC
R11
CRS
D4
ROSC
RC
CSW
4
2
CVDD
RSW
SGND
019aaa839
Fig 13. Schematic diagram of a fixed frequency application with new SW pin circuitry
Remark: In the applications as shown in Figure 2 and Figure 3 note that R10 is not
mounted and that capacitor "R11" is replaced by a 0  resistor.
A controlled preheat current where the current would appear as shown in Figure 14 is not
possible. There is no free pin available and a sense resistor would lead to additional
power losses.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
ignition
ILL
preheat
burn
t
019aaa840
Fig 14. Controlled preheat current
Section 4.1 and Section 4.2 describe how a controlled preheat can be approximated for
both a feedback controlled frequency application and a fixed frequency application. Proof
of concept is shown in Section 11.2 that these approximations of a controlled preheat are
adequate solutions to prevent lamp glow.
If filament specifications are unknown, a rule of thumb is that the optimal ratio between the
filament resistance at ignition and cold filament resistance is approximately 5 : 1. With a
preheat time between 500 ms and 600 ms this ratio can be reached. With a cold start not
only is the ignition voltage is higher but also the starting voltage. Both ignition and starting
cause more damage in the case of a cold start.
4.1 Start-up of a feedback controlled frequency application
Since the operating frequency is determined by operation on the edge of hard switching
as explained in Section 3.2, the start-up behavior of the application has been optimized
for this mode of operation. The circuit that connects to pin SW is different to the default
circuit as shown in Data sheet UBA2024.
The timing components ROSC and COSC are chosen in such a way that the oscillator starts
at a required preheat frequency, typically approximately 10 kHz above the ignition
frequency.
It is now possible to set the starting frequency and consequently, the time needed to reach
the ignition frequency by choosing the right values for the timing components ROSC and
COSC. This method allows the designer to program both the preheat frequency and time.
A major advantage of this method, compared to a discrete solution using a PTC resistor, is
that the same preheat energy is applied as with the discrete solution but without using an
expensive PTC. In addition, a PTC resistor has to dissipate power to remain tripped
during operation. Using a preheat time of at least 400 ms increases the switch cycle life
time of the application and reduces the need for the saturation current through the coil as
the ignition voltage decreases. The calculation tool will calculate ROSC for a given COSC
and a default preheat time of 600 ms and will return the actual preheat time. If another
preheat time is required, the user can change ROSC and immediately see the effect on the
calculated preheat time.
AN10966
Application note
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
In principle, the minimum frequency is determined by the zero voltage feedback control
loop, but to avoid problems (with e.g. a cold burner) a safeguard frequency is introduced
by adding a resistor in parallel with CSW, see Figure 15. This extra resistor RSW will
determine the minimum frequency of the oscillator.
U1
HV
6
7
VDD
ROSC
FS
3
8
RC
UBA2024BP
OUT
PGND
5
1
4
2
SW
SGND
RSW
CSW
COSC
019aaa841
Fig 15. SW circuit for a frequency controlled feedback operated application
The voltage on the SW pin determines the amplitude and as a consequence, the
frequency on the RC pin. Resistor RSW will limit the voltage on the SW pin because CSW
will be charged with a current of 280 nA. At a level of 280 nA  4.7 M = 1.32 V, CSW will
no longer be charged and the frequency will no longer increase. The time needed to reach
this voltage is determined by CSW.
Default values for resistor RSW and capacitor CSW used in the calculation tool for a
preheat time of 600 ms are 4.7 M and 470 nF, respectively.
With fixed frequency operation applying the standard application from the datasheet,
where the operating frequency is determined by the values of resistor ROSC and capacitor
COSC, the preheat frequency starts at 2.5 times the operating frequency. The preheat
current as a function of time will look similar to curve (2) in Figure 16, referred to as
Quasi-preheat, starting at 100 kHz. However, in the frequency controlled feedback
operation where resistor ROSC and capacitor COSC only determine the starting frequency
of the IC and CDVDT determines the operating frequency, the preheat current will look
similar to curve (3) in Figure 16. The advantage of the latter is that more energy is put into
the filaments during the quasi-preheat which results in a more predictable ignition and an
increased filament lifetime.
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
ignition
preheat
Ii
burn
(2)
(3)
(1)
t
019aaa842
(1) Controlled preheat.
(2) Quasi-preheat starting at 100 kHz.
(3) Preheat starting at fign + 10 kHz.
Fig 16. Preheat and ignition, preheat current as a function of time
In Figure 16 curve (1) represents the preheat current from a system with controlled
preheat (see Figure 14), where the frequency is constant during preheat and decreases to
accomplish ignition after the preheat time has passed. Note that preheating at a frequency
of approximately 10 kHz above the ignition frequency results in a good approximation of
controlled preheat system, e.g. UBA2028.
The recommended value for COSC for frequency controlled feedback operation is 1200 pF.
Lower values of COSC slightly decrease the duty cycle of the half-bridge output and lead to
higher hard switching losses on the leading edge of the half-bridge output voltage.
Smaller values for COSC can be used for fixed frequency operation; see Section 4.2.
4.2 Start-up of a fixed frequency application
The time needed to sweep down (set by CSW only as RSW is not present when the IC is
used in the standard application shown in the datasheet) from the start frequency to the
resonance frequency can be used as an approximation for the ignition time. The sweep
time is typically CSW (nF)  10.3 ms. The ignition time is shorter for large values because
the lamp ignites before the resonance frequency is reached. The typical ignition time is 1 s
when CSW = 330 nF. A larger CSW increases the sweep time and improves the preheating
of the electrodes. However, the rise of the pre-ignition lamp ignition voltage is also slower.
Both a quasi-preheat that is too short and a voltage rise that is too slow increase the glow
time of the lamp. This reduces the lifetime of the lamp. During the glow phase the lamp is
ignited, but the filaments and the gas inside the lamp are not at their final operating
temperature. The UBA2024 has a mechanism to push extra energy into the lamp during
this glow phase, which is described in the UBA2024 datasheet. This will make the lamp
reach its final light output quicker which gives a longer lamp lifetime. Typical values for
CSW are between 33 nF and 330 nF when the IC is used in the standard fixed frequency
operation mode.
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
U1
HV
6
7
VDD
ROFFS
FS
3
8
ROSC
RC
UBA2024BP
OUT
PGND
5
1
4
2
SW
CSWF
SGND
CSW
RSW
COSC
019aaa843
Fig 17. SW circuit for a fixed frequency operated application
In Figure 17 a schematic diagram of the SW circuitry is shown which also provides a
starting frequency of approximately 10 kHz above the ignition frequency. In this operation
the operating frequency is still determined by ROSC and COSC according to Equation 5.
The starting frequency is determined by the offset voltage that is determined by the
voltage divider ROFFS and RSW. The capacitor CSW now works as a filter for this offset
voltage. After start up CSWF will be charged further until the IC has reached the operating
frequency. This preheating method is similar to the solid blue curve shown in Figure 16.
The default component values used in the calculation tool are CSWF = 470 nF,
CSW = 10 nF, RSW = 10 k and COSC = 220 pF. These defaults are used by the calculation
tool to determine E48 values for both ROFFS and ROSC, resulting in a preheat time of
600 ms. Finally the tool will also return the actual preheat time using the calculated ROFFS
and ROSC.
5. Design of a 26 W non-dimmable CFL
This section explains the selection criteria for the component values. It also clarifies how
to enter the appropriate component values into the application development tool. With the
calculation tool and the help of some practical guidelines it should be easy to set-up
designs of different lamp powers. Throughout this document the light source itself is
referred to as the burner. The tool is intended for all use cases of burners operating at
50 V to 130 V; 8 W to 24 W. In this application note, a PL-C 4P, 26 W burner with a
specified power of 24 W operating at 80 V is taken as an example.
5.1 Selecting a buffer capacitor and fusistor
Lamp power of a resonant tank with burner always depends on the bus voltage. When
using 220 V (AC) or 110 V (AC) with a voltage doubler, this relation is more relaxed than
for rectified 110 V (AC).
A bus voltage ripple ratio of between 15 % and 20 % determined by the buffer capacitor is
recommended for proper operation. If the buffer capacitor has a value resulting in a ripple
ratio of less than 15 %, the application will draw higher than necessary charge current
peaks from the mains which reduces the power factor. In the tool, this ratio is calculated
and returned to the user. Choosing a smaller buffer capacitor will lead to a higher ripple
and a lower average bus voltage.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
As a result of this, the LC-tank may have to provide more resonant gain which requires
larger resonant capacitors. So choosing a smaller buffer capacitor does not necessarily
lead to a smaller application. In addition, the Crest factor of the lamp power would become
worse. The following table shows recommended values for the buffer capacitor and
fusistor for a standard input configuration as shown in Figure 1 per power range of the
application running on a mains voltage of 120 V (AC) and 60 Hz.
Table 3.
Advised values for the standard input configuration
Lamp power range[1]
CBUS
RFUS[2]
4W
10 F; 200 V
18  (0.5 W)
5 W to 6 W
15 F; 200 V
12  (0.5 W)
7 W to 8 W
15 F; 200 V
12  (1 W)
9 W to 11 W
22 F; 200 V
5.6  (1 W)
12 W to 14 W
22 F; 200 V
5.6  (2 W)
15 W to 18 W
22 F; 200 V
5.6  (2 W)
19 W to 22 W
33 F; 200 V
3.3  (2 W)
23 W to 26 W
33 F; 200 V
3.3  (2 W)
[1]
Overall lamp power including driver circuit.
[2]
Minimum continuous power rating.
5.2 Using the calculation tool
This section describes how to use the calculation tool and how to interpret the results.
5.2.1 Input values
The application development tool calculates the component values based on the following
input parameters:
•
•
•
•
•
•
•
Burner power
Burner operating voltage
Burner ignition voltage
Filament resistance
Maximum filament current
Mains input voltage and frequency (typical operating voltage)
Combined value of the DC blocking capacitors
Figure 18 shows the part of the application development tool where the input parameters
can be entered. The example shows the design of a 26 W lamp. This is the total lamp
power, which means 24 W burner power and about 2 W loss in the electronic ballast. The
burner used in this example is a replaceable burner. It is based on a G24q-3 fitting with
the following parameters.
•
•
•
•
•
AN10966
Application note
Burner power = 24 W
Burner voltage = 80 V
Ignition voltage = 460 V
Warm filament resistance = 9 
Maximum filament current = 320 A
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
The following actions need to be taken:
1. Enter the burner parameters
2. Enter the mains voltage to be used for the 26 W lamp (120 V)
3. Enter the value of the buffer capacitor (33 F)
4. Enter the mains frequency (60 Hz)
5. Enter the total value of the blocking capacitors (300 nF)
6. Enter the required operating frequency (40 kHz)
When using burners with an operating voltage up to 73 V, the resonant tank does not have
to provide resonant gain. When the UBA2024B is used with burners that have a high
operating voltage, the resonant tank provides resonant gain and the frequency is
regulated on the edge of hard switching. This frequency regulation is a protection feature
of the UBA2024, intended for self-protection during ignition. The advantage of resonant
gain is that no voltage doubler capacitors are needed which consume a lot of space in a
retrofit CFL. The frequency at which the UBA2024B will run no longer depends on its RC
timing components provided fmin is selected about 5 kHz below the operating frequency.
This will increase the accuracy of the system. The disadvantage of switching on the edge
of hard switching is that there are small switching losses in the half-bridge. The additional
switching losses amount to less than 15 mW for this application.
IC selection
UBA2024BP
Burner power
24 W
Burner operating voltage
80 V
Ignition voltage
460 V
Warm filament resistance
9 Ω
Maximum filament current
320 mA
Mains voltage
120 V
Buffer capacitor
33 μF
Mains frequency
60 Hz
DC blocking capacitance
Desired operating frequency
300 nF
40.0 kHz
019aaa844
Fig 18. Entering the design parameters for a 26 W lamp
Based on the burner parameters, mains voltage and frequency, the buffer capacitor,
selected DC blocking capacitors and the operating frequency required, the calculation of
the LC resonance tank can be executed by pressing the Optimize! button (Figure 23). The
application development tool then calculates recommended values for the resonance
inductor, capacitor and the dV/dt capacitor. The operating frequency is also calculated.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
5.3 Calculation algorithm
After entering all the necessary parameters, the calculation will proceed by returning
recommended values for the resonance capacitor and inductor. If the calculated filament
current is higher than the entered maximum value, the resonance capacitor will be split so
that the requirement for the filament current is met. The required capacitance will be
instantaneously returned as E12 values.
The next step in the calculation is to achieve a zero voltage switching condition by
determining the average coil current during the non-overlap time. The target is to have no
difference between the actual average coil current during the non-overlap time and coil
current during the non-overlap time. A value is now determined for the dV/dt capacitor that
meets these requirements. This capacitance value will also be returned as an E12 value.
Then fine tune to the zero voltage switching condition after the dV/dt capacitor value has
been adapted to an E12 value. Again, the target is to have no difference between the
actual average coil current during the non-overlap time and the required coil current
during the non-overlap time. This is achieved by changing the actual operating frequency
fburn and the resonance inductor Lres, under the following constraints:
• Average lamp power equals the required burner power
• Actual operating frequency is less than or equal to the tank's resonant peak frequency
increased with 2 kHz when the lamp is ignited
The final step is to enter values for the RC timing components ROSC and COSC. A value
higher than 1 nF is recommended for COSC (default is 1.2 nF), and a ROSC value is
advised to set the preheat time to 600 ms. Enter a realistic value that approximates the
advised value and the tool will calculate the preheat time instantaneously.
5.4 Calculation results
Once the calculation is complete the tool will display graphs of the average burner power
as a function of the rectified bridge voltage (see Figure 19), burner power as a function of
frequency (see Figure 20) and burner voltage, filament current and frequency as a
function of time during start-up (preheat and ignition) (see Figure 21) of the application.
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
019aaa845
33
Plamp
(W)
23
13
3
100
150
200
250
300
350
Vbridge (V)
Fig 19. Average burner power as a function of the rectified bridge voltage
019aaa846
30
Plamp
(W)
20
10
0
30
40
50
60
f (kHz)
Fig 20. Burner power as a function of frequency
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
019aaa847
1200
95
f
(kHz)
85
Vlamp
(V)
Ifil
(mA)
f
75
800
65
Ifil
55
400
45
35
Vlamp
25
1000
0
0
200
400
600
800
t (ms)
Fig 21. Burner voltage, filament current and frequency as a function of time during
start-up
The tool will also display a graph of the calculated lamp power at fburn  3 kHz, which will
immediately warn the user if the solution is on a steep slope of the power transfer curve of
the resonant tank. This graph is shown in Figure 22.
019aaa848
24.4
1.0
output
power
(W)
24.175
24.051
ΔP
(%)
0.5
0.52
24.0
0.0
−0.5
−1.0
23.6
23.2
24.051
−1.5
−2.05
−2.0
fburn (−3 kHz)
fburn
actual operating frequency
fburn (+3 kHz)
−2.5
Fig 22. Calculated lamp power variation
The power variation is shown in both W and as a percentage relative to the power at fburn.
The numerical output of the tool is shown in Figure 23 “Input/output data fields”.
AN10966
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22 of 43
AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Optimize
UBA2024BP
IC selection
Burner power
24
W
Burner operating voltage
80
V
460
V
Warm filament resistance
9
Ω
Maximum filament current
320
mA
Mains voltage
120
V (AC)
Ignition voltage
300 mA
120 V(AC)
0%
RSW
4700
kΩ
CSW
470
nF
ROFFS
removed!
kΩ
CSWF
shorted!
nF
ROSC
26.1
kΩ
COSC
1200
pF
43.4
kHz
Buffer capacitor
33
μF
Mains frequency
60
Hz
DC blocking capacitance
300
nF
Actual operating frequency
Desired operating frequency
40.0
kHz
Bridge voltage ripple
PLAMP(AVG)
24.0
W
Starting frequency
83.45
kHz
RLAMP(AVG)
269.42
Ω
Peak frequency
45.59
kHz
Ignition frequency
69.71
kHz
16.9 %
LC-tank
1st estimate
C resonance lamp
10
nF
0
nF
Ignition peak current
2014.8
mA
10
nF
Ignition peak energy
2695.5
mJ
690.0
ms
0.000
mA
C resonance parallel
C resonance total
L resonance
0.6641
CdV/dt
0.68
10.92 nF
0.79 mH
mH
Preheat time
nF
Start-up
Δ I(tD)
Minimum
Average
Maximum
Bridge voltage
139.8
154.8
168.2
V
Power with entered L and C
19.9
24.0
27.9
W
Ambient temperature
FET RMS current
331
371
411
mA
ILL(BURN)
218
218
218
ID(BURN)
249
300
331
−7
ILW(BURN) =
(ID(BURN)2 + ILL(BURN)2)
Phase shift ICOIL and VHB
OK
70
°C
Average power loss in FET
0.46 W
W
mA
Total power loss
0.55 W
W
349
mA
Case temperature
113.3
°C
371
411
mA
Junction temperature
122.1
°C
−17
−25
019aaa849
Fig 23. Input/output data fields
Calculation results are listed along with the entered burner properties, mains voltage and
the required operating frequency which comprise of:
• Component values of the resonant tank (CRS, CRP, LR and CDVDT)
• Actual operating frequency
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
•
•
•
•
•
•
•
•
•
•
•
•
Average lamp power and resistance
Various frequencies (fstart, fignition, fmin, fPEAK)
The preheat time (tph)
Start-up condition
Coil current balance during the non-overlap time (I(tD))
The advised resonant tank components which are calculated at the beginning of the
algorithm (CR(ADV) (total capacitance of CRS + CRP not rounded to E12 values) and
LR(ADV))
Ignition peak current and energy through the coil
Power dissipation and IC temperatures (case and junction)
The advised oscillator resistance for a preheat time of 600 ms (ROSC(ADV))
The voltage across the filaments (VFILAM)
The lamp currents in the burn state of the application (ILL(BURN), ID(BURN), ILH(BURN))
The minimum and maximum rectified mains voltage (VBRIDGE(MIN), VBRIDGE(MAX)-D)
5.4.1 Coil
On completion of the calculation, the tool also returns the most important coil
requirements (example in Figure 24). Together with the inductance entered in Figure 23
and the operating temperature of the inductor there is enough information to design a coil.
Due to losses in the inductor, its operating temperature is higher than the lamp ambient
temperature. When the coil is properly designed, the inductor temperature increase will be
around 40 C above the ambient temperature. When a warm lamp is switched off and
then on again, the inductor should not saturate at this inductor temperature.
Ignition frequency
69.71 kHz
Ignition peak current
2014.8 mA
Ignition peak energy
2695.5 mJ
019aaa850
Fig 24. Coil design parameters
5.4.2 Thermal properties
In this section the estimated dissipated power and junction temperature in the IC are
calculated. See Figure 25 for an example. When the maximum ambient temperature at
which the lamp needs to operate is entered, the anticipated junction temperature is
calculated. The junction temperature must not exceed 150 C. If the junction temperature
does exceed 150 C, the expected operating life time of the IC is significantly reduced.
The maximum stress allowed during the ignition phase is 2500 mA (peak) for the
UBA2024B at a case temperature of 25 C (repetition rate is less than once per hour). The
maximum stress period must not be longer than 1 second.
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Ambient temperature
70 °C
Average power loss in FET
0.42 W
Total power loss
0.51 W
Case temperature
110.4 °C
Junction temperature
118.5 °C
019aaa851
Fig 25. Dissipated power and expected case and junction temperature in the IC
5.5 Choosing the other components
A bridge cell or separate diodes such as the 1N5062 can be used for the rectifier bridge.
The 1N4007 diodes can also be used but they are not avalanche rugged.
For a lamp current  150 mA with CDVDT = 220 pF and for a current  150 mA with
CDVDT = 100 pF, the value of CVDD and CFS is 10 nF.
The recommended half-bridge capacitors (CHB1 and CHB2) are greater than 150 nF when
fout = 40 kHz to 50 kHz and greater than 220 nF when fout = 25 kHz to 30 kHz.
The resonance frequency of the input pi filter, consisting of LFILT and CHB (CHB being the
effective capacitor as seen on pin HV of the IC (the series capacitance of CHB1) and
CHB2), must be at least two times lower than the nominal output frequency.
Remark: Performance and lifetime cannot be guaranteed by using the values given in this
Section. The lamp and the UBA2024 performance interact strongly with each other and
need to be qualified together as a combination.
5.6 Checking the tolerance sensitivity
In this section the stability of the result provided is verified with respect to mains
fluctuations and component tolerances. After the tool has provided a solution for how to
dimension the application, it can also be used to show the result when for example the
mains voltage changes  5 %, or when the resonant capacitors tolerances are taken into
account.
An example in Figure 26 illustrates the effect on the output power when Vmains increases
by 10 % from 120 V (AC) to 132 V (AC). When the increased mains voltage is entered the
actual burning frequency must be adapted to return I(tD) = 0 to zero, so the user manually
optimizes for operation on the edge of hard switching. In order to achieve the condition
I(tD) = 0, the frequency must be decreased. The field PLAMP(AVG) shows what the power is
in this situation, and for the application in question this will result in a 22 % power increase
from 24 W to 29.4 W.
AN10966
Application note
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25 of 43
AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Optimize
UBA2024BP
IC selection
Burner power
24
W
Burner operating voltage
80
V
460
V
Warm filament resistance
9
Ω
Maximum filament current
320
mA
Mains voltage
120
V (AC)
Ignition voltage
300 mA
132 V(AC)
10%
RSW
4700
kΩ
CSW
470
nF
ROFFS
removed!
kΩ
CSWF
shorted!
nF
ROSC
26.1
kΩ
COSC
1200 pF
34.9
Buffer capacitor
33
μF
Mains frequency
60
Hz
DC blocking capacitance
300
nF
Actual operating frequency
Desired operating frequency
40.0
kHz
Bridge voltage ripple
PLAMP(AVG)
29.4
W
Starting frequency
83.45
kHz
RLAMP(AVG)
220.38
Ω
Peak frequency
34.02
kHz
Ignition frequency
70.37
kHz
kHz
14.2 %
LC-tank
1st estimate
C resonance lamp
10
nF
0
nF
Ignition peak current
2033.9
mA
10
nF
Ignition peak energy
2746.9
mJ
650.0
ms
0.661
mA
C resonance parallel
C resonance total
L resonance
0.6641
CdV/dt
0.68
10.92 nF
0.79 mH
mH
Preheat time
nF
Start-up
Δ I(tD)
Minimum
Average
Maximum
Bridge voltage
158.9
172.5
185.2
V
Power with entered L and C
23.5
29.4
34.9
W
Ambient temperature
FET RMS current
343
408
471
mA
ILL(BURN)
175
175
175
ID(BURN)
294
368
343
−7
ILW(BURN) =
(ID(BURN)2 + ILL(BURN)2)
Phase shift ICOIL and VHB
OK
70
°C
Average power loss in FET
0.58 W
W
mA
Total power loss
0.67 W
W
437
mA
Case temperature
123.0
°C
408
471
mA
Junction temperature
133.7
°C
−19
−28
019aaa852
Fig 26. Sensitivity to mains variation (+ 10 %)
This procedure has also been carried out when tolerances of the resonant capacitors and
inductor are taken into account. The result of this exercise is shown in Table 4.
AN10966
Application note
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Rev. 1 — 13 December 2010
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AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Table 4.
Calculated tolerance sensitivity results
Parameter
(nominal)
Vmains = 120 V (AC) = 100 %
CRS = 10 nF = 100 %
L = 0.66 mH = 100 %
Vmains ( 10 %) Vmains (+ 10 %) CRS ( 5 %)
CRS (+ 5 %)
L( 10 %)
L (+ 10 %)
frequency
(43.4 kHz)
49.1 kHz
34.9 kHz
44.7 kHz
42.3 kHz
45.5 kHz
41.6 kHz
PLAMP(AVG)
(24 W)
20.280 W
29.449 W
23.374 W
24.615 W
25.226 W
22.908 W
6. Building the application
6.1 Reference board
6.1.1 External lamp detection circuit
The NXP Semiconductors evaluation board contains an additional lamp detection circuit
which is not required for mass production applications such as CFLi (see Figure 29). The
functioning of this detection circuit is described in this section.
During start-up, preheat and ignition phases, the voltage at the SW pin (pin 1) increases
from 0 V to 1.32 V. At the same time the amplitude of the signal on the RC pin (pin 7)
increases by the same amount. However, if the lamp is not ignited, because it is broken or
missing, the sweep voltage will remain below the 3 V level or even drop to 0 V. The IC will
not operate in Zero Voltage Switching mode (ZVS). Large currents flow in the half-bridge
causing dissipation in the IC to exceed the maximum value. The half-bridge can only
withstand the high dissipation until the junction temperature reaches 150 °C.
At start-up the RC oscillator starts with an amplitude of 2 V on pin RC (pin 8). The
half-bridge frequency is now running at approximately 15 % above the nominal ignition
frequency. When the burner is connected to the circuit the half-bridge operates in ZVS
and the CSW capacitor charges. R6, R7 and C12 create an average DC voltage of the
oscillator voltage on pin RC, which is basically half the amplitude. That voltage is then fed
to the base of Q2-2, which functions as a comparator.
At the same time that CSW is charging, C11 is charged by R3 from VDD. This takes place
with a time constant of (R3//R4)  C11. The charging stops when the voltage on C11
reaches 1.6 V. The voltage on C11 is fed to the emitter of Q2-2 to compare it with its base
voltage.
Under normal conditions during start-up, when the lamp is connected the average DC
voltage from RC rises above 1.6 V at the end of the charging period for C11. The base
emitter voltage of Q2-2 will remain reverse biased and will not turn on. If non-ZVS is
detected in the half-bridge driver switches due to an unconnected or broken lamp, the
charging of CSW stops and the voltage on CSW drops to 0 V. The average DC voltage on
the RC pin reduces to less than 1 V and Q2-2 starts to conduct.
Q2-2 drives the latching transistor Q1-1 and the fault condition is latched by the left diode
of the double diode, D5. At the same time the right diode of D5 will stop the UBA2024B
half-bridge oscillator. The latch can be reset by power cycling the mains voltage with less
than 1 s delay (for the test circuit this depends on the discharge time of C11 and R4). The
latch circuit is designed in such a way that it is not noise sensitive. However, it is better to
keep it away from the large signal tracks.
AN10966
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Typically, the circuit triggers within 0.5 s from start-up when no lamp is connected and also
when a lamp is removed during operation. When the protection has tripped, the dissipated
power in the IC is about 0.6 W. The IC can dissipate this power continuously.
Ensure that there is some reaction time margin (at room temperature) when choosing
C11. Also, consider voltage derating of MLCC capacitors when low voltage types are
used. It is advisable to choose an X7R type of at least 10 V.
The protection circuit places additional capacitive loading (about 5 pF) on pin RC. This
can be significant in fixed frequency operation for small values of COSC . In this case, the
value of COSC is compensated for this effect by lowering ROSC from 200 k to 191 k
(E96 series), giving an operating frequency of 45.9 kHz instead of 43.3 kHz. When the
circuit is used it is advisable to add the extra 5 pF to COSC; see Equation 4.
This additional capacitance can be ignored when the IC is working on the edge of hard
switching, since a COSC = 1.2 nF is recommended to improve the duty cycle of the
half-bridge output voltage.
019aaa363
Fig 27. Photo reference board UBA2024BP (DIP8)
AN10966
Application note
019aaa364
Fig 28. Photo reference board UBA2024BT (SO14)
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xxxxxxxxxxxxxxxxxxxxxxxxx xxxxxxxxxxxxxxxxxxxx xxx
HV
D1
1N4007
K1
W2
2
1
D2
1N4007
3.3 Ω
MKDS 1.5/2
120 VAC
C1
33 μF
200 V
6
5
J2
J1
4
D3
1N4007
C3
150 nF
250 V
T2
3
FS
J3
1
1
1
2
2
2
NM
D4
1N4007
6
7
VDD
VDD
WE_Bobbin EF20
C2
150 nF
250 V
W3
3
2
1.5 mH
R1
NXP Semiconductors
AN10966
Application note
U1
L1
C6
10 nF
C4
TBF
NM
RC
R2
26.1 kΩ
R8
C8
10 nF
0Ω
RC
UBA2024BP
OUT
NM
8
3
R10
TBF
NM
5
1
C7
PGND
0.82 nF
4
500 V
2
SW R11
0Ω
C5
470 nF
R9
SGND 4.7 MΩ
C9
1.2 nF
GND
K3
1
W4 C1
W5 C2
optional ''Lamp Detection Circuit''
MKDS 1.5/2
K2
2
1
C10
10 nF
2000 V
W6 C3
VDD
R3
220 kΩ
W7 C4
4
5
T1-2
BC847BPN
MKDS 1.5/2
3
R6
R7
1 MΩ
1 MΩ
2
3
RC
1
2
R4
33 kΩ
C11
3.3 μF
R5
180 kΩ
C12
220 pF
D5
BAV70W
6
T1-1
BC847BPN
1
GND
019aaa853
Fig 29. Circuit diagram of the UBA2024BP reference board with optional lamp detection circuit
AN10966
29 of 43
© NXP B.V. 2010. All rights reserved.
J1, J2 and J3 are 0  resistors.
UBA2024BP:
J1 = 0.66 mH, default set for 26 W.
J2 = 0.98 mH, 13 W.
J3 = 1.09 mH, 18 W.
Do NOT short more than one jumper at the same time.
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Rev. 1 — 13 December 2010
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2
AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
6.2 Bill of materials
The bill of materials is given in Table 5 for the application example with a PL-C 4P 26W
lamp, including the external lamp detection circuit.
Table 5.
Components used to build the application around the UBA2024BP for driving a PL-C 4P 26W CFL
This table applies to both the UBA2024BP and UBA2024BT reference boards
Reference
Description
Remarks
Value
R1
resistor, fusible; 3R3 / 5 %, 2W NFR
fusistor
3.3 ; 2 W
R2
resistor, thick film, 26K1 / 1 %, 0W1 0603
oscillator resistor
26.1 k; 0.1 W; 1 %
R3[1]
resistor, thick film, 220K / 5 %, 0W1 0603
R4[1]
resistor, thick film, 33K / 5 %, 0W1 0603
33 k; 0.1 W
R5[1]
resistor, thick film, 180K / 5 %, 0W1 0603
180 k; 0.1 W
R6, R7[1]
resistor, thick film, 1M / 5 %, 0W1 0603
1 M; 0.1 W
R8, R11
resistor, thick film, 0R / 1 %, 0W1 0603
R9
resistor, thick film, 4M7 / 1 %, 0W1 0603
R10
220 k; 0.1 W
short
0
not applicable
not mounted
not applicable
C1
capacitor, Al, El, 47 F, 20 %, 200V KXG
high temperature electrolytic
type
47 F; 200 V
C2, C3
capacitor, 150 n, 10 %, 250V DME
C4
not applicable
not mounted
not applicable
C5
capacitor, ceramic, 470n, 10 %, 10V X5R 0603
470 nF; 10 V; 10 %
C6, C8
capacitor, ceramic, 10n, 20 %, 50V X7R 0603
10 nF; 50 V
C7
capacitor, ceramic, 0.82 n, 10 %, 500V
X7R 1206
10 nF; 50 V
C9
capacitor, ceramic, 1n2, 5 %, 50V X7R 0603
oscillator capacitor
1.2 nF; 50 V; 5 %
C10
capacitor, 10n, 5 %, 2KV MKP
lamp capacitor
10 nF; 2 kV; 5 %
C11[1]
capacitor, ceramic, 33, 20 %, 10V Y5V 0805
3.3 F; 10 V
C12[1]
capacitor, ceramic, 220p, 5 %, 50V COG 0603
220 pF; 50 V; 5 %
D1, D2, D3,
D4
diode, standard, 1KV, 1A
mains rectifier diode
D5[1]
diode, small signal, dual, 70V, 200mA
double diode common cathode BAV70W
L1
Inductor RF choke 1m5H, 1R7, 0A43, 10 %
radial type
1.5 mH, 0.43 A
T1[1]
Tor, dual, NPN/PNP, 45V, 100mA
PNP and NPN diode in one
BC847BPN
T2
RF choke, T-H BOBBIN EF-20
E-20 core (select inductance
with jumper
0.66 mH; J1 in place
U1[2]
UBA2024BP, UBA2024BT
CFL driver IC
UBA2024BP
4.7 M; 0.1 W; 1 %
150 nF; 250 V
1N4007
[1]
Component(s) needed for the optional lamp detection circuit.
[2]
2 versions of the demo board are available for the UBA2024BP in a DIP8 package and the UBA2024BT in a SO14 package.
AN10966
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
7. Layout considerations
The UBA2024B PCB layout has a considerable influence on the performance of the IC.
Issues to be taken into account are:
• Coils with open magnetic circuits should not be placed opposite the IC (on the other
side of the PCB). If an axial filter inductor is used for LFILT, it should be placed in the
same direction as the IC to minimize magnetic field pick-up.
• The oscillator pin (pin 7, RC) and the sweep pin (pin 8, SW) should be shielded from
output/lamp by a ground track.
• Components on pins 7 and 8 should be placed as close to the IC as possible.
• Capacitors CVDD and CFS should be placed close to the IC.
• Mains input wires must not run parallel or near the half-bridge signal (pin 5, OUT) or
near the output of the lamp inductor, bypassing the input filter.
• If the UBA2024BT is used, all SGND pins need to be soldered to a copper plane for
effective heat transfer. This copper plane is underneath the IC and extends on both
sides of the IC as far as possible. Fixing the IC to the board using thermal conductive
glue also helps to keep the IC cool.
8. Quick measurements
Table 6 compares the calculated values from the application development tool with the
measured values. The measurements were carried out at 25 C.
Table 6.
Measured values compared with the calculated values
Values
Lamp power
(W)
fburn
(kHz)
Tph
(ms)
fstart
(kHz)
fign
(kHz)
ILL
(mA)
ID
(mA)
ILH
(mA)
Iign(pk)
(mA)
calculated
24
43.4
690
83.4
69.7
218
300
371
2010
measured
23.4
51.5
560
81.7
70
279
279
404
1960
Table 7.
AN10966
Application note
Components used in the application as calculated by the calculation spreadsheet
for driving a PL-C 4P 26 W burner
Component
Value
LRES
0.66 mH
CRS
10 nF
CRP
not mounted
CDVDT
0.82 nF
ROSC
26.1 k
COSC
1200 pF
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
9. Start-up waveforms
The measured waveforms are shown of the lamp voltage and lamp current (Figure 30)
and coil current (Figure 31) during preheat.
Ilamp
Vlamp
019aaa854
L = 0.66 mH
CRS = 10 nF
CDVDT = 0.82 nF
CBUF = 33 F
RFUS = 3.3 
Fig 30. Start-up waveforms showing lamp voltage and current
Remark: Note that the lamp ignites without glow. If lamp glow was present it would
indicate a lamp current before the lamp has ignited. This is not the case here. The ignition
peak voltage is 470 V.
AN10966
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Icoil
Vlamp
019aaa855
Fig 31. Start-up waveforms showing lamp voltage and coil current
The measured peak value of the coil current equals 1960 mA.
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
10. Steady state waveforms
The waveforms in Figure 32 are shown 15 minutes after power on.
Vbridge
Vout
Ilamp
Vlamp
019aaa856
Fig 32. Steady state waveforms of the application at ambient temperature of 25 °C
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Vout
019aaa857
Fig 33. Steady state behavior, showing VOUT and ICOIL at Vbridge(min)
In Figure 33 the measured coil current is shown during the dead time at the trailing edge
of the half-bridge voltage. Hard switching is seen here, while the frequency is controlled
by zero crossing of the coil current. The shape of VOUT has been changed such that the
slopes are not so steep by increasing the calculated CDVDT from 0.68 nF to 0.82 nF.
AN10966
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AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
11. What if...
This section shows examples of practical problems such as coil saturation and lamp glow.
11.1 Coil saturation
Figure 34 illustrates what happens when the coil goes into saturation during ignition.
Ilamp
inductor starts to saturate
inductor heavily saturated
019aaa858
Fig 34. The coil current during ignition when the coil is saturated
In this case the coil current will show excessive peaks which in turn results in the
integrated half-bridge switches going into saturation and consequently damaging the IC.
AN10966
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NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
11.2 Lamp glow
Lamp glow is mainly caused by improper preheating of the filaments
Either a quasi-preheat that is too short or a voltage rise that is too slow will increase the
glow time of the lamp. This reduces the lifetime of the lamp. During the glow phase the
lamp is ignited, but the filaments and the gas inside the lamp are not at their final
operating temperature.
Ilamp
glow phase
Vlamp
019aaa859
Fig 35. Lamp glow caused by improper preheating
In Figure 35 it is clear that there is still a high voltage present at the lamp while at the
same time lamp current is flowing. When the filaments and gas inside the lamp have
reached their normal operating temperature, the voltage at the lamp will drop to its normal
operating value.
This is the preheating method shown in Figure 16 referred to as quasi-preheat and
starting at 100 kHz.
AN10966
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AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
Ilamp
Vlamp
019aaa860
Fig 36. Proper ignition of the lamp due to proper preheating, without glow
Figure 36 shows the ignition of a lamp that is preheated as shown in Figure 16 where
preheating starts at the ignition frequency plus an additional 10 kHz. Note that there is no
lamp glow present due to the filaments having enough time to reach the correct operating
temperature. This method of preheating will increase the life time of the lamp and ensure
that it will pass any on/off test of minimum 10,000 repetitions.
AN10966
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AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
12. References
AN10966
Application note
[1]
Application note AN10713 — 18 W CFL lamp design using UBA2024 application
development tool and application examples
[2]
Data sheet UBA2024 — Half-bridge power IC for CFL lamps
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AN10966
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UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
13. Legal information
13.1 Definitions
Draft — The document is a draft version only. The content is still under
internal review and subject to formal approval, which may result in
modifications or additions. NXP Semiconductors does not give any
representations or warranties as to the accuracy or completeness of
information included herein and shall have no liability for the consequences of
use of such information.
13.2 Disclaimers
Limited warranty and liability — Information in this document is believed to
be accurate and reliable. However, NXP Semiconductors does not give any
representations or warranties, expressed or implied, as to the accuracy or
completeness of such information and shall have no liability for the
consequences of use of such information.
In no event shall NXP Semiconductors be liable for any indirect, incidental,
punitive, special or consequential damages (including - without limitation - lost
profits, lost savings, business interruption, costs related to the removal or
replacement of any products or rework charges) whether or not such
damages are based on tort (including negligence), warranty, breach of
contract or any other legal theory.
Notwithstanding any damages that customer might incur for any reason
whatsoever, NXP Semiconductors’ aggregate and cumulative liability towards
customer for the products described herein shall be limited in accordance
with the Terms and conditions of commercial sale of NXP Semiconductors.
Right to make changes — NXP Semiconductors reserves the right to make
changes to information published in this document, including without
limitation specifications and product descriptions, at any time and without
notice. This document supersedes and replaces all information supplied prior
to the publication hereof.
Suitability for use — NXP Semiconductors products are not designed,
authorized or warranted to be suitable for use in life support, life-critical or
safety-critical systems or equipment, nor in applications where failure or
AN10966
Application note
malfunction of an NXP Semiconductors product can reasonably be expected
to result in personal injury, death or severe property or environmental
damage. NXP Semiconductors accepts no liability for inclusion and/or use of
NXP Semiconductors products in such equipment or applications and
therefore such inclusion and/or use is at the customer’s own risk.
Applications — Applications that are described herein for any of these
products are for illustrative purposes only. NXP Semiconductors makes no
representation or warranty that such applications will be suitable for the
specified use without further testing or modification.
Customers are responsible for the design and operation of their applications
and products using NXP Semiconductors products, and NXP Semiconductors
accepts no liability for any assistance with applications or customer product
design. It is customer’s sole responsibility to determine whether the NXP
Semiconductors product is suitable and fit for the customer’s applications and
products planned, as well as for the planned application and use of
customer’s third party customer(s). Customers should provide appropriate
design and operating safeguards to minimize the risks associated with their
applications and products.
NXP Semiconductors does not accept any liability related to any default,
damage, costs or problem which is based on any weakness or default in the
customer’s applications or products, or the application or use by customer’s
third party customer(s). Customer is responsible for doing all necessary
testing for the customer’s applications and products using NXP
Semiconductors products in order to avoid a default of the applications and
the products or of the application or use by customer’s third party
customer(s). NXP does not accept any liability in this respect.
Export control — This document as well as the item(s) described herein
may be subject to export control regulations. Export might require a prior
authorization from national authorities.
13.3 Trademarks
Notice: All referenced brands, product names, service names and trademarks
are the property of their respective owners.
All information provided in this document is subject to legal disclaimers.
Rev. 1 — 13 December 2010
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AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
14. Tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
The UBA2024 family . . . . . . . . . . . . . . . . . . . . .3
UBA2024 application range . . . . . . . . . . . . . . . .3
Advised values for the standard input
configuration . . . . . . . . . . . . . . . . . . . . . . . . . .18
Calculated tolerance sensitivity results . . . . . .27
Components used to build the application
around the UBA2024BP for driving
a PL-C 4P 26W CFL . . . . . . . . . . . . . . . . . . . .30
Measured values compared with
the calculated values . . . . . . . . . . . . . . . . . . . .31
Components used in the application
as calculated by the calculation spreadsheet
for driving a PL-C 4P 26 W burner . . . . . . . . . .31
AN10966
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AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
15. Figures
Fig 1.
Fig 2.
Fig 3.
Fig 4.
Fig 5.
Fig 6.
Fig 7.
Fig 8.
Fig 9.
Fig 10.
Fig 11.
Fig 12.
Fig 13.
Fig 14.
Fig 15.
Fig 16.
Fig 17.
Fig 18.
Fig 19.
Fig 20.
Fig 21.
Fig 22.
Fig 23.
Fig 24.
Fig 25.
Fig 26.
Fig 27.
Fig 28.
Fig 29.
Fig 30.
Fig 31.
Fig 32.
Fig 33.
Fig 34.
Mains input configurations for 100 V (AC)
to 120 V (AC) . . . . . . . . . . . . . . . . . . . . . . . . . . . . .4
Application diagram for the UBA2024BP . . . . . . . .5
Application diagram for the UBA2024BT . . . . . . . .5
Application diagram for the UBA2024BP with
inductive preheating. . . . . . . . . . . . . . . . . . . . . . . .6
Lamp currents . . . . . . . . . . . . . . . . . . . . . . . . . . . .6
Resonant tank with a burner driven by
a square wave voltage . . . . . . . . . . . . . . . . . . . . . .7
Resonant gain in a fixed frequency application
with a resonant gain tank . . . . . . . . . . . . . . . . . . . .8
Resonant gain in a feedback controlled
frequency application . . . . . . . . . . . . . . . . . . . . . . .8
Example of a transfer function for a resonant
gain LC-tank loaded with an ignited burner . . . . .10
Half-bridge output voltage and coil current
with CDVDT controlled frequency . . . . . . . . . . . . .10
Half-bridge output voltage and coil current with
zero crossing controlled frequency . . . . . . . . . . .12
Hard switching occurrence. . . . . . . . . . . . . . . . . .13
Schematic diagram of a fixed frequency
application with new SW pin circuitry. . . . . . . . . .13
Controlled preheat current . . . . . . . . . . . . . . . . . .14
SW circuit for a frequency controlled feedback
operated application . . . . . . . . . . . . . . . . . . . . . .15
Preheat and ignition, preheat current as
a function of time . . . . . . . . . . . . . . . . . . . . . . . . .16
SW circuit for a fixed frequency operated
application . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .17
Entering the design parameters for
a 26 W lamp. . . . . . . . . . . . . . . . . . . . . . . . . . . . .19
Average burner power as a function of the
rectified bridge voltage. . . . . . . . . . . . . . . . . . . . .21
Burner power as a function of frequency . . . . . . .21
Burner voltage, filament current and frequency
as a function of time during start-up. . . . . . . . . . .22
Calculated lamp power variation . . . . . . . . . . . . .22
Input/output data fields. . . . . . . . . . . . . . . . . . . . .23
Coil design parameters . . . . . . . . . . . . . . . . . . . .24
Dissipated power and expected case and
junction temperature in the IC . . . . . . . . . . . . . . .25
Sensitivity to mains variation (+ 10 %). . . . . . . . .26
Photo reference board UBA2024BP (DIP8). . . . .28
Photo reference board UBA2024BT (SO14) . . . .28
Circuit diagram of the UBA2024BP reference
board with optional lamp detection circuit . . . . . .29
Start-up waveforms showing lamp voltage
and current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . .32
Start-up waveforms showing lamp voltage and
coil current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .33
Steady state waveforms of the application at
ambient temperature of 25 °C . . . . . . . . . . . . . . .34
Steady state behavior, showing VOUT and ICOIL
at Vbridge(min) . . . . . . . . . . . . . . . . . . . . . . . . . . . . .35
The coil current during ignition when the coil
is saturated . . . . . . . . . . . . . . . . . . . . . . . . . . . . .36
AN10966
Application note
Fig 35. Lamp glow caused by improper preheating . . . . 37
Fig 36. Proper ignition of the lamp due to proper
preheating, without glow . . . . . . . . . . . . . . . . . . . 38
All information provided in this document is subject to legal disclaimers.
Rev. 1 — 13 December 2010
© NXP B.V. 2010. All rights reserved.
42 of 43
AN10966
NXP Semiconductors
UBA2024B CFL ballast up to 120 V (AC) without voltage doubler
16. Contents
1
1.1
1.2
1.3
2
3
3.1
3.2
3.2.1
3.2.2
3.2.3
4
4.1
4.2
5
5.1
5.2
5.2.1
5.3
5.4
5.4.1
5.4.2
5.5
5.6
6
6.1
6.1.1
6.2
7
8
9
10
11
11.1
11.2
12
13
13.1
13.2
13.3
14
15
16
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
UBA2024 family features . . . . . . . . . . . . . . . . . 4
System benefits . . . . . . . . . . . . . . . . . . . . . . . . 4
UBA2024B benefits . . . . . . . . . . . . . . . . . . . . . 4
Circuit diagrams . . . . . . . . . . . . . . . . . . . . . . . . 5
Modes of lamp power control . . . . . . . . . . . . . . 7
Fixed frequency operation . . . . . . . . . . . . . . . . 8
Feedback controlled frequency operation . . . . 9
Feedback controlled frequency using CDVDT . 10
Feedback controlled frequency using zero
crossing of the coil current . . . . . . . . . . . . . . . 11
Losses due to hard switching . . . . . . . . . . . . . 12
Preheating . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Start-up of a feedback controlled frequency
application . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Start-up of a fixed frequency application. . . . . 16
Design of a 26 W non-dimmable CFL. . . . . . . 17
Selecting a buffer capacitor and fusistor. . . . . 17
Using the calculation tool . . . . . . . . . . . . . . . . 18
Input values . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Calculation algorithm . . . . . . . . . . . . . . . . . . . 20
Calculation results . . . . . . . . . . . . . . . . . . . . . 20
Coil . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Thermal properties . . . . . . . . . . . . . . . . . . . . . 24
Choosing the other components. . . . . . . . . . . 25
Checking the tolerance sensitivity . . . . . . . . . 25
Building the application . . . . . . . . . . . . . . . . . 27
Reference board . . . . . . . . . . . . . . . . . . . . . . . 27
External lamp detection circuit . . . . . . . . . . . . 27
Bill of materials . . . . . . . . . . . . . . . . . . . . . . . . 30
Layout considerations. . . . . . . . . . . . . . . . . . . 31
Quick measurements. . . . . . . . . . . . . . . . . . . . 31
Start-up waveforms . . . . . . . . . . . . . . . . . . . . . 32
Steady state waveforms . . . . . . . . . . . . . . . . . 34
What if... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Coil saturation. . . . . . . . . . . . . . . . . . . . . . . . . 36
Lamp glow . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
References . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
Legal information. . . . . . . . . . . . . . . . . . . . . . . 40
Definitions . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Disclaimers . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Trademarks. . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Tables . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Please be aware that important notices concerning this document and the product(s)
described herein, have been included in section ‘Legal information’.
© NXP B.V. 2010.
All rights reserved.
For more information, please visit: http://www.nxp.com
For sales office addresses, please send an email to: salesaddresses@nxp.com
Date of release: 13 December 2010
Document identifier: AN10966