Driving High Brightness LEDs for Wide Input DC to DC Applications

Driving High Intensity LED Strings in DC to DC Applications
D. Solley, ON Semiconductor, Phoenix, AZ
Abstract
Improvements in high brightness LED
technology offer enhanced energy efficient lighting
solutions in a world where the cost of energy
continues to rise along with increased demand.
Unlike traditional lighting that has been in existence
over the last 50-100 years, LEDs require new driver
solutions that address the challenges of providing a
constant regulated current to a load that can vary in
voltage by +/-30% over process and temperature
variation. This paper will focus on driving high
brightness LEDs from low voltage DC and AC sources
commonly used in lighting applications.
Figure 1: LED Spectral response compared to
Photometric Eye Sensitivity
Solid State Lighting
LED Characteristics
Today, LEDs provide efficient, robust, and
intense light sources. A conventional white LED
consists of a blue LED coated with a yellow
phosphor resulting in a secondary emission of
light that approximates “white” light. The
spectrum of an OSRAM White Golden Dragon is
shown in Figure 1. Alternatively red, green and
blue LEDs may be mixed to create white light.
The arguments for adopting solid state lighting
(SSL) are compelling. LEDs turn on instantly;
even at -40 °C. Life expectancy of 50,000 hrs
makes them attractive in locations where
maintenance is difficult and costly. LEDs are fully
dimmable, with no IR or UV components in the
beam. The small point source facilitates simpler
lens design, while their compact size allows easy
integration into a variety of products. SSL may
play a role in reducing global pollution. The
International Energy Agency estimates that 1900
Mt of C02 was emitted for lighting in 2005. The
figure represents 70% of light passenger car
emissions. At an International Materials Forum in
2005, N. Stath illustrated the efficiency of a
variety of light sources since 1879. Fluorescent
light sources show efficacies between 80-100
Lumens/Watt; mercury and CFL sources
between 40-60 Lumens/Watt; incandescent and
halogen
light
sources
between
10-20
Lumens/Watt. Recent announcements by LED
manufacturers
(Cree,
Nichia)
mention
demonstrated
efficacy
between
130-150
Lumens/Watt exceeding that available from
fluorescent and metal halide sources. While
efficacy continues to increase, costs per lumen
continue to improve at a rapid pace, driven by
technology and manufacturing advances.
Due to the steep V/I curve of the LED and to
achieve optimum performance, it is critical to
drive LEDs with a constant current to achieve
the specified brightness and color. For high
brightness power LEDs the current ranges from
150 – 1500 mA, 350 mA being a common value.
Manufacturer’s data for several product families
is shown in Table 1. From the data, the minimum
and maximum voltage for a generic LED was
created. The output voltage swing for 3 to 6
generic LED strings is included at the end of the
table for reference. It is apparent that +/- 30%
variations can be expected in normal operation.
Table 1.White LED Forward Voltage Comparison
Vendor
(Model)
Cree
(XR-E)
Luxeon
(K2)
Osram
(Golden)
Osram
(Platinum)
“Generic
LED”
3 LEDs
4 LEDs
5 LEDs
6 LEDs
Current
(A)
0.35
0.70
0.35
0.70
1.00
Vmin (V)
@ TJmax (°C)
2.36
2.81
2.54
2.72
2.78
Vmax (V)
@ 25 °C
3.90
4.35
4.23
4.41
4.95
0.35
2.30
3.80
0.70
2.41
4.30
0.35
0.70
0.35
0.70
0.35
0.70
0.35
0.70
0.35
0.7
2.30
2.41
6.90
7.22
9.20
9.62
11.50
12.03
13.80
14.43
4.23
4.95
12.69
14.85
16.92
19.80
21.15
24.75
25.38
29.70
Figure 2: OSRAM OSTAR™
Driver Definition
There are numerous ways that LEDs may be
combined in series or series/parallel strings.
A typical interior automotive lighting application
might require 200 lumens. Depending on LED
selection, this requires a series string of 3 to 6
LEDs. Figure 2 shows a multi LED device
package. Hence to drive any LED combination
that may be envisaged, an efficient, high density,
cost effective constant current converter with
both a wide input (8 V to 19 V) and a wide output
(6.9 V to 30 V) range is required. A pure buck or
a pure boost switching regulator topology is not
sufficiently flexible. What is proposed for the
LED driver is a current regulated, non inverting
buck/boost converter. A high side current
sensing scheme is preferable since for
automotive applications the LED string cathode
is connected to chassis ground.
Also, to
maximize converter performance, a sensing
scheme having a low loss (e.g. 200 mV) is
required. The schematic of the LED driver is
illustrated in Figure 3.
Figure 3: Schematic of Buck Boost Converter showing low drop high side current sensing.
Theory of Operation
The simplified power stage is shown in Figure
4 for clarity. To minimize power dissipation in
the power circuit, low ripple current is required.
So the converter is run in continuous current
mode (CCM). For this analysis, all power
components are assumed ideal. Switches Q1,
Q2 turn on for time D*Ts (D duty cycle, Ts
switching period) charging inductor L1 from
input Vin. When Q1 and Q2 turn off, diodes
D1, D2 deliver the inductor energy to the
output Vout. For the inductor flux (V*µs) to
remain in equilibrium each switching cycle, the
V*µs product across the inductor during each
switch interval must balance.
Figure 4: Power Stage showing BJT base
drives
Vin ⋅ D ⋅ TS = Vout ⋅ (1 − D ) ⋅ TS
(1)
Rearranging equation 1 the voltage gain of
buck boost is given by:
Vout = Vin ⋅
D
(1 − D )
(2)
Varying the duty cycle will vary the output such
that when D is below 0.5, the converter is in
buck mode, when D is above 0.5, the
converter is in boost mode and when D equals
0.5, the voltage gain Vout/Vin is unity.
The ripple current in the inductor is given by
expression
V ⋅ D ⋅ TS
ΔI L1 = in
L1
(3)
Assuming Vin = 12 V and D*TS = 0.5*5 µs, a
value for L1 of 68 μH in equation 3 will
maintain +/-30% ripple current in a 700 mA
application maintaining CCM operation.
MOSFETs or BJTs can be selected as the
primary switches Q1/Q2. NSS40500UW3T2G
and
NSS40501UW3T2G
from
ON
Semiconductor’s e²PowerEdge family of BJTs
were chosen for cost/performance criteria.
They feature ultra low saturation voltage and
high current gain capability (Figure 5).
Figure 5: Collector Emitter Saturation Voltage
versus Collector Current
Turn on, turn off, saturation voltage and
storage time of a BJT are controlled by the
magnitudes of turn on IB1 and turn off IB2 base
currents. The drive currents are identified in
Figure 4. The values of the base drive
resistors R2, R3 and R4 in the schematic may
be adjusted to optimize performance. The
efficiency of the converter can be improved if
the storage time of Q2 is less than Q1. The
reasons for this will be discussed later. Q2’s
storage time can be reduced if it is held out of
saturation by the addition of D3a/b shown in
Figure 3. Once Q2 is near saturation,
additional base current flows through diode
D3a and into the collector junction. This
diversion of base current IB1 reduces the
stored charge in the base region and allows a
faster turn off. Typical TS of 1.5 µs is reduced
to a few hundred nanoseconds.
The controller used to demonstrate the buck
boost topology is ON Semiconductor’s
NCP3063. A functional block diagram is
shown in the Figure 6.
Figure 6: Block Diagram of
NCP3063
This device consists of a 1.25 V reference,
comparator, oscillator, an active current limit
circuit, a driver and a high current output
switch. In its traditional operating mode, the
NCP3063 is a hysteretic, dc - dc converter that
uses a gated oscillator to regulate the output
voltage. Voltage feedback from the output is
sensed at pin 5, and gates the oscillator on/off
to regulate the output. The oscillator frequency
and off-time of the output switch are
programmed by the value selected for the
timing capacitor, CT. CT is charged and
discharged by a 1 to 6 ratio internal current
source and sink, generating a ramp at pin 3.
The ramp is controlled by two comparators
whose levels are set 500 mV apart. In normal
operation, D is fixed at 6/7 or 0.86. The “gated
oscillator” mode is used to protect the LED
string if a LED fails ‘open”. A zener diode
between Vout and pin 5 will clamp the output at
a voltage VZ + 1.25 V.
The NCP3063 can also operate as a
conventional PWM controller, by injecting
current into the CT pin. The control current
may be developed either from the input
source, providing voltage feedforward (via R5)
or from the output current sensing circuit (via
R6). In both cases the slope of the oscillator
ramp changes causing D to vary. In Figure 3,
the current sense resistor R9 is placed in
series with Vout, to satisfy the high side sensing
requirement. The bandgap reference U3,
together with dual NPN transistors Q4a,b and
R13, R14 create two equal current sinks. If U3
is a 1.25 V bandgap and R13, R14 equal 1.24
kΩ (1%) two 1 mA current sinks are formed.
Resistors R10, R11 level shift the current
sense signal IOUT*R9 to satisfy the input
requirements of U2. To create a 210 mV
reference for the current loop, the expression
1 mA * (R10-R11) = 210 mV must be satisfied.
Hence R10 is selected to be 210 Ω larger in
absolute value than R11. Current regulation is
set by the equation Iout * R9 = 210 mV. If R9 is
0.6 Ω, the programmed current is set for 350
mA. The difference between the 210 mV set
point and the current sense is amplified by U2
to create an error voltage. This error voltage
and R6 drives a programmed current into the
CT pin to regulate the LED current.
Because the converter is switching at 200 kHz,
MLCC capacitors in SMT packages can
provide cost effective filtering. Low value
MLCC capacitors (10 µF) have very small ESR
(2 mΩ) and ESL (100 nH) values. When used
in single or parallel combinations they form a
“perfect” capacitor. Ripple voltage is due only
to charging and discharging the capacitor by
the inductor. Two 10 µF, 1210 capacitors are
employed across the input and output of the
driver. The ripple voltage across the input
capacitor = D*Ts* Δ I (L1) / Cin. The ripple
voltage developed across the output capacitor
is given by (1-D)*Ts* Δ I (L1) / Cout.
Converter Waveforms
The voltage waveforms at both the input
(upper trace) and output (lower trace) of the
inductor L1 were measured while the
difference waveform (middle trace) gives the
voltage across the inductor. Figure 7 shows
the converter operating in buck mode, while
Figure 8 illustrates boost operation.
Figure 7: Buck Mode from 12 Vin to 8
Vout
Figure 8: Boost Mode from 12 Vin to
16 Vout
It is evident from Figures 7 and 8 that the
inductor waveforms differ from a classic buck
boost. The voltage across the inductor is
clamped at (Vout–Vin) for the duration of the
storage delay interval TD. During this interval
Q2 is off and Q1 is on for its storage time.
During this period, power is delivered to the
output via Q1 and not by D1. Efficiency
improvement is observed as the VCE(sat) of the
PNP device (100 mV) is less that the voltage
drop across the Schottky diode D1 (300 mV).
If the time delay intervals were reversed and
Q1 turned off first, power would cycle through
the inductor L1, switch Q2 and diode D1. No
power would be delivered to the load until Q2
turned off. The efficiency of the converter is
shown in figure 9 and varied between 75 and
80%. The data was taken with Vin at 12 V
while the output was varied between 11 V and
26 V at 700 mA constant current load.
The V*µs balance expression
equation 1 is modified as follows.
given
in
Vin ⋅ D ⋅ TS ± (Vout − Vin ) ⋅ TD = Vout (1 − D − TD ) ⋅ TS
Conclusions
ON
Semiconductor’s
latest
monolithic
NCP3063
controller
and
family
of
e2PowerEdge ultra low saturation bipolar
transistors are combined to create a non
inverting buck boost topology optimized to
drive strings of LED’s at a constant current. A
high side, low drop, current sensing scheme
has
been
implemented,
targeted
for
automotive and other high efficiency
applications. The output from the current
sense is used to vary the slope of the oscillator
ramp and achieve duty cycle modulation,
independent of the gated oscillator function
provided by the IC. The classic transfer
function of the buck boost converter is
modified by the storage time interval between
the NPN and PNP bipolar switches.
EFFICIENCY, %
85
80
75
References
70
65
60
10
12
14
16
18
20
22
24
26
Vin, Volts
Figure 9: Measurements of Converter
Efficiency
28
N. Stath, “Nano Technology drives LED
Advancements” International Materials Forum
2005
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