LTC1778/LTC1778-1 Wide Operating Range, No RSENSETM Step-Down Controller U FEATURES DESCRIPTIO ■ The LTC®1778 is a synchronous step-down switching regulator controller optimized for CPU power. The controller uses a valley current control architecture to deliver very low duty cycles with excellent transient response without requiring a sense resistor. Operating frequency is selected by an external resistor and is compensated for variations in VIN. ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ No Sense Resistor Required True Current Mode Control Optimized for High Step-Down Ratios tON(MIN) ≤ 100ns Extremely Fast Transient Response Stable with Ceramic COUT Dual N-Channel MOSFET Synchronous Drive Power Good Output Voltage Monitor (LTC1778) Adjustable On-Time (LTC1778-1) Wide VIN Range: 4V to 36V ±1% 0.8V Voltage Reference Adjustable Current Limit Adjustable Switching Frequency Programmable Soft-Start Output Overvoltage Protection Optional Short-Circuit Shutdown Timer Micropower Shutdown: IQ < 30µA Available in a 16-Pin Narrow SSOP Package Discontinuous mode operation provides high efficiency operation at light loads. A forced continuous control pin reduces noise and RF interference, and can assist secondary winding regulation by disabling discontinuous operation when the main output is lightly loaded. Fault protection is provided by internal foldback current limiting, an output overvoltage comparator and optional short-circuit shutdown timer. Soft-start capability for supply sequencing is accomplished using an external timing capacitor. The regulator current limit level is user programmable. Wide supply range allows operation from 4V to 36V at the input and from 0.8V up to (0.9)VIN at the output. U APPLICATIO S ■ , LTC and LT are registered trademarks of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6100678, 6580258, 5847554, 6304066 Notebook and Palmtop Computers Distributed Power Systems U ■ TYPICAL APPLICATIO RON 1.4MΩ Efficiency vs Load Current ION VIN RUN/SS M1 Si4884 TG CC 500pF SW ITH RC 20k INTVCC BG PGOOD PGND + DB CMDSH-3 LTC1778 SGND L1 1.8µH CB 0.22µF BOOST + M2 Si4874 CVCC 4.7µF CIN 10µF 50V ×3 D1 B340A COUT 180µF 4V ×2 VIN 5V TO 28V VOUT = 2.5V VIN = 5V 90 VOUT 2.5V 10A EFFICIENCY (%) CSS 0.1µF 100 VIN = 25V 80 70 R2 30.1k VFB R1 14k 60 0.01 1 0.1 LOAD CURRENT (A) 10 1778 F01b 1778 F01a Figure 1. High Efficiency Step-Down Converter 1778fb 1 LTC1778/LTC1778-1 W W W AXI U U ABSOLUTE RATI GS (Note 1) Input Supply Voltage (VIN, ION)................. 36V to – 0.3V Boosted Topside Driver Supply Voltage (BOOST) ................................................... 42V to – 0.3V SW Voltage .................................................. 36V to – 5V EXTVCC, (BOOST – SW), RUN/SS, PGOOD Voltages ....................................... 7V to – 0.3V FCB, VON, VRNG Voltages .......... INTVCC + 0.3V to – 0.3V ITH, VFB Voltages...................................... 2.7V to – 0.3V TG, BG, INTVCC, EXTVCC Peak Currents .................... 2A TG, BG, INTVCC, EXTVCC RMS Currents .............. 50mA Operating Ambient Temperature Range (Note 4) LTC1778E ........................................... – 40°C to 85°C LTC1778I .......................................... – 40°C to 125°C Junction Temperature (Note 2) ............................ 125°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C U U W PACKAGE/ORDER I FOR ATIO ORDER PART NUMBER TOP VIEW RUN/SS 1 16 BOOST PGOOD 2 15 TG VRNG 3 14 SW FCB 4 LTC1778EGN LTC1778IGN 13 PGND ITH 5 11 INTVCC ION 7 10 VIN VFB 8 9 15 TG VRNG 3 14 SW FCB 4 SGND 6 GN PART MARKING EXTVCC GN PACKAGE 16-LEAD PLASTIC SSOP 1778 1778I TJMAX = 125°C, θJA = 130°C/ W 16 BOOST VON 2 ITH 5 12 BG SGND 6 TOP VIEW RUN/SS 1 ORDER PART NUMBER LTC1778EGN-1 13 PGND 12 BG 11 INTVCC ION 7 10 VIN VFB 8 9 GN PART MARKING EXTVCC 17781 GN PACKAGE 16-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 130°C/ W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS 900 15 2000 30 µA µA 0.800 0.800 0.808 0.812 V V Main Control Loop IQ Input DC Supply Current Normal Shutdown Supply Current VFB Feedback Reference Voltage ITH = 1.2V (Note 3) LTC1778E ITH = 1.2V (Note 3) LTC1778I ∆VFB(LINEREG) Feedback Voltage Line Regulation VIN = 4V to 30V, ITH = 1.2V (Note 3) ∆VFB(LOADREG) Feedback Voltage Load Regulation ITH = 0.5V to 1.9V (Note 3) IFB Feedback Input Current VFB = 0.8V gm(EA) Error Amplifier Transconductance ITH = 1.2V (Note 3) VFCB Forced Continuous Threshold IFCB Forced Continuous Pin Current VFCB = 0.8V tON On-Time ION = 30µA, VON = 0V (LTC1778-1) ION = 15µA, VON = 0V (LTC1778-1) tON(MIN) Minimum On-Time ION = 180µA ● ● 0.792 0.792 0.002 ● %/V – 0.05 – 0.3 % –5 ±50 nA ● 1.4 1.7 2 mS ● 0.76 0.8 0.84 V –1 –2 µA 198 396 233 466 268 536 ns ns 50 100 ns 1778fb 2 LTC1778/LTC1778-1 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 15V unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN tOFF(MIN) Minimum Off-Time ION = 30µA VSENSE(MAX) Maximum Current Sense Threshold VPGND – VSW VRNG = 1V, VFB = 0.76V VRNG = 0V, VFB = 0.76V VRNG = INTVCC, VFB = 0.76V VSENSE(MIN) Minimum Current Sense Threshold VPGND – VSW VRNG = 1V, VFB = 0.84V VRNG = 0V, VFB = 0.84V VRNG = INTVCC, VFB = 0.84V ∆VFB(OV) Output Overvoltage Fault Threshold VFB(UV) Output Undervoltage Fault Threshold VRUN/SS(ON) RUN Pin Start Threshold VRUN/SS(LE) RUN Pin Latchoff Enable Threshold VRUN/SS(LT) RUN Pin Latchoff Threshold RUN/SS Pin Falling IRUN/SS(C) Soft-Start Charge Current VRUN/SS = 0V – 0.5 IRUN/SS(D) Soft-Start Discharge Current VRUN/SS = 4.5V, VFB = 0V 0.8 VIN(UVLO) Undervoltage Lockout VIN Falling VIN(UVLOR) Undervoltage Lockout Release VIN Rising TG RUP TG Driver Pull-Up On Resistance TG RDOWN BG RUP ● ● ● 113 79 158 TYP MAX UNITS 250 400 ns 133 93 186 153 107 214 mV mV mV – 67 – 47 – 93 mV mV mV 5.5 7.5 9.5 % 520 600 680 mV 0.8 1.5 2 V 4 4.5 V 3.5 4.2 V – 1.2 –3 µA 1.8 3 µA ● 3.4 3.9 V ● 3.5 4 V TG High 2 3 Ω TG Driver Pull-Down On Resistance TG Low 2 3 Ω BG Driver Pull-Up On Resistance BG High 3 4 Ω BG RDOWN BG Driver Pull-Down On Resistance BG Low 1 2 Ω TG tr TG Rise Time CLOAD = 3300pF 20 ns TG tf TG Fall Time CLOAD = 3300pF 20 ns BG tr BG Rise Time CLOAD = 3300pF 20 ns BG tf BG Fall Time CLOAD = 3300pF 20 ns ● RUN/SS Pin Rising Internal VCC Regulator VINTVCC Internal VCC Voltage 6V < VIN < 30V, VEXTVCC = 4V ∆VLDO(LOADREG) Internal VCC Load Regulation ICC = 0mA to 20mA, VEXTVCC = 4V VEXTVCC EXTVCC Switchover Voltage ICC = 20mA, VEXTVCC Rising ∆VEXTVCC EXTVCC Switch Drop Voltage ICC = 20mA, VEXTVCC = 5V ∆VEXTVCC(HYS) EXTVCC Switchover Hysteresis ● 4.7 ● 4.5 5 5.3 V – 0.1 ±2 % 4.7 150 V 300 200 mV mV PGOOD Output (LTC1778 Only) ∆VFBH PGOOD Upper Threshold VFB Rising 5.5 7.5 9.5 ∆VFBL ∆VFB(HYS) VPGL PGOOD Lower Threshold VFB Falling – 5.5 – 7.5 – 9.5 % PGOOD Hysteresis VFB Returning 1 2 % PGOOD Low Voltage IPGOOD = 5mA 0.15 0.4 V Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD as follows: LTC1778E: TJ = TA + (PD • 130°C/W) Note 3: The LTC1778 is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). % Note 4: The LTC1778E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC1778I is guaranteed over the full – 40°C to 125°C operating temperature range. 1778fb 3 LTC1778/LTC1778-1 U W TYPICAL PERFOR A CE CHARACTERISTICS Transient Response (Discontinuous Mode) Transient Response VOUT 50mV/DIV Start-Up RUN/SS 2V/DIV VOUT 50mV/DIV VOUT 1V/DIV IL 5A/DIV IL 5A/DIV 20µs/DIV LOAD STEP 0A TO 10A VIN = 15V VOUT = 2.5V FCB = 0V FIGURE 9 CIRCUIT 20µs/DIV LOAD STEP 1A TO 10A VIN = 15V VOUT = 2.5V FCB = INTVCC FIGURE 9 CIRCUIT 1778 G01 Efficiency vs Load Current 100 DISCONTINUOUS MODE CONTINUOUS MODE 70 50 0.001 FCB = 5V FIGURE 9 CIRCUIT 0.1 0.01 1 LOAD CURRENT (A) 280 ILOAD = 1A 90 ILOAD = 10A 80 0 5 25 10 15 20 INPUT VOLTAGE (V) IOUT = 0A 240 200 30 5 Frequency vs Load Current 0 250 ITH VOLTAGE (V) 200 ∆VOUT (%) FIGURE 9 CIRCUIT 2.0 –0.1 –0.2 100 1.5 CONTINUOUS MODE 1.0 –0.3 DISCONTINUOUS MODE 0.5 50 2 4 6 LOAD CURRENT (A) 8 10 1778 G26 –0.4 25 ITH Voltage vs Load Current 2.5 FIGURE 9 CIRCUIT CONTINUOUS MODE 0 20 1778 G05 Load Regulation 0 15 10 1778 G04 300 FREQUENCY (kHz) 260 INPUT VOLTAGE (V) 1778 G03 150 IOUT = 10A 220 10 DISCONTINUOUS MODE FCB = 0V FIGURE 9 CIRCUIT 85 VIN = 10V VOUT = 2.5V EXTVCC = 5V FIGURE 9 CIRCUIT 60 Frequency vs Input Voltage 300 FREQUENCY (kHz) 80 1778 G19 VIN = 15V VOUT = 2.5V RLOAD = 0.25Ω 95 EFFICIENCY (%) 90 50ms/DIV 1778 G02 Efficiency vs Input Voltage 100 EFFICIENCY (%) IL 5A/DIV 0 2 6 4 LOAD CURRENT (A) 8 10 1778 G06 0 0 10 5 LOAD CURRENT (A) 15 1778 G07 1778fb 4 LTC1778/LTC1778-1 U W TYPICAL PERFOR A CE CHARACTERISTICS Current Sense Threshold vs ITH Voltage VRNG = 200 On-Time vs ION Current On-Time vs VON Voltage 10k 2V VVON = 0V 1k ON-TIME (ns) 0 100 400 200 10 0 1.0 1.5 2.0 ITH VOLTAGE (V) 0.5 2.5 10 ION CURRENT (µA) 1 3.0 200 150 100 50 50 25 75 0 TEMPERATURE (°C) Maximum Current Sense Threshold vs VRNG Voltage 100 150 MAXIMUM CURRENT SENSE THRESHOLD (mV) MAXIMUM CURRENT SENSE THRESHOLD (mV) 250 VRNG = 1V 125 100 75 50 25 0 0 125 0.2 0.4 VFB (V) 0.6 Maximum Current Sense Threshold vs RUN/SS Voltage 100 75 50 25 0 2.5 3 RUN/SS VOLTAGE (V) 200 150 100 50 0 3.5 1778 G23 0.5 0.75 1.0 1.25 1.5 VRNG VOLTAGE (V) 1.75 150 Feedback Reference Voltage vs Temperature 0.82 VRNG = 1V 140 130 120 110 100 –50 –25 2.0 1778 G10 FEEDBACK REFERENCE VOLTAGE (V) MAXIMUM CURRENT SENSE THRESHOLD (mV) 125 2 250 Maximum Current Sense Threshold vs Temperature VRNG = 1V 1.5 0.8 300 1778 G09 1778 G22 150 3 1778 G21 Current Limit Foldback IION = 30µA VVON = 0V 2 1 VON VOLTAGE (V) 0 1778 G20 On-Time vs Temperature 0 –50 –25 0 100 1778 G08 ON-TIME (ns) 600 –100 300 IION = 30µA 800 0.7V 0.5V 100 –200 MAXIMUM CURRENT SENSE THRESHOLD (mV) 1000 1.4V 1V ON-TIME (ns) CURRENT SENSE THRESHOLD (mV) 300 50 25 0 75 TEMPERATURE (°C) 100 125 1778 G11 0.81 0.80 0.79 0.78 –50 –25 75 0 25 50 TEMPERATURE (°C) 100 125 1778 G12 1778fb 5 LTC1778/LTC1778-1 U W TYPICAL PERFOR A CE CHARACTERISTICS Input and Shutdown Currents vs Input Voltage 2.0 1200 60 EXTVCC OPEN INPUT CURRENT (µA) 1.6 1.4 1.2 50 800 40 SHUTDOWN 600 30 400 20 200 SHUTDOWN CURRENT (µA) 1000 1.8 gm (mS) INTVCC Load Regulation 0 –0.1 ∆INTVCC (%) Error Amplifier gm vs Temperature –0.2 –0.3 –0.4 10 EXTVCC = 5V 1.0 –50 –25 0 50 25 0 75 TEMPERATURE (°C) 100 125 20 15 25 10 INPUT VOLTAGE (V) 5 30 1778 G13 35 8 –0.25 2 0 –50 FCB PIN CURRENT (µA) 3 FCB PIN CURRENT (µA) 0 2 –0.50 –0.75 –1.00 50 25 0 75 TEMPERATURE (°C) 100 125 –1.50 –50 –25 50 25 75 0 TEMPERATURE (°C) 1778 G14 UNDERVOLTAGE LOCKOUT THRESHOLD (V) RUN/SS THRESHOLD (V) 4.5 LATCHOFF ENABLE 4.0 3.5 LATCHOFF THRESHOLD 75 0 25 50 TEMPERATURE (°C) PULL-UP CURRENT 100 125 –2 –50 –25 50 25 0 75 TEMPERATURE (°C) 100 125 1778 G16 Undervoltage Lockout Threshold vs Temperature 5.0 –25 0 1778 G15 RUN/SS Latchoff Thresholds vs Temperature 3.0 –50 PULL-DOWN CURRENT 1 –1 –1.25 –25 50 RUN/SS Pin Current vs Temperature 10 4 10 30 40 20 INTVCC LOAD CURRENT (mA) 1778 G25 FCB Pin Current vs Temperature 6 0 1778 G24 EXTVCC Switch Resistance vs Temperature EXTVCC SWITCH RESISTANCE (Ω) –0.5 0 0 100 125 1778 G17 4.0 3.5 3.0 2.5 2.0 –50 –25 75 0 25 50 TEMPERATURE (C) 100 125 1778 G18 1778fb 6 LTC1778/LTC1778-1 U U U PI FU CTIO S RUN/SS (Pin 1): Run Control and Soft-Start Input. A capacitor to ground at this pin sets the ramp time to full output current (approximately 3s/µF) and the time delay for overcurrent latchoff (see Applications Information). Forcing this pin below 0.8V shuts down the device. PGOOD (Pin 2, LTC1778): Power Good Output. Open drain logic output that is pulled to ground when the output voltage is not within ±7.5% of the regulation point. ION (Pin 7): On-Time Current Input. Tie a resistor from VIN to this pin to set the one-shot timer current and thereby set the switching frequency. VFB (Pin 8): Error Amplifier Feedback Input. This pin connects the error amplifier input to an external resistive divider from VOUT. EXTVCC (Pin 9): External VCC Input. When EXTVCC exceeds 4.7V, an internal switch connects this pin to INTVCC and shuts down the internal regulator so that controller and gate drive power is drawn from EXTVCC. Do not exceed 7V at this pin and ensure that EXTVCC < VIN. VON (Pin 2, LTC1778-1): On-Time Voltage Input. Voltage trip point for the on-time comparator. Tying this pin to the output voltage or an external resistive divider from the output makes the on-time proportional to VOUT. The comparator input defaults to 0.7V when the pin is grounded or unavailable (LTC1778) and defaults to 2.4V when the pin is tied to INTVCC. Tie this pin to INTVCC in high VOUT applications to use a lower RON value. VIN (Pin 10): Main Input Supply. Decouple this pin to PGND with an RC filter (1Ω, 0.1µF). VRNG (Pin 3): Sense Voltage Range Input. The voltage at this pin is ten times the nominal sense voltage at maximum output current and can be set from 0.5V to 2V by a resistive divider from INTVCC. The nominal sense voltage defaults to 70mV when this pin is tied to ground, 140mV when tied to INTVCC. BG (Pin 12): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC. FCB (Pin 4): Forced Continuous Input. Tie this pin to ground to force continuous synchronous operation at low load, to INTVCC to enable discontinuous mode operation at low load or to a resistive divider from a secondary output when using a secondary winding. ITH (Pin 5): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V with 0.8V corresponding to zero sense voltage (zero current). SGND (Pin 6): Signal Ground. All small-signal components and compensation components should connect to this ground, which in turn connects to PGND at one point. INTVCC (Pin 11): Internal 5V Regulator Output. The driver and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 4.7µF low ESR tantalum capacitor. PGND (Pin 13): Power Ground. Connect this pin closely to the source of the bottom N-channel MOSFET, the (–) terminal of CVCC and the (–) terminal of CIN. SW (Pin 14): Switch Node. The (–) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below ground up to VIN. TG (Pin 15): Top Gate Drive. Drives the top N-channel MOSFET with a voltage swing equal to INTVCC superimposed on the switch node voltage SW. BOOST (Pin 16): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below INTVCC up to VIN + INTVCC. 1778fb 7 LTC1778/LTC1778-1 W FU CTIO AL DIAGRA U U RON VIN 2 VON** 7 ION 4 FCB 10 VIN 9 EXTVCC + 4.7V 0.7V CIN 2.4V + 1µA – 0.8V REF 1 0.8V 5V REG + – F tON = 16 VVON (10pF) IION R S TG Q FCNT 14 L1 SWITCH LOGIC IREV VOUT DB – – M1 SW + ICMP CB 15 ON 20k + BOOST INTVCC 11 SHDN 1.4V BG OV 12 + COUT CVCC M2 VRNG PGND 3 × 13 PGOOD* 0.7V 2 3.3µA R2 1 240k + 1V Q2 Q4 – Q6 ITHB 0.74V UV VFB 8 Q3 Q1 R1 + Q5 SGND 6 OV + – – 0.8V – SS + RUN SHDN 1.2µA EA + – – ×4 6V + 0.6V 0.8V *LTC1778 **LTC1778-1 0.86V 5 ITH RC CC1 0.6V 1 RUN/SS CSS 1778 FD 1778fb 8 LTC1778/LTC1778-1 U OPERATIO Main Control Loop The LTC1778 is a current mode controller for DC/DC step-down converters. In normal operation, the top MOSFET is turned on for a fixed interval determined by a one-shot timer OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator ICMP trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by sensing the voltage between the PGND and SW pins using the bottom MOSFET on-resistance . The voltage on the ITH pin sets the comparator threshold corresponding to inductor valley current. The error amplifier EA adjusts this voltage by comparing the feedback signal VFB from the output voltage with an internal 0.8V reference. If the load current increases, it causes a drop in the feedback voltage relative to the reference. The ITH voltage then rises until the average inductor current again matches the load current. At low load currents, the inductor current can drop to zero and become negative. This is detected by current reversal comparator IREV which then shuts off M2, resulting in discontinuous operation. Both switches will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.8V) to initiate another cycle. Discontinuous mode operation is disabled by comparator F when the FCB pin is brought below 0.8V, forcing continuous synchronous operation. The operating frequency is determined implicitly by the top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an ontime that is proportional to the ideal duty cycle, thus holding frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor RON. Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage exits a ±7.5% window around the regulation point. Furthermore, in an overvoltage condition, M1 is turned off and M2 is turned on and held on until the overvoltage condition clears. Foldback current limiting is provided if the output is shorted to ground. As VFB drops, the buffered current threshold voltage ITHB is pulled down by clamp Q3 to a 1V level set by Q4 and Q6. This reduces the inductor valley current level to one sixth of its maximum value as VFB approaches 0V. Pulling the RUN/SS pin low forces the controller into its shutdown state, turning off both M1 and M2. Releasing the pin allows an internal 1.2µA current source to charge up an external soft-start capacitor CSS. When this voltage reaches 1.5V, the controller turns on and begins switching, but with the ITH voltage clamped at approximately 0.6V below the RUN/SS voltage. As CSS continues to charge, the soft-start current limit is removed. INTVCC/EXTVCC Power Power for the top and bottom MOSFET drivers and most of the internal controller circuitry is derived from the INTVCC pin. The top MOSFET driver is powered from a floating bootstrap capacitor CB. This capacitor is recharged from INTVCC through an external Schottky diode DB when the top MOSFET is turned off. When the EXTVCC pin is grounded, an internal 5V low dropout regulator supplies the INTVCC power from VIN. If EXTVCC rises above 4.7V, the internal regulator is turned off, and an internal switch connects EXTVCC to INTVCC. This allows a high efficiency source connected to EXTVCC, such as an external 5V supply or a secondary output from the converter, to provide the INTVCC power. Voltages up to 7V can be applied to EXTVCC for additional gate drive. If the input voltage is low and INTVCC drops below 3.5V, undervoltage lockout circuitry prevents the power switches from turning on. 1778fb 9 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO The basic LTC1778 application circuit is shown in Figure 1. External component selection is primarily determined by the maximum load current and begins with the selection of the sense resistance and power MOSFET switches. The LTC1778 uses the on-resistance of the synchronous power MOSFET for determining the inductor current. The desired amount of ripple current and operating frequency largely determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter and COUT is chosen with low enough ESR to meet the output voltage ripple and transient specification. resulting in nominal sense voltages of 50mV to 200mV. Additionally, the VRNG pin can be tied to SGND or INTVCC in which case the nominal sense voltage defaults to 70mV or 140mV, respectively. The maximum allowed sense voltage is about 1.33 times this nominal value. Choosing the LTC1778 or LTC1778-1 The gate drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC1778 applications. If the input voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered. Maximum Sense Voltage and VRNG Pin Inductor current is determined by measuring the voltage across a sense resistance that appears between the PGND and SW pins. The maximum sense voltage is set by the voltage applied to the VRNG pin and is equal to approximately (0.133)VRNG. The current mode control loop will not allow the inductor current valleys to exceed (0.133)VRNG/RSENSE. In practice, one should allow some margin for variations in the LTC1778 and external component values and a good guide for selecting the sense resistance is: RSENSE = VRNG 10 • IOUT(MAX) An external resistive divider from INTVCC can be used to set the voltage of the VRNG pin between 0.5V and 2V The LTC1778 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the breakdown voltage V(BR)DSS, threshold voltage V(GS)TH, on-resistance RDS(ON), reverse transfer capacitance CRSS and maximum current IDS(MAX). When the bottom MOSFET is used as the current sense element, particular attention must be paid to its onresistance. MOSFET on-resistance is typically specified with a maximum value RDS(ON)(MAX) at 25°C. In this case, additional margin is required to accommodate the rise in MOSFET on-resistance with temperature: RDS(ON)(MAX) = RSENSE ρT The ρT term is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance 2.0 ρT NORMALIZED ON-RESISTANCE The LTC1778 has an open-drain PGOOD output that indicates when the output voltage is within ±7.5% of the regulation point. The LTC1778-1 trades the PGOOD pin for a VON pin that allows the on-time to be adjusted. Tying the VON pin high results in lower values for RON which is useful in high VOUT applications. The VON pin also provides a means to adjust the on-time to maintain constant frequency operation in applications where VOUT changes and to correct minor frequency shifts with changes in load current. Finally, the VON pin can be used to provide additional current limiting in positive-to-negative converters and as a control input to synchronize the switching frequency with a phase locked loop. Power MOSFET Selection 1.5 1.0 0.5 0 – 50 50 100 0 JUNCTION TEMPERATURE (°C) 150 1778 F02 Figure 2. RDS(ON) vs. Temperature 1778fb 10 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO The power dissipated by the top and bottom MOSFETs strongly depends upon their respective duty cycles and the load current. When the LTC1778 is operating in continuous mode, the duty cycles for the MOSFETs are: VOUT VIN V –V = IN OUT VIN DTOP = DBOT The resulting power dissipation in the MOSFETs at maximum output current are: PTOP = DTOP IOUT(MAX)2 ρT(TOP) RDS(ON)(MAX) + k VIN2 IOUT(MAX) CRSS f PBOT = DBOT IOUT(MAX)2 ρT(BOT) RDS(ON)(MAX) Both MOSFETs have I2R losses and the top MOSFET includes an additional term for transition losses, which are largest at high input voltages. The constant k = 1.7A–1 can be used to estimate the amount of transition loss. The bottom MOSFET losses are greatest when the bottom duty cycle is near 100%, during a short-circuit or at high input voltage. Tying a resistor RON from VIN to the ION pin yields an ontime inversely proportional to VIN. For a step-down converter, this results in approximately constant frequency operation as the input supply varies: f= VOUT [ HZ ] VVON RON(10pF) To hold frequency constant during output voltage changes, tie the VON pin to VOUT or to a resistive divider from VOUT when VOUT > 2.4V. The VON pin has internal clamps that limit its input to the one-shot timer. If the pin is tied below 0.7V, the input to the one-shot is clamped at 0.7V. Similarly, if the pin is tied above 2.4V, the input is clamped at 2.4V. In high VOUT applications, tying VON to INTVCC so that the comparator input is 2.4V results in a lower value for RON. Figures 3a and 3b show how RON relates to switching frequency for several common output voltages. 1000 SWITCHING FREQUENCY (kHz) with temperature, typically about 0.4%/°C as shown in Figure 2. For a maximum junction temperature of 100°C, using a value ρT = 1.3 is reasonable. VOUT = 3.3V VOUT = 1.5V 100 100 1000 RON (kΩ) Operating Frequency The operating frequency of LTC1778 applications is determined implicitly by the one-shot timer that controls the on-time tON of the top MOSFET switch. The on-time is set by the current into the ION pin and the voltage at the VON pin (LTC1778-1) according to: V tON = VON (10pF ) IION VON defaults to 0.7V in the LTC1778. 10000 1778 F03a Figure 3a. Switching Frequency vs RON for the LTC1778 and LTC1778-1 (VON = 0V) 1000 SWITCHING FREQUENCY (kHz) The choice of operating frequency is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance in order to maintain low output ripple voltage. VOUT = 2.5V VOUT = 12V VOUT = 5V VOUT = 3.3V 100 100 1000 RON (kΩ) 10000 1778 F03b Figure 3b. Switching Frequency vs RON for the LTC1778-1 (VON = INTVCC) 1778fb 11 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO Because the voltage at the ION pin is about 0.7V, the current into this pin is not exactly inversely proportional to VIN, especially in applications with lower input voltages. To correct for this error, an additional resistor RON2 connected from the ION pin to the 5V INTVCC supply will further stabilize the frequency. VIN(MIN) = VOUT tON + tOFF(MIN) tON A plot of maximum duty cycle vs frequency is shown in Figure 5. 5V RON 0.7 V Changes in the load current magnitude will also cause frequency shift. Parasitic resistance in the MOSFET switches and inductor reduce the effective voltage across the inductance, resulting in increased duty cycle as the load current increases. By lengthening the on-time slightly as current increases, constant frequency operation can be maintained. This is accomplished with a resistive divider from the ITH pin to the VON pin and VOUT. The values required will depend on the parasitic resistances in the specific application. A good starting point is to feed about 25% of the voltage change at the ITH pin to the VON pin as shown in Figure 4a. Place capacitance on the VON pin to filter out the ITH variations at the switching frequency. The resistor load on ITH reduces the DC gain of the error amp and degrades load regulation, which can be avoided by using the PNP emitter follower of Figure 4b. Minimum Off-time and Dropout Operation The minimum off-time tOFF(MIN) is the smallest amount of time that the LTC1778 is capable of turning on the bottom MOSFET, tripping the current comparator and turning the MOSFET back off. This time is generally about 250ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, Inductor Selection Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: ⎛ V ⎞⎛ V ⎞ ∆IL = ⎜ OUT ⎟ ⎜ 1 − OUT ⎟ VIN ⎠ ⎝ f L ⎠⎝ Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage 2.0 SWITCHING FREQUENCY (MHz) RON2 = due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: 1.5 DROPOUT REGION 1.0 0.5 0 0 1.0 0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN) 1778 F05 Figure 5. Maximum Switching Frequency vs Duty Cycle RVON1 30k RVON1 3k VON VOUT CVON 0.01µF RVON2 100k LTC1778 RC ITH VOUT 10k CVON 0.01µF RVON2 10k INTVCC LTC1778 RC Q1 2N5087 ITH CC CC (4a) VON 1778 F04 (4b) Figure 4. Correcting Frequency Shift with Load Current Changes 1778fb 12 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a tradeoff between component size, efficiency and operating frequency. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). The largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to: ⎛ VOUT ⎞ ⎛ VOUT ⎞ L=⎜ 1 − ⎟⎜ ⎟ ⎝ f ∆IL(MAX) ⎠ ⎝ VIN(MAX) ⎠ Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko. Schottky Diode D1 Selection The Schottky diode D1 shown in Figure 1 conducts during the dead time between the conduction of the power MOSFET switches. It is intended to prevent the body diode of the bottom MOSFET from turning on and storing charge during the dead time, which can cause a modest (about 1%) efficiency loss. The diode can be rated for about one half to one fifth of the full load current since it is on for only a fraction of the duty cycle. In order for the diode to be effective, the inductance between it and the bottom MOSFET must be as small as possible, mandating that these components be placed adjacently. The diode can be omitted if the efficiency loss is tolerable. CIN and COUT Selection The input capacitance CIN is required to filter the square wave current at the drain of the top MOSFET. Use a low ESR capacitor sized to handle the maximum RMS current. IRMS ≅ IOUT(MAX) VOUT VIN VIN –1 VOUT This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT(MAX) / 2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor. The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The output ripple ∆VOUT is approximately bounded by: ⎛ 1 ⎞ ∆VOUT ≤ ∆IL ⎜ ESR + ⎟ 8 fCOUT ⎠ ⎝ Since ∆IL increases with input voltage, the output ripple is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications providing that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. To dampen input voltage transients, add a small 5µF to 50µF aluminum electrolytic capacitor with an ESR in the range of 0.5Ω to 2Ω. High performance through-hole capacitors may also be used, Kool Mµ is a registered trademark of Magnetics, Inc. 1778fb 13 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO but an additional ceramic capacitor in parallel is recommended to reduce the effect of their lead inductance. Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor CB connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from INTVCC when the switch node is low. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + INTVCC. The boost capacitor needs to store about 100 times the gate charge required by the top MOSFET. In most applications 0.1µF to 0.47µF, X5R or X7R dielectric capacitor is adequate. Discontinuous Mode Operation and FCB Pin The FCB pin determines whether the bottom MOSFET remains on when current reverses in the inductor. Tying this pin above its 0.8V threshold enables discontinuous operation where the bottom MOSFET turns off when inductor current reverses. The load current at which current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and will vary with changes in VIN. Tying the FCB pin below the 0.8V threshold forces continuous synchronous operation, allowing current to reverse at light loads and maintaining high frequency operation. In addition to providing a logic input to force continuous operation, the FCB pin provides a means to maintain a flyback winding output when the primary is operating in discontinuous mode. The secondary output VOUT2 is normally set as shown in Figure 6 by the turns ratio N of the + VIN CIN VIN 1N4148 OPTIONAL EXTVCC CONNECTION 5V < VOUT2 < 7V TG • + LTC1778 EXTVCC SW R4 T1 1:N FCB R3 • + VOUT2 COUT2 1µF VOUT1 ⎛ R4 ⎞ VOUT 2(MIN) = 0.8V⎜ 1 + ⎟ ⎝ R3 ⎠ Fault Conditions: Current Limit and Foldback The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC1778, the maximum sense voltage is controlled by the voltage on the VRNG pin. With valley current control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limit is: ILIMIT = VSNS(MAX) RDS(ON) 1 + ∆IL ρT 2 The current limit value should be checked to ensure that ILIMIT(MIN) > IOUT(MAX). The minimum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the largest power loss in the converter. Note that it is important to check for self-consistency between the assumed MOSFET junction temperature and the resulting value of ILIMIT which heats the MOSFET switches. Caution should be used when setting the current limit based upon the RDS(ON) of the MOSFETs. The maximum current limit is determined by the minimum MOSFET onresistance. Data sheets typically specify nominal and maximum values for RDS(ON), but not a minimum. A reasonable assumption is that the minimum RDS(ON) lies the same amount below the typical value as the maximum lies above it. Consult the MOSFET manufacturer for further guidelines. COUT BG SGND transformer. However, if the controller goes into discontinuous mode and halts switching due to a light primary load current, then VOUT2 will droop. An external resistor divider from VOUT2 to the FCB pin sets a minimum voltage VOUT2(MIN) below which continuous operation is forced until VOUT2 has risen above its minimum. PGND 1778 F06 Figure 6. Secondary Output Loop and EXTVCC Connection To further limit current in the event of a short circuit to ground, the LTC1778 includes foldback current limiting. If the output falls by more than 25%, then the maximum sense voltage is progressively lowered to about one sixth of its full value. 1778fb 14 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO INTVCC Regulator An internal P-channel low dropout regulator produces the 5V supply that powers the drivers and internal circuitry within the LTC1778. The INTVCC pin can supply up to 50mA RMS and must be bypassed to ground with a minimum of 4.7µF low ESR tantalum capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. Applications using large MOSFETs with a high input voltage and high frequency of operation may cause the LTC1778 to exceed its maximum junction temperature rating or RMS current rating. Most of the supply current drives the MOSFET gates unless an external EXTVCC source is used. In continuous mode operation, this current is IGATECHG = f(Qg(TOP) + Qg(BOT)). The junction temperature can be estimated from the equations given in Note 2 of the Electrical Characteristics. For example, the LTC1778CGN is limited to less than 14mA from a 30V supply: will start-up using the internal linear regulator until the boosted output supply is available. External Gate Drive Buffers The LTC1778 drivers are adequate for driving up to about 30nC into MOSFET switches with RMS currents of 50mA. Applications with larger MOSFET switches or operating at frequencies requiring greater RMS currents will benefit from using external gate drive buffers such as the LTC1693. Alternately, the external buffer circuit shown in Figure 7 can be used. Note that the bipolar devices reduce the signal swing at the MOSFET gate, and benefit from an increased EXTVCC voltage of about 6V. 10Ω TG TJ = 70°C + (14mA)(30V)(130°C/W) = 125°C For larger currents, consider using an external supply with the EXTVCC pin. INTVCC BOOST Q1 FMMT619 GATE OF M1 Q2 FMMT720 10Ω BG Q3 FMMT619 GATE OF M2 Q4 FMMT720 PGND SW 1778 F07 Figure 7. Optional External Gate Driver EXTVCC Connection The EXTVCC pin can be used to provide MOSFET gate drive and control power from the output or another external source during normal operation. Whenever the EXTVCC pin is above 4.7V the internal 5V regulator is shut off and an internal 50mA P-channel switch connects the EXTVCC pin to INTVCC. INTVCC power is supplied from EXTVCC until this pin drops below 4.5V. Do not apply more than 7V to the EXTVCC pin and ensure that EXTVCC ≤ VIN. The following list summarizes the possible connections for EXTVCC: 1. EXTVCC grounded. INTVCC is always powered from the internal 5V regulator. 2. EXTVCC connected to an external supply. A high efficiency supply compatible with the MOSFET gate drive requirements (typically 5V) can improve overall efficiency. 3. EXTVCC connected to an output derived boost network. The low voltage output can be boosted using a charge pump or flyback winding to greater than 4.7V. The system Soft-Start and Latchoff with the RUN/SS Pin The RUN/SS pin provides a means to shut down the LTC1778 as well as a timer for soft-start and overcurrent latchoff. Pulling the RUN/SS pin below 0.8V puts the LTC1778 into a low quiescent current shutdown (IQ < 30µA). Releasing the pin allows an internal 1.2µA current source to charge up the external timing capacitor CSS. If RUN/SS has been pulled all the way to ground, there is a delay before starting of about: tDELAY = 1.5V CSS = 1.3s/µF CSS 1.2µA ( ) When the voltage on RUN/SS reaches 1.5V, the LTC1778 begins operating with a clamp on ITH of approximately 0.9V. As the RUN/SS voltage rises to 3V, the clamp on ITH is raised until its full 2.4V range is available. This takes an additional 1.3s/µF, during which the load current is folded back until the output reaches 75% of its final value. The pin can be driven from logic as shown in Figure 7. Diode D1 1778fb 15 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO reduces the start delay while allowing CSS to charge up slowly for the soft-start function. After the controller has been started and given adequate time to charge up the output capacitor, CSS is used as a short-circuit timer. After the RUN/SS pin charges above 4V, if the output voltage falls below 75% of its regulated value, then a short-circuit fault is assumed. A 1.8µA current then begins discharging CSS. If the fault condition persists until the RUN/SS pin drops to 3.5V, then the controller turns off both power MOSFETs, shutting down the converter permanently. The RUN/SS pin must be actively pulled down to ground in order to restart operation. The overcurrent protection timer requires that the soft-start timing capacitor CSS be made large enough to guarantee that the output is in regulation by the time CSS has reached the 4V threshold. In general, this will depend upon the size of the output capacitance, output voltage and load current characteristic. A minimum soft-start capacitor can be estimated from: CSS > COUT VOUT RSENSE (10 – 4 [F/V s]) Generally 0.1µF is more than sufficient. Overcurrent latchoff operation is not always needed or desired. Load current is already limited during a shortcircuit by the current foldback circuitry and latchoff operation can prove annoying during troubleshooting. The feature can be overridden by adding a pull-up current greater than 5µA to the RUN/SS pin. The additional current prevents the discharge of CSS during a fault and also shortens the soft-start period. Using a resistor to V IN as shown in Figure 8a is simple, but slightly increases shutdown current. Connecting a resistor to INTV CC as INTVCC RSS* VIN 3.3V OR 5V D1 RUN/SS RSS* D2* RUN/SS 2N7002 CSS CSS 1778 F08 *OPTIONAL TO OVERRIDE OVERCURRENT LATCHOFF (8a) (8b) Figure 8. RUN/SS Pin Interfacing with Latchoff Defeated shown in Figure 8b eliminates the additional shutdown current, but requires a diode to isolate CSS . Any pull-up network must be able to pull RUN/SS above the 4.2V maximum threshold of the latchoff circuit and overcome the 4µA maximum discharge current. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC1778 circuits: 1. DC I2R losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows through L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if RDS(ON) = 0.01Ω and RL = 0.005Ω, the loss will range from 15mW to 1.5W as the output current varies from 1A to 10A. 2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss ≅ (1.7A–1) VIN2 IOUT CRSS f 3. INTVCC current. This is the sum of the MOSFET driver and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high efficiency source, such as an output derived boost network or alternate supply if available. 4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. 1778fb 16 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO Other losses, including COUT ESR loss, Schottky diode D1 conduction loss during dead time and inductor core loss generally account for less than 2% additional loss. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current, then there is no change in efficiency. Checking Transient Response Selecting a standard value of 1.8µH results in a maximum ripple current of: ∆IL = ⎛ 2.5V ⎞ ⎜1– ⎟ = 5.1A 28V ⎠ 250kHz 1.8µH ⎝ 2.5V ( )( ) Next, choose the synchronous MOSFET switch. Choosing a Si4874 (RDS(ON) = 0.0083Ω (NOM) 0.010Ω (MAX), θJA = 40°C/W) yields a nominal sense voltage of: VSNS(NOM) = (10A)(1.3)(0.0083Ω) = 108mV The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ∆ILOAD (ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in Figure 9 will provide adequate compensation for most applications. For a detailed explanation of switching control loop theory see Application Note 76. Tying VRNG to 1.1V will set the current sense voltage range for a nominal value of 110mV with current limit occurring at 146mV. To check if the current limit is acceptable, assume a junction temperature of about 80°C above a 70°C ambient with ρ150°C = 1.5: Design Example Because the top MOSFET is on for such a short time, an Si4884 RDS(ON)(MAX) = 0.0165Ω, CRSS = 100pF, θJA = 40°C/W will be sufficient. Checking its power dissipation at current limit with ρ100°C = 1.4: As a design example, take a supply with the following specifications: VIN = 7V to 28V (15V nominal), VOUT = 2.5V ±5%, IOUT(MAX) = 10A, f = 250kHz. First, calculate the timing resistor with VON = VOUT: RON = 2.5V = 1.42MΩ (0.7V)(250kHz)(10pF ) and choose the inductor for about 40% ripple current at the maximum VIN: ⎛ 2.5V ⎞ L= ⎜ 1− ⎟ = 2.3µH 28V ⎠ 250kHz 0.4 10A ⎝ 2.5V ( )( )( ) ILIMIT ≥ 146mV 1 5.1A = 12A 2 (1.5)(0.010Ω) ( ) + and double check the assumed TJ in the MOSFET: PBOT = 28V – 2 .5V 12A 28V 2 ( ) (1.5)(0.010Ω) = 1.97 W TJ = 70°C + (1.97W)(40°C/W) = 149°C 2.5V 12A 28V 2 ( ) (1.4)(0.0165Ω) + 2 (1.7)(28V) (12A)(100pF )(250kHz) PTOP = = 0.30W + 0.40W = 0.7 W TJ = 70°C + (0.7W)(40°C/W) = 98°C The junction temperatures will be significantly less at nominal current, but this analysis shows that careful attention to heat sinking will be necessary in this circuit. 1778fb 17 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO CIN is chosen for an RMS current rating of about 5A at 85°C. The output capacitors are chosen for a low ESR of 0.013Ω to minimize output voltage changes due to inductor ripple current and load steps. The ripple voltage will be only: ∆VOUT(RIPPLE) = ∆IL(MAX) (ESR) = (5.1A) (0.013Ω) = 66mV However, a 0A to 10A load step will cause an output change of up to: ∆VOUT(STEP) = ∆ILOAD (ESR) = (10A) (0.013Ω) = 130mV An optional 22µF ceramic output capacitor is included to minimize the effect of ESL in the output ripple. The complete circuit is shown in Figure 9. PC Board Layout Checklist When laying out a PC board follow one of the two suggested approaches. The simple PC board layout requires a dedicated ground plane layer. Also, for higher currents, it is recommended to use a multilayer board to help with heat sinking power components. 1 R3 11k R4 39k RPG 100k 2 3 CC1 500pF 4 RC 20k CC2 100pF 5 6 7 R1 14.0k R2 30.1k 8 C2 6.8nF LTC1778 RUN/SS BOOST PGOOD TG VRNG SW FCB ITH SGND 16 15 • Place CIN, COUT, MOSFETs, D1 and inductor all in one compact area. It may help to have some components on the bottom side of the board. • Place LTC1778 chip with pins 9 to 16 facing the power components. Keep the components connected to pins 1 to 8 close to LTC1778 (noise sensitive components). • Use an immediate via to connect the components to ground plane including SGND and PGND of LTC1778. Use several bigger vias for power components. • Use compact plane for switch node (SW) to improve cooling of the MOSFETs and to keep EMI down. • Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. • Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power component. You can connect the copper areas to any DC net (VIN, VOUT, GND or to any other DC rail in your system). DB CMDSH-3 CB 0.22µF M1 Si4884 L1 1.8µH + PGND BG INTVCC ION VIN VFB EXTVCC VIN 5V TO 28V CIN 10µF 35V ×3 14 13 M2 Si4874 D1 B340A COUT1-2 180µF 4V ×2 COUT3 22µF 6.3V X7R VOUT 2.5V 10A 12 11 + CSS 0.1µF • The ground plane layer should not have any traces and it should be as close as possible to the layer with power MOSFETs. CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 1.4MΩ 1778 F09 CIN: UNITED CHEMICON THCR60EIHI06ZT COUT1-2: CORNELL DUBILIER ESRE181E04B L1: SUMIDA CEP125-1R8MC-H Figure 9. Design Example: 2.5V/10A at 250kHz 1778fb 18 LTC1778/LTC1778-1 U W U U APPLICATIO S I FOR ATIO When laying out a printed circuit board, without a ground plane, use the following checklist to ensure proper operation of the controller. These items are also illustrated in Figure 10. • Segregate the signal and power grounds. All small signal components should return to the SGND pin at one point which is then tied to the PGND pin close to the source of M2. • Place M2 as close to the controller as possible, keeping the PGND, BG and SW traces short. • Connect the input capacitor(s) CIN close to the power MOSFETs. This capacitor carries the MOSFET AC current. • Keep the high dV/dT SW, BOOST and TG nodes away from sensitive small-signal nodes. • Connect the INTVCC decoupling capacitor CVCC closely to the INTVCC and PGND pins. • Connect the top driver boost capacitor CB closely to the BOOST and SW pins. • Connect the VIN pin decoupling capacitor CF closely to the VIN and PGND pins. CSS 2 3 4 RUN/SS BOOST PGOOD TG VRNG SW FCB PGND 15 DB 14 5 BG ITH + M1 13 D1 RC CC2 12 CIN VIN M2 CVCC 6 7 SGND INTVCC ION VIN 8 VFB EXTVCC – 11 10 R1 R2 L 16 + CC1 CB LTC1778 1 9 – VOUT COUT CF + RF RON BOLD LINES INDICATE HIGH CURRENT PATHS 1778 F10 Figure 10. LTC1778 Layout Diagram 1778fb 19 LTC1778/LTC1778-1 U TYPICAL APPLICATIO S 1.5V/10A at 300kHz from 3.3V Input CSS 0.1µF 1 RR2 39k RR1 11k RPG 100k 2 3 CC1 680pF 4 RC 20k 5 CC2 100pF 6 7 R1 10k 8 LTC1778 RUN/SS BOOST PGOOD TG VRNG SW DB CMDSH-3 16 CB 0.22µF 15 CIN1-2 22µF 6.3V ×2 M1 IRF7811A 14 L1, 0.68µH + PGND FCB BG ITH SGND INTVCC ION VIN VFB EXTVCC 13 M2 IRF7811A D1 B320B COUT 270µF 2V ×2 + VIN 3.3V CIN3 330µF 6.3V VOUT 1.5V 10A 12 CVCC 4.7µF 11 10 5V 9 RON 576k R2 8.87k 1778 TA01 CIN1-2: MURATA GRM42-2X5R226K6.3 COUT: CORNELL DUBILIER ESRE271M02B 1.2V/6A at 300kHz CSS 0.1µF 1 RPG 100k 2 3 CC1 470pF 4 RC 20k 5 CC2 100pF 6 7 R1 20k R2 10k 8 C2 2200pF LTC1778 RUN/SS BOOST PGOOD TG VRNG FCB SW PGND BG ITH SGND INTVCC ION VIN VFB EXTVCC 16 15 DB CMDSH-3 CB 0.22µF 14 M1 1/2 FDS6982S L1 1.8µH 13 CIN 10µF 25V ×2 M2 1/2 FDS6982S + COUT1 180µF 2V VIN 5V TO 25V VOUT 1.2V 6A COUT2 10µF 6.3V 12 11 CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 510k 1778 TA02 CIN: TAIYO YUDEN TMK432BJ106MM COUT1: CORNELL DUBILIER ESRD181M02B COUT2: TAIYO YUDEN JMK316BJ106ML L1: TOKO 919AS-1R8N 1778fb 20 LTC1778/LTC1778-1 U TYPICAL APPLICATIO S Single Inductor, Positive Output Buck/Boost CSS 0.1µF 1 2 3 4 CC1 1nF RC 47k 5 CC2 220pF 6 7 C1 100pF 8 LTC1778-1 RUN/SS BOOST VON TG VRNG SW DB CMDSH-3 16 CB 0.22µF 15 CIN 22µF D2 50V IR 12CWQ03FN ×2 M1 IRF7811A 14 PGND BG ITH SGND INTVCC ION VIN VFB EXTVCC 13 M2 IRF7811A COUT 100µF 20V ×6 M3 Si4888 12 CVCC 4.7µF 11 D1 B340A RF 1Ω 10 CF 0.1µF 9 PGND CIN: MARCON THER70EIH226ZT COUT: AVX TPSV107M020R0085 L1: SCHOTT 36835-1 RON1 1.5M 1% RON2 1.5M 1% R2 140k 1% VIN 6V TO 18V VOUT 12V L1 4.8µH + FCB R1 10k 1% VIN IOUT 18V 6A 12V 5A 6V 3.3A 1778 TA04 12V/5A at 300kHz LTC1778-1 16 1 RUN/SS BOOST 2 3 CC1 2.2nF 4 RC 20k 5 CC2 100pF 6 7 R1 10k R2 140k 8 C2 2200pF VON TG VRNG SW FCB ITH SGND ION VFB 15 DB CMDSH-3 CB 0.22µF 14 CIN 22µF 50V M1 L1 10µH + PGND BG INTVCC VIN EXTVCC 13 M2 D1 VIN 14V TO 28V VOUT 12V 5A COUT 220µF 16V 12 11 + CSS 0.1µF CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 1.6M 1778 TA05 CIN: UNITED CHEMICON THCR70E1H226ZT COUT: SANYO 16SV220M L1: SUMIDA CDRH127-100 M1, M2: FAIRCHILD FDS6680A D1: DIODES, INC. B340A (847) 696-2000 (619) 661-6835 (847) 956-0667 (408) 822-2126 (805) 446-4800 1778fb 21 LTC1778/LTC1778-1 U TYPICAL APPLICATIO S Positive-to-Negative Converter, –5V/5A at 300kHz CSS 0.1µF 1 2 3 4 CC1 4700pF RC 10k 5 CC2 100pF 6 7 R1 10k R2 52.3k 8 LTC1778-1 16 RUN/SS BOOST VON TG VRNG SW FCB ITH SGND ION VFB 15 VIN IOUT 20V 8A 10V 6.7A 5V 5A DB CMDSH-3 CB 0.22µF 14 CIN1 10µF 25V ×2 M1 IRF7811A BG INTVCC VIN EXTVCC VIN 5V TO 20V L1 2.7µH + PGND CIN2 10µF 35V 13 M2 IRF7822 D1 B340A 12 11 CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF COUT 100µF 6V ×3 VOUT –5V RON 698k 1778 TA06 CIN1: TAIYO YUDEN TMK432BJ106MM CIN2: SANYO 35CV10GX COUT: PANASONIC EEFUD0J101R L1: PANASONIC ETQPAF2R7H 1778fb 22 LTC1778/LTC1778-1 U PACKAGE DESCRIPTIO GN Package 16-Lead Plastic SSOP (Narrow 0.150) (LTC DWG # 05-08-1641) .189 – .196* (4.801 – 4.978) .045 ±.005 16 15 14 13 12 11 10 9 .254 MIN .009 (0.229) REF .150 – .165 .229 – .244 (5.817 – 6.198) .0165 ± .0015 .150 – .157** (3.810 – 3.988) .0250 BSC RECOMMENDED SOLDER PAD LAYOUT 1 .015 ± .004 × 45° (0.38 ± 0.10) .007 – .0098 (0.178 – 0.249) 2 3 4 5 6 7 .0532 – .0688 (1.35 – 1.75) 8 .004 – .0098 (0.102 – 0.249) 0° – 8° TYP .016 – .050 (0.406 – 1.270) NOTE: 1. CONTROLLING DIMENSION: INCHES INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) .008 – .012 (0.203 – 0.305) TYP .0250 (0.635) BSC GN16 (SSOP) 0204 3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 1778fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LTC1778/LTC1778-1 U TYPICAL APPLICATIO Typical Application 2.5V/3A at 1.4MHz CSS 0.1µF LTC1778 RUN/SS BOOST 1 RPG 100k 2 3 4 CC1 470pF RC 33k 8 C2 2200pF SW SGND 7 R2 24.9k VRNG 15 CB 0.22µF 14 CIN 10µF 25V M1 1/2 Si9802 L1, 1µH + PGND BG ITH 6 R1 11.5k TG FCB 5 CC2 100pF PGOOD 16 DB CMDSH-3 INTVCC ION VIN VFB EXTVCC 13 M2 1/2 Si9802 VIN 9V TO 18V VOUT 2.5V 3A COUT 120µF 4V 12 11 CVCC 4.7µF 10 RF 1Ω 9 CF 0.1µF RON 220k 1778 TA03 CIN: TAIYO YUDEN TMK432BJ106MM COUT: CORNELL DUBILIER ESRD121M04B L1: TOKO A921CY-1R0M RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1622 550kHz Step-Down Controller 8-Pin MSOP; Synchronizable; Soft-Start; Current Mode LTC1625/LTC1775 No RSENSE Current Mode Synchronous Step-Down Controller 97% Efficiency; No Sense Resistor; 16-Pin SSOP LTC1628-PG Dual, 2-Phase Synchronous Step-Down Controller Power Good Output; Minimum Input/Output Capacitors; 3.5V ≤ VIN ≤ 36V LTC1628-SYNC Dual, 2-Phase Synchronous Step-Down Controller Synchronizable 150kHz to 300kHz LTC1709-7 High Efficiency, 2-Phase Synchronous Step-Down Controller with 5-Bit VID Up to 42A Output; 0.925V ≤ VOUT ≤ 2V LTC1709-8 High Efficiency, 2-Phase Synchronous Step-Down Controller Up to 42A Output; VRM 8.4; 1.3V ≤ VOUT ≤ 3.5V LTC1735 High Efficiency, Synchronous Step-Down Controller Burst Mode® Operation; 16-Pin Narrow SSOP; 3.5V ≤ VIN ≤ 36V LTC1736 High Efficiency, Synchronous Step-Down Controller with 5-Bit VID Mobile VID; 0.925V ≤ VOUT ≤ 2V; 3.5V ≤ VIN ≤ 36V LTC1772 SOT-23 Step-Down Controller Current Mode; 550kHz; Very Small Solution Size LTC1773 Synchronous Step-Down Controller Up to 95% Efficiency, 550kHz, 2.65V ≤ VIN ≤ 8.5V, 0.8V ≤ VOUT ≤ VIN, Synchronizable to 750kHz LTC1876 2-Phase, Dual Synchronous Step-Down Controller with Step-Up Regulator 3.5V ≤ VIN ≤ 36V, Power Good Output, 300kHz Operation LTC3713 Low VIN High Current Synchronous Step-Down Controller 1.5V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)VIN, IOUT Up to 20A LTC3778 Low VOUT, No RSENSE Synchronous Step-Down Controller 0.6V ≤ VOUT ≤ (0.9)VIN, 4V ≤ VIN ≤ 36V, IOUT Up to 20A 60V Synchronous Step-Down Controller Current Mode, Output Slew Rate Control ® LT 3800 Burst Mode is a registered trademark of Linear Technology Corporation. 1778fb 24 Linear Technology Corporation LT/LT 0405 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2001