PIN CONFIGURATIONS Low offset voltage: 1 μV Input offset drift: 0.005 μV/°C Rail-to-rail input and output swing 5 V/2.7 V single-supply operation High gain, CMRR, PSRR: 130 dB Ultralow input bias current: 20 pA Low supply current: 700 μA/op amp Overload recovery time: 50 μs No external capacitors required 1 NC –IN A +IN A V– 8 NC V+ OUT A NC AD8551 4 5 NC = NO CONNECT 01101-001 FEATURES Figure 1. 8-Lead MSOP (RM Suffix) 8 NC +IN A 3 AD8551 V– 4 APPLICATIONS 7 V+ 6 OUT A 5 NC NC = NO CONNECT Temperature sensors Pressure sensors Precision current sensing Strain gage amplifiers Medical instrumentation Thermocouple amplifiers 01101-002 NC 1 –IN A 2 Figure 2. 8-Lead SOIC (R Suffix) OUT A –IN A +IN A V– 1 8 AD8552 4 5 V+ OUT B –IN B +IN B 01101-003 Data Sheet Zero-Drift, Single-Supply, Rail-to-Rail Input/Output Operational Amplifiers AD8551/AD8552/AD8554 Figure 3. 8-Lead TSSOP (RU Suffix) GENERAL DESCRIPTION With an offset voltage of only 1 μV and drift of 0.005 μV/°C, the AD8551/AD8552/AD8554 are perfectly suited for applications in which error sources cannot be tolerated. Temperature, position and pressure sensors, medical equipment, and strain gage amplifiers benefit greatly from nearly zero drift over their operating temperature range. The rail-to-rail input and output swings provided by the AD8551/AD8552/AD8554 make both high-side and low-side sensing easy. The AD8551/AD8552/AD8554 are specified for the extended industrial/automotive temperature range (−40°C to +125°C). The AD8551 single amplifier is available in 8-lead MSOP and 8-lead narrow SOIC packages. The AD8552 dual amplifier is available in 8-lead narrow SOIC and 8-lead TSSOP surfacemount packages. The AD8554 quad is available in 14-lead narrow SOIC and 14-lead TSSOP packages. Rev. F +IN A 3 8 V+ AD8552 V– 4 7 OUT B 6 –IN B 5 +IN B 01101-004 –IN A 2 Figure 4. 8-Lead SOIC (R Suffix) OUT A –IN A +IN A V+ +IN B –IN B OUT B 1 14 AD8554 7 8 OUT D –IN D +IN D V– +IN C –IN C OUT C 01101-005 The AD8551/AD8552/AD8554 provide the benefits previously found only in expensive auto-zeroing or chopper-stabilized amplifiers. Using Analog Devices, Inc. topology, these new zero-drift amplifiers combine low cost with high accuracy. No external capacitors are required. OUT A 1 Figure 5. 14-Lead TSSOP (RU Suffix) OUT A 1 14 OUT D –IN A 2 13 –IN D +IN A 3 12 +IN D V+ 4 AD8554 11 V– +IN B 5 10 +IN C –IN B 6 9 –IN C OUT B 7 8 OUT C 01101-006 This family of amplifiers has ultralow offset, drift, and bias current. The AD8551, AD8552, and AD8554 are single, dual, and quad amplifiers featuring rail-to-rail input and output swings. All are guaranteed to operate from 2.7 V to 5 V with a single supply. Figure 6. 14-Lead SOIC (R Suffix) Document Feedback Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 ©1999–2015 Analog Devices, Inc. All rights reserved. Technical Support www.analog.com AD8551/AD8552/AD8554 Data Sheet TABLE OF CONTENTS Features .............................................................................................. 1 1/f Noise Characteristics ........................................................... 16 Applications ....................................................................................... 1 Intermodulation Distortion ...................................................... 17 General Description ......................................................................... 1 Broadband and External Resistor Noise Considerations ...... 18 Pin Configurations ........................................................................... 1 Output Overdrive Recovery...................................................... 18 Revision History ............................................................................... 2 Input Overvoltage Protection ................................................... 18 Specifications..................................................................................... 3 Output Phase Reversal ............................................................... 18 Electrical Characteristics ............................................................. 3 Capacitive Load Drive ............................................................... 19 Absolute Maximum Ratings............................................................ 5 Power-Up Behavior .................................................................... 19 Thermal Characteristics .............................................................. 5 Applications Information .............................................................. 20 ESD Caution .................................................................................. 5 A 5 V Precision Strain Gage Circuit ........................................ 20 Typical Performance Characteristics ............................................. 6 3 V Instrumentation Amplifier ................................................ 20 Functional Description .................................................................. 14 A High Accuracy Thermocouple Amplifier ........................... 21 Amplifier Architecture .............................................................. 14 Precision Current Meter ............................................................ 21 Basic Auto-Zero Amplifier Theory .......................................... 14 Precision Voltage Comparator.................................................. 21 High Gain, CMRR, PSRR .......................................................... 15 Outline Dimensions ....................................................................... 22 Maximizing Performance Through Proper Layout ............... 16 Ordering Guide .......................................................................... 24 REVISION HISTORY 6/15—Rev. E to Rev. F Change to Input Voltage Parameter, Table 3 ................................. 5 3/07—Rev. B to Rev. C Changes to Specifications Section ...................................................3 11/12—Rev.D to Rev. E Changes to Figure 68 ...................................................................... 21 Updated Outline Dimensions (RM-8) ......................................... 22 Changes to Ordering Guide .......................................................... 24 2/07—Rev. A to Rev. B Updated Format .................................................................. Universal Changes to Figure 54...................................................................... 16 Deleted Spice Model Section ........................................................ 19 Deleted Figure 63, Renumbered Sequentially ............................ 19 Changes to Ordering Guide .......................................................... 24 9/08—Rev. C to Rev. D Changes to Ordering Guide .......................................................... 23 11/02—Rev. 0 to Rev. A Edits to Figure 60 ............................................................................ 16 Updated Outline Dimensions ....................................................... 20 Rev. F | Page 2 of 24 Data Sheet AD8551/AD8552/AD8554 SPECIFICATIONS ELECTRICAL CHARACTERISTICS VS = 5 V, VCM = 2.5 V, VO = 2.5 V, TA = 25°C, unless otherwise noted. Table 1. Parameter INPUT CHARACTERISTICS Offset Voltage Symbol Conditions Min VOS Typ Max Unit 1 5 10 50 1.5 300 4 70 200 150 400 5 μV μV pA nA pA nA pA pA pA pA V dB dB dB dB μV/°C −40°C ≤ TA ≤ +125°C Input Bias Current AD8551/AD8554 AD8552 AD8552 Input Offset Current AD8551/AD8554 AD8552 AD8552 Input Voltage Range Common-Mode Rejection Ratio CMRR Large Signal Voltage Gain1 AVO Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage High IB 10 1.0 160 2.5 20 150 30 150 −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +85°C −40°C ≤ TA ≤ +125°C IOS −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +85°C −40°C ≤ TA ≤ +125°C ΔVOS/ΔT VOH Output Voltage Low VOL Output Short-Circuit Limit Current ISC VCM = 0 V to +5 V −40°C ≤ TA ≤ +125°C RL = 10 kΩ, VO = 0.3 V to 4.7 V −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +125°C RL = 100 kΩ to GND RL = 100 kΩ to GND @ −40°C to +125°C RL = 10 kΩ to GND RL = 10 kΩ to GND @ −40°C to +125°C RL = 100 kΩ to V+ RL = 100 kΩ to V+ @ −40°C to +125°C RL = 10 kΩ to V+ RL = 10 kΩ to V+ @ −40°C to +125°C 0 120 115 125 120 4.99 4.99 4.95 4.95 ±25 −40°C to +125°C Output Current IO −40°C to +125°C POWER SUPPLY Power Supply Rejection Ratio Supply Current/Amplifier DYNAMIC PERFORMANCE Slew Rate Overload Recovery Time Gain Bandwidth Product NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density 1 PSRR ISY SR VS = 2.7 V to 5.5 V −40°C ≤ TA ≤ +125°C VO = 0 V −40°C ≤ TA ≤ +125°C RL = 10 kΩ GBP en p-p en p-p en in 0 Hz to 10 Hz 0 Hz to 1 Hz f = 1 kHz f = 10 Hz Gain testing is dependent upon test bandwidth. Rev. F | Page 3 of 24 120 115 140 130 145 135 0.005 4.998 4.997 4.98 4.975 1 2 10 15 ±50 ±40 ±30 ±15 130 130 850 1000 0.4 0.05 1.5 1.0 0.32 42 2 0.04 10 10 30 30 975 1075 0.3 V V V V mV mV mV mV mA mA mA mA dB dB μA μA V/μs ms MHz μV p-p μV p-p nV/√Hz fA/√Hz AD8551/AD8552/AD8554 Data Sheet VS = 2.7 V, VCM = 1.35 V, VO = 1.35 V, TA = 25°C, unless otherwise noted. Table 2. Parameter INPUT CHARACTERISTICS Offset Voltage Symbol Conditions Min VOS Typ Max Unit 1 5 10 50 1.5 300 4 50 200 150 400 2.7 μV μV pA nA pA nA pA pA pA pA V dB dB dB dB μV/°C −40°C ≤ TA ≤ +125°C Input Bias Current AD8551/AD8554 AD8552 AD8552 Input Offset Current AD8551/AD8554 AD8552 AD8552 Input Voltage Range Common-Mode Rejection Ratio IB Large Signal Voltage Gain1 AVO Offset Voltage Drift OUTPUT CHARACTERISTICS Output Voltage High 10 1.0 160 2.5 10 150 30 150 −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +85°C −40°C ≤ TA ≤ +125°C IOS −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +85°C −40°C ≤ TA ≤ +125°C CMRR ΔVOS/ΔT VOH Output Voltage Low VOL Short-Circuit Limit ISC VCM = 0 V to 2.7 V −40°C ≤ TA ≤ +125°C RL = 10 kΩ, VO = 0.3 V to 2.4 V −40°C ≤ TA ≤ +125°C −40°C ≤ TA ≤ +125°C RL = 100 kΩ to GND RL = 100 kΩ to GND @ −40°C to +125°C RL = 10 kΩ to GND RL = 10 kΩ to GND @ −40°C to +125°C RL = 100 kΩ to V+ RL = 100 kΩ to V+ @ −40°C to +125°C RL = 10 kΩ to V+ RL = 10 kΩ to V+ @ −40°C to +125°C 0 115 110 110 105 2.685 2.685 2.67 2.67 ±10 −40°C to +125°C Output Current IO −40°C to +125°C POWER SUPPLY Power Supply Rejection Ratio Supply Current/Amplifier DYNAMIC PERFORMANCE Slew Rate Overload Recovery Time Gain Bandwidth Product NOISE PERFORMANCE Voltage Noise Voltage Noise Density Current Noise Density 1 PSRR ISY SR VS = 2.7 V to 5.5 V −40°C ≤ TA ≤ +125°C VO = 0 V −40°C ≤ TA ≤ +125°C 2.697 2.696 2.68 2.675 1 2 10 15 ±15 ±10 ±10 ±5 130 130 750 950 0.04 10 10 20 20 900 1000 V V V V mV mV mV mV mA mA mA mA dB dB μA μA RL = 10 kΩ 0.5 0.05 1 V/μs ms MHz 0 Hz to 10 Hz f = 1 kHz f = 10 Hz 1.6 75 2 μV p-p nV/√Hz fA/√Hz GBP en p-p en in 120 115 130 130 140 130 0.005 Gain testing is dependent upon test bandwidth. Rev. F | Page 4 of 24 Data Sheet AD8551/AD8552/AD8554 ABSOLUTE MAXIMUM RATINGS THERMAL CHARACTERISTICS Table 3. Parameter Supply Voltage Input Voltage Differential Input Voltage1 ESD (Human Body Model) Output Short-Circuit Duration to GND Storage Temperature Range Operating Temperature Range Junction Temperature Range Lead Temperature Range (Soldering, 60 sec) 1 Rating 6V GND −0.3 V to VS + 0.3 V ±5.0 V 2000 V Indefinite −65°C to +150°C −40°C to +125°C −65°C to +150°C 300°C Table 4. Package Type 8-Lead MSOP (RM) 8-Lead TSSOP (RU) 8-Lead SOIC (R) 14-Lead TSSOP (RU) 14-Lead SOIC (R) ESD CAUTION Differential input voltage is limited to ±5.0 V or the supply voltage, whichever is less. Stresses at or above those listed under Absolute Maximum Ratings may cause permanent damage to the product. This is a stress rating only; functional operation of the product at these or any other conditions above those indicated in the operational section of this specification is not implied. Operation beyond the maximum operating conditions for extended periods may affect product reliability. Rev. F | Page 5 of 24 θJA 190 240 158 180 120 θJC 44 43 43 36 36 Unit °C/W °C/W °C/W °C/W °C/W AD8551/AD8552/AD8554 Data Sheet TYPICAL PERFORMANCE CHARACTERISTICS 180 180 NUMBER OF AMPLIFIERS 140 120 100 80 60 140 120 100 80 60 40 40 20 20 0 –2.5 –1.5 –0.5 1.5 0.5 OFFSET VOLTAGE (µV) VSY = 5V VCM = 2.5V TA = 25°C 160 0 –2.5 01101-007 NUMBER OF AMPLIFIERS 160 2.5 –1.5 Figure 7. Input Offset Voltage Distribution at 2.7 V 2.5 Figure 10. Input Offset Voltage Distribution at 5 V 12 50 VSY = 5V TA = –40°C, +25°C, +85°C 40 VSY = 5V VCM = 2.5V TA = –40°C TO +125°C 10 +85°C 30 NUMBER OF AMPLIFIERS INPUT BIAS CURRENT (pA) 0.5 –0.5 1.5 OFFSET VOLTAGE (µV) 01101-010 VSY = 2.7V VCM = 1.35V TA = 25°C 20 10 +25°C 0 –40°C –10 8 6 4 2 0 1 4 2 3 INPUT COMMON-MODE VOLTAGE (V) 0 01101-008 –30 5 Figure 8. Input Bias Current vs. Common-Mode Voltage 1500 1 0 2 3 4 INPUT OFFSET DRIFT (nV/°C) 5 6 01101-011 –20 Figure 11. Input Offset Voltage Drift Distribution at 5 V 10k VSY = 5V TA = 125°C VSY = 5V TA = 25°C 1000 OUTPUT VOLTAGE (mV) 500 0 –500 –1000 100 SOURCE 10 SINK 1 –2000 0 1 4 2 3 INPUT COMMON-MODE VOLTAGE (V) 5 0.1 0.0001 Figure 9. Input Bias Current vs. Common-Mode Voltage 0.001 0.01 0.1 1 LOAD CURRENT (mA) 10 100 Figure 12. Output Voltage to Supply Rail vs. Load Current at 5 V Rev. F | Page 6 of 24 01101-012 –1500 01101-009 INPUT BIAS CURRENT (pA) 1k Data Sheet AD8551/AD8552/AD8554 800 10k TA = +25°C 100 SOURCE 10 SINK 1 1 0.01 0.1 LOAD CURRENT (mA) 0.001 100 10 500 400 300 200 100 0 01101-013 0.1 0.0001 600 3 4 2 SUPPLY VOLTAGE (V) 5 6 Figure 16. Supply Current per Amplifier vs. Supply Voltage Figure 13. Output Voltage to Supply Rail vs. Load Current at 2.7 V 60 0 VCM = 2.5V VSY = 5V 50 40 OPEN-LOOP GAIN (dB) –250 –500 –750 VSY = 2.7V CL = 0pF RL = ∞ 0 30 45 20 90 10 135 0 180 –10 225 –20 270 –25 75 0 25 50 TEMPERATURE (°C) 100 125 150 –40 10k 1M FREQUENCY (Hz) 10M 100M Figure 17. Open-Loop Gain and Phase Shift vs. Frequency at 2.7 V Figure 14. Input Bias Current vs. Temperature 60 1.0 VCM = 2.5V VSY = 5V 100k 50 5V 40 OPEN-LOOP GAIN (dB) 0.8 2.7V 0.6 0.4 0.2 VSY = 5V CL = 0pF RL = ∞ 0 30 45 20 90 10 135 0 180 –10 225 –20 270 PHASE SHIFT (Degrees) –50 01101-014 –1000 –75 01101-017 –30 0 –75 –50 –25 0 75 25 50 TEMPERATURE (°C) 100 125 150 –40 10k 100k 1M FREQUENCY (Hz) 10M 100M Figure 18. Open-Loop Gain and Phase Shift vs. Frequency at 5 V Figure 15. Supply Current vs. Temperature Rev. F | Page 7 of 24 01101-018 –30 01101-015 SUPPLY CURRENT (mA) INPUT BIAS CURRENT (pA) 1 0 PHASE SHIFT (Degrees) OUTPUT VOLTAGE (mV) 1k 700 01101-016 SUPPLY CURRENT PER AMPLIFIER (µA) VSY = 2.7V TA = 25°C AD8551/AD8552/AD8554 Data Sheet 60 300 50 AV = –100 270 240 OUTPUT IMPEDANCE (Ω) 40 CLOSED-LOOP GAIN (dB) VSY = 5V VSY = 2.7V CL = 0pF RL = 2kΩ 30 20 AV = –10 10 AV = +1 0 –10 210 180 150 120 AV = 100 90 –20 60 –30 30 AV = 10 1k 10k 100k FREQUENCY (Hz) 1M 10M 0 100 01101-019 –40 100 Figure 19. Closed-Loop Gain vs. Frequency at 2.7 V 1k 1M 10k 100k FREQUENCY (Hz) 10M Figure 22. Output Impedance vs. Frequency at 5 V 60 AV = –100 40 CLOSED-LOOP GAIN (dB) VSY = 2.7V CL = 300pF RL = 2kΩ AV = 1 VSY = 5V CL = 0pF RL = 2kΩ 50 30 20 AV = –10 10 AV = +1 0 –10 –30 1k 10k 100k FREQUENCY (Hz) 1M 10M 500mV 01101-020 2µs –40 100 Figure 20. Closed-Loop Gain vs. Frequency at 5 V 01101-023 –20 Figure 23. Large Signal Transient Response at 2.7 V 300 270 VSY = 5V CL = 300pF RL = 2kΩ AV = 1 VSY = 2.7V 210 180 150 120 90 AV = 100 60 0 100 AV = 1 1k 10k 100k FREQUENCY (Hz) 1M 5µs 10M 1V Figure 21. Output Impedance vs. Frequency at 2.7 V Figure 24. Large Signal Transient Response at 5 V Rev. F | Page 8 of 24 01101-024 AV = 10 30 01101-021 OUTPUT IMPEDANCE (Ω) 240 01101-022 AV = 1 Data Sheet AD8551/AD8552/AD8554 45 VSY = ±1.35V CL = 50pF VSY = ±2.5V RL = 2kΩ TA = 25°C 40 35 30 25 20 +OS 15 –OS 10 5 0 10 Figure 25. Small Signal Transient Response at 2.7 V 100 1k CAPACITANCE (pF) 10k Figure 28. Small Signal Overshoot vs. Load Capacitance at 5 V VSY = ±2.5V CL = 50pF 0V RL = ∞ AV = 1 VIN VSY = ±2.5V VIN = –200mV p-p (RET TO GND) CL = 0pF RL = 10kΩ AV = –100 VOUT 0V 20µs 1V 01101-029 01101-026 50mV 5µs BOTTOM SCALE: 1V/DIV TOP SCALE: 200mV/DIV Figure 29. Positive Overvoltage Recovery Figure 26. Small Signal Transient Response at 5 V 50 VSY = ±1.35V RL = 2kΩ TA = 25°C VIN 40 0V 35 30 25 +OS 0V 20 15 –OS 10 VOUT 5 20µs 100 1k CAPACITANCE (pF) 10k 01101-027 0 10 VSY = ±2.5V VIN = 200mV p-p (RET TO GND) CL = 0pF RL = 10kΩ AV = –100 1V BOTTOM SCALE: 1V/DIV TOP SCALE: 200mV/DIV Figure 30. Negative Overvoltage Recovery Figure 27. Small Signal Overshoot vs. Load Capacitance at 2.7 V Rev. F | Page 9 of 24 01101-030 SMALL SIGNAL OVERSHOOT (%) 45 01101-028 50mV 5µs 01101-025 SMALL SIGNAL OVERSHOOT (%) RL = ∞ AV = 1 AD8551/AD8552/AD8554 Data Sheet 140 VS = ±2.5V RL = 2kΩ AV = –100 VIN = 60mV p-p VSY = ±1.35V 120 PSRR (dB) 100 80 60 +PSRR 40 0 100 1M 10M Figure 34. PSRR vs. Frequency at ±1.35 V Figure 31. No Phase Reversal 140 140 VSY = 2.7V VSY = ±2.5V 120 120 100 100 PSRR (dB) CMRR (dB) 100k 10k FREQUENCY (Hz) 1k 01101-034 20 01101-031 1V 200µs –PSRR 80 60 80 +PSRR 60 40 40 20 20 1k 10k 100k FREQUENCY (Hz) 1M 10M 0 100 01101-032 0 100 Figure 32. CMRR vs. Frequency at 2.7 V 10k 100k FREQUENCY (Hz) 1k 10M Figure 35. PSRR vs. Frequency at ±2.5 V 140 3.0 VSY = ±1.35V RL = 2kΩ AV = 1 THD+N < 1% TA = 25°C VSY = 5V 120 OUTPUT SWING (V p-p) 2.5 100 80 60 40 1.5 1.0 1k 10k 100k FREQUENCY (Hz) 1M 10M Figure 33. CMRR vs. Frequency at 5 V 0 100 1k 10k FREQUENCY (Hz) 100k 1M Figure 36. Maximum Output Swing vs. Frequency at 2.7 V Rev. F | Page 10 of 24 01101-036 0 100 2.0 0.5 20 01101-033 CMRR (dB) 1M 01101-035 –PSRR Data Sheet AD8551/AD8552/AD8554 5.5 VSY = ±2.5V RL = 2kΩ AV = 1 THD+N < 1% TA = 25°C 5.0 156 en (nV/√Hz) 3.5 3.0 2.5 2.0 130 104 78 1.5 52 1.0 1k 10k FREQUENCY (Hz) 100k 1M 01101-037 0 100 0 0.5 1.0 1.5 FREQUENCY (kHz) 2.0 2.5 01101-040 26 0.5 Figure 40. Voltage Noise Density at 2.7 V from 0 Hz to 2.5 kHz Figure 37. Maximum Output Swing vs. Frequency at 5 V VSY = ±1.35V AV = 10000 VSY = 2.7V RS = 0Ω 112 en (nV/√Hz) 96 0V 80 64 48 32 2mV 0 Figure 38. 0.1 Hz to 10 Hz Noise at 2.7 V 5 10 15 FREQUENCY (kHz) 20 25 01101-041 16 01101-038 1s Figure 41. Voltage Noise Density at 2.7 V from 0 Hz to 25 kHz VSY = ±2.5V AV = 10000 VSY = 5V RS = 0Ω 91 en (nV/√Hz) 78 65 52 39 26 2mV 13 0 Figure 39. 0.1 Hz to 10 Hz Noise at 5 V 0.5 1.0 1.5 FREQUENCY (kHz) 2.0 2.5 Figure 42. Voltage Noise Density at 5 V from 0 Hz to 2.5 kHz Rev. F | Page 11 of 24 01101-042 1s 01101-039 OUTPUT SWING (V p-p) 4.5 4.0 VSY = 2.7V RS = 0Ω 182 AD8551/AD8552/AD8554 Data Sheet 150 VSY = 5V RS = 0Ω VSY = 2.7V TO 5.5V POWER SUPPLY REJECTION (dB) 112 en (nV/√Hz) 96 80 64 48 32 145 140 135 130 5 10 15 FREQUENCY (kHz) 20 25 125 –75 01101-043 0 Figure 43. Voltage Noise Density at 5 V from 0 Hz to 25 kHz –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 150 01101-045 16 Figure 45. Power Supply Rejection vs. Temperature 50 VSY = 5V RS = 0Ω VSY = 2.7V 40 SHORT-CIRCUIT CURRENT (mA) 168 120 96 72 48 24 30 ISC– 20 10 0 ISC+ –10 –20 –30 0 5 FREQUENCY (Hz) 10 –50 –75 Figure 44. Voltage Noise Density at 5 V from 0 Hz to 10 Hz –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 150 Figure 46. Output Short-Circuit Current vs. Temperature Rev. F | Page 12 of 24 01101-046 –40 01101-044 en (nV/√Hz) 144 Data Sheet AD8551/AD8552/AD8554 100 250 SHORT-CIRCUIT CURRENT (mA) ISC– 60 40 20 0 –20 ISC+ –40 –60 –100 –75 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 150 01101-047 –80 VSY = 2.7V 225 200 175 150 RL = 1kΩ 100 75 RL = 100kΩ 25 0 –75 RL = 10kΩ –50 –25 0 25 50 75 TEMPERATURE (°C) 100 125 150 01101-048 OUTPUT VOLTAGE TO SUPPLY RAIL (mV) 250 50 200 175 150 RL = 1kΩ 125 100 75 50 RL = 10kΩ 25 0 –75 –50 –25 0 25 50 75 TEMPERATURE (°C) RL = 100kΩ 100 125 150 Figure 49. Output Voltage to Supply Rail vs. Temperature Figure 47. Output Short-Circuit Current vs. Temperature 125 VSY = 5.0V 225 Figure 48. Output Voltage to Supply Rail vs. Temperature Rev. F | Page 13 of 24 01101-049 OUTPUT VOLTAGE TO SUPPLY RAIL (mV) VSY = 5.0V 80 AD8551/AD8552/AD8554 Data Sheet FUNCTIONAL DESCRIPTION The AD8551/AD8552/AD8554 are CMOS amplifiers and achieve their high degree of precision through auto-zero stabilization. This autocorrection topology allows the AD8551/AD8552/AD8554 to maintain its low offset voltage over a wide temperature range and over its operating lifetime. AMPLIFIER ARCHITECTURE Each AD8551/AD8552/AD8554 op amps consist of two amplifiers, a main amplifier and a secondary amplifier, used to correct the offset voltage of the main amplifier. Both consist of a rail-to-rail input stage, allowing the input common-mode voltage range to reach both supply rails. The input stage consists of an NMOS differential pair operating concurrently with a parallel PMOS differential pair. The outputs from the differential input stages are combined in another gain stage whose output is used to drive a rail-to-rail output stage. The wide voltage swing of the amplifier is achieved by using two output transistors in a common-source configuration. The output voltage range is limited by the drain-to-source resistance of these transistors. As the amplifier is required to source or sink more output current, the rDS of these transistors increases, raising the voltage drop across these transistors. Simply put, the output voltage does not swing as close to the rail under heavy output current conditions as it does with light output current. This is a characteristic of all rail-to-rail output amplifiers. Figure 12 and Figure 13 show how close the output voltage can get to the rails with a given output current. The output of the AD8551/AD8552/ AD8554 is short-circuit protected to approximately 50 mA of current. some improvements made over time. The AD8551/AD8552/ AD8554 design offers a number of significant performance improvements over previous versions while attaining a very substantial reduction in device cost. This section offers a simplified explanation of how the AD8551/AD8552/AD8554 are able to offer extremely low offset voltages and high open-loop gains. As noted in the Amplifier Architecture section, each AD8551/ AD8552/AD8554 op amp contains two internal amplifiers. One is used as the primary amplifier, the other as an autocorrection, or nulling, amplifier. Each amplifier has an associated input offset voltage that can be modeled as a dc voltage source in series with the noninverting input. In Figure 50 and Figure 51 these are labeled as VOSX, where x denotes the amplifier associated with the offset: A for the nulling amplifier and B for the primary amplifier. The open-loop gain for the +IN and −IN inputs of each amplifier is given as AX. Both amplifiers also have a third voltage input with an associated open-loop gain of BX. There are two modes of operation determined by the action of two sets of switches in the amplifier: an auto-zero phase and an amplification phase. Auto-Zero Phase In this phase, all φA switches are closed and all φB switches are opened. Here, the nulling amplifier is taken out of the gain loop by shorting its two inputs together. Of course, there is a degree of offset voltage, shown as VOSA, inherent in the nulling amplifier which maintains a potential difference between the +IN and −IN inputs. The nulling amplifier feedback loop is closed through φB2 and VOSA appears at the output of the nulling amp and on CM1, an internal capacitor in the AD8551/AD8552/AD8554. Mathematically, this is expressed in the time domain as VOA[t] = AAVOSA[t] − BAVOA[t] (1) which can be expressed as VOA t AAVOSA t 1 BA (2) This demonstrates that the offset voltage of the nulling amplifier times a gain factor appears at the output of the nulling amplifier and, thus, on the CM1 capacitor. The AD8551/AD8552/AD8554 amplifiers have exceptional gain, yielding greater than 120 dB of open-loop gain with a load of 2 kΩ. Because the output transistors are configured in a common-source configuration, the gain of the output stage, and thus the openloop gain of the amplifier, is dependent on the load resistance. Open-loop gain decreases with smaller load resistances. This is another characteristic of rail-to-rail output amplifiers. BASIC AUTO-ZERO AMPLIFIER THEORY VIN+ AB VIN– ФB ФA VOUT BB VOA VOSA + ФB AA CM2 VNB –BA ФA CM1 VNA Autocorrection amplifiers are not a new technology. Various IC implementations have been available for more than 15 years with Rev. F | Page 14 of 24 Figure 50. Auto-Zero Phase of the AD8551/AD8552/AD8554 01101-050 The AD8551/AD8552/AD8554 are high precision, rail-to-rail operational amplifiers that can be run from a single-supply voltage. Their typical offset voltage of less than 1 μV allows these amplifiers to be easily configured for high gains without risk of excessive output voltage errors. The extremely small temperature drift of 5 nV/°C ensures a minimum of offset voltage error over its entire temperature range of −40°C to +125°C, making the AD8551/AD8552/AD8554 amplifiers ideal for a variety of sensitive measurement applications in harsh operating environments, such as underhood and braking/suspension systems in automobiles. Data Sheet AD8551/AD8552/AD8554 Amplification Phase When the φB switches close and the φA switches open for the amplification phase, this offset voltage remains on CM1 and, essentially, corrects any error from the nulling amplifier. The voltage across CM1 is designated as VNA. Furthermore, VIN is designated as the potential difference between the two inputs to the primary amplifier, or VIN = (VIN+ − VIN−). Thus, the nulling amplifier can be expressed as VOA [t ] = A A (VIN [t ] − VOSA [t ]) − B AVNA [t ] (3) VIN+ VOUT AB VIN– BB VOA VOSA + ФA ФB AA VOUT [t ] = AB (VIN [t ] + VOSB ) + BBVNB V VOUT [t ] = A BVIN [t ] + A BVOSB + B B A A VIN [t ] + OSB 1 + BA ФA VOUT [t ] = VIN [t ](AB + AB BB ) + Because φA is now open and there is no place for CM1 to discharge, the voltage (VNA), at the present time (t), is equal to the voltage at the output of the nulling amp (VOA) at the time when φA was closed. If the period of the autocorrection switching frequency is labeled tS, then the amplifier switches between phases every 0.5 × tS. Therefore, in the amplification phase (4) (5) VOUT [t ] ≈ VIN [t ]AA BA + AA (VOSA + VOSB ) time invariant; therefore, Equation 5 can be rearranged and rewritten as AA (1 + BA )VOSA − AA BAVOSA 1 + BA (6) or V VOA [t ] = AA VIN [t ] + OSA 1 + BA (11) Most obvious is the gain product of both the primary and nulling amplifiers. This AABA term is what gives the AD8551/AD8552/ AD8554 its extremely high open-loop gain. To understand how VOSA and VOSB relate to the overall effective input offset voltage of the complete amplifier, establish the generic amplifier equation of VOUT = k × (VIN + VOS , EFF ) (12) where k is the open-loop gain of an amplifier and VOS, EFF is its effective offset voltage. Putting Equation 12 into the form of Equation 11 gives For the sake of simplification, assume that the autocorrection frequency is much faster than any potential change in VOSA or VOSB. This is a valid assumption because changes in offset voltage are a function of temperature variation or long-term wear time, both of which are much slower than the auto-zero clock frequency of the AD8551/AD8552/AD8554. This effectively renders VOS VOA [t ] = AAVIN [t ] + (10) Also, the gain product of AABB is much greater than AB. These allow Equation 10 to be simplified to Substituting Equation 4 and Equation 2 into Equation 3 yields 1 AA B AVOSA t − t S 2 VOA [t ] = AAVIN [t ] + AAVOSA [t ] − 1 + BA AA BAVOSA + ABVOSA 1 + BA AA = AB and BA = BB and BA >> 1 Figure 51. Output Phase of the Amplifier 1 VNA [t ] = VNA t − t S 2 (9) The AD8551/AD8552/AD8554 architecture is optimized in such a way that CM1 VNA Combining terms, VNB –BA (8) In the amplification phase, VOA = VNB, so this can be rewritten as CM2 01101-051 ФB the nulling amplifier has greatly reduced its own offset voltage error even before correcting the primary amplifier. This results in the primary amplifier output voltage becoming the voltage at the output of the AD8551/AD8552/AD8554 amplifiers. It is equal to (7) From these equations, the auto-zeroing action becomes evident. Note the VOS term is reduced by a 1 + BA factor. This shows how VOUT [t ] ≈ VIN [t ]AA BA + VOS , EFF AA BA (13) Thus, it is evident that VOS , EFF ≈ VOSA + VOSB BA (14) The offset voltages of both the primary and nulling amplifiers are reduced by the Gain Factor BA. This takes a typical input offset voltage from several millivolts down to an effective input offset voltage of submicrovolts. This autocorrection scheme is the outstanding feature of the AD8551/AD8552/AD8554 series that continues to earn the reputation of being among the most precise amplifiers available on the market. HIGH GAIN, CMRR, PSRR Common-mode and power supply rejection are indications of the amount of offset voltage an amplifier has as a result of a change in its input common-mode or power supply voltages. As shown in the previous section, the autocorrection architecture of the Rev. F | Page 15 of 24 AD8551/AD8552/AD8554 Data Sheet PC board at one end of the component (TA1) is different from the temperature at the other end (TA2), the resulting Seebeck voltages are not equal, resulting in a thermal voltage error. MAXIMIZING PERFORMANCE THROUGH PROPER LAYOUT To achieve the maximum performance of the extremely high input impedance and low offset voltage of the AD8551/ AD8552/AD8554, care is needed in laying out the circuit board. The PC board surface must remain clean and free of moisture to avoid leakage currents between adjacent traces. Surface coating of the circuit board reduces surface moisture and provides a humidity barrier, reducing parasitic resistance on the board. The use of guard rings around the amplifier inputs further reduces leakage currents. Figure 52 shows proper guard ring configuration, and Figure 53 shows the top view of a surfacemount layout. The guard ring does not need to be a specific width, but it should form a continuous loop around both inputs. By setting the guard ring voltage equal to the voltage at the noninverting input, parasitic capacitance is minimized as well. For further reduction of leakage currents, components can be mounted to the PC board using Teflon standoff insulators. This thermocouple error can be reduced by using dummy components to match the thermoelectric error source. Placing the dummy component as close as possible to its partner ensures both Seebeck voltages are equal, thus canceling the thermocouple error. Maintaining a constant ambient temperature on the circuit board further reduces this error. The use of a ground plane helps distribute heat throughout the board and reduces EMI noise pickup. COMPONENT LEAD VSC1 + SURFACE-MOUNT COMPONENT + VSC2 VTS1 + SOLDER + VTS2 PC BOARD TA1 TA2 IF TA1 ≠ TA2, THEN VTS1 + VSC1 ≠ VTS2 + VSC2 COPPER TRACE 01101-054 AD8551/AD8552/AD8554 allows it to quite effectively minimize offset voltages. The technique also corrects for offset errors caused by common-mode voltage swings and power supply variations. This results in superb CMRR and PSRR figures in excess of 130 dB. Because the autocorrection occurs continuously, these figures can be maintained across the entire temperature range of the device, from −40°C to +125°C. Figure 54. Mismatch in Seebeck Voltages Causes Thermoelectric Voltage Error RF R1 VOUT VIN AD8551/ AD8552/ AD8554 RS = R1 VOUT VIN VIN AD8552 NOTES 1. RS SHOULD BE PLACED IN CLOSE PROXIMITY AND ALIGNMENT TO R1 TO BALANCE SEEBECK VOLTAGES. AD8552 01101-055 AV = 1 + (RF/R1) VOUT Figure 55. Using Dummy Components to Cancel Thermoelectric Voltage Errors VIN VOUT 01101-052 AD8552 Figure 52. Guard Ring Layout and Connections to Reduce PC Board Leakage Currents V+ R1 R2 AD8552 VIN1 R2 R1 GUARD RING VREF VREF GUARD RING V– 01101-053 VIN2 Figure 53. Top View of AD8552 SOIC Layout with Guard Rings Other potential sources of offset error are thermoelectric voltages on the circuit board. This voltage, also called Seebeck voltage, occurs at the junction of two dissimilar metals and is proportional to the temperature of the junction. The most common metallic junctions on a circuit board are solder-to-board trace and solder-to-component lead. Figure 54 shows a cross-section of the thermal voltage error sources. If the temperature of the 1/f NOISE CHARACTERISTICS Another advantage of auto-zero amplifiers is their ability to cancel flicker noise. Flicker noise, also known as 1/f noise, is noise inherent in the physics of semiconductor devices, and it increases 3 dB for every octave decrease in frequency. The 1/f corner frequency of an amplifier is the frequency at which the flicker noise is equal to the broadband noise of the amplifier. At lower frequencies, flicker noise dominates, causing higher degrees of error for sub-Hertz frequencies or dc precision applications. Because the AD8551/AD8552/AD8554 amplifiers are selfcorrecting op amps, they do not have increasing flicker noise at lower frequencies. In essence, low frequency noise is treated as a slowly varying offset error and is greatly reduced as a result of autocorrection. The correction becomes more effective as the noise frequency approaches dc, offsetting the tendency of the noise to increase exponentially as frequency decreases. This allows the AD8551/AD8552/AD8554 to have lower noise near dc than standard low noise amplifiers that are susceptible to 1/f noise. Rev. F | Page 16 of 24 Data Sheet AD8551/AD8552/AD8554 INTERMODULATION DISTORTION The AD8551/AD8552/AD8554 can be used as a conventional op amp for gain/ bandwidth combinations up to 1.5 MHz. The auto-zero correction frequency of the device is fixed at 4 kHz. Although a trace amount of this frequency feeds through to the output, the amplifier can be used at much higher frequencies. Figure 56 shows the spectral output of the AD8552 with the amplifier configured for unity gain and the input grounded. the amplifier. Figure 58 shows the spectral output of an AD8552 configured as a high gain stage (+60 dB) with a 1 mV input signal applied. The relative levels of all IMD products and harmonic distortion add up to produce an output error of −60 dB relative to the input signal. At unity gain, these add up to only −120 dB relative to the input signal. 0 VSY = 5V AV = 60dB OUTPUT SIGNAL 1V rms @ 200Hz –20 OUTPUT SIGNAL (dB) The 4 kHz auto-zero clock frequency appears at the output with less than 2 μV of amplitude. Harmonics are also present, but at reduced levels from the fundamental auto-zero clock frequency. The amplitude of the clock frequency feedthrough is proportional to the closed-loop gain of the amplifier. Like other autocorrection amplifiers, at higher gains there is more clock frequency feedthrough. Figure 57 shows the spectral output with the amplifier configured for a gain of 60 dB. –60 –80 IMD < 100µV rms –120 VSY = 5V AV = 0dB 0 1 2 3 4 5 6 FREQUENCY (kHz) 7 8 9 10 01101-058 –100 0 –20 –40 OUTPUT SIGNAL (dB) Figure 58. Spectral Analysis of AD8552 in High Gain with a 1 mV Input Signal –40 For most low frequency applications, the small amount of autozero clock frequency feedthrough does not affect the precision of the measurement system. If it is desired, the clock frequency feedthrough can be reduced through the use of a feedback capacitor around the amplifier. However, this reduces the bandwidth of the amplifier. Figure 59 and Figure 60 show a configuration for reducing the clock feedthrough and the corresponding spectral analysis at the output. The −3 dB bandwidth of this configuration is 480 Hz. –60 –80 –100 –140 0 1 2 3 4 5 6 FREQUENCY (kHz) 7 8 9 10 01101-056 –120 Figure 56. Spectral Analysis of AD8552 Output in Unity Gain Configuration 3.3nF 0 –20 100Ω VIN = 1mV rms @ 200Hz –40 –60 01101-059 OUTPUT SIGNAL (dB) 100kΩ VSY = 5V AV = 60dB Figure 59. Reducing Autocorrection Clock Noise Using a Feedback Capacitor –80 0 VSY = 5V AV = 60dB –100 –20 1 2 3 4 5 6 FREQUENCY (kHz) 7 8 9 10 Figure 57. Spectral Analysis of AD8551/AD8552/AD8554 Output with +60 dB Gain When an input signal is applied, the output contains some degree of intermodulation distortion (IMD). This is another characteristic feature of all autocorrection amplifiers. IMD appears as sum and difference frequencies between the input signal and the 4 kHz clock frequency (and its harmonics) and is at a level similar to, or less than, the clock feedthrough at the output. The IMD is also proportional to the closed-loop gain of Rev. F | Page 17 of 24 –40 –60 –80 –100 –120 0 1 2 3 4 5 6 FREQUENCY (kHz) 7 8 9 10 Figure 60. Spectral Analysis Using a Feedback Capacitor 01101-060 0 01101-057 –140 OUTPUT SIGNAL –120 AD8551/AD8552/AD8554 Data Sheet BROADBAND AND EXTERNAL RESISTOR NOISE CONSIDERATIONS The total broadband noise output from any amplifier is primarily a function of three types of noise: input voltage noise from the amplifier, input current noise from the amplifier, and Johnson noise from the external resistors used around the amplifier. Input voltage noise, or en, is strictly a function of the amplifier used. The Johnson noise from a resistor is a function of the resistance and the temperature. Input current noise, or in, creates an equivalent voltage noise proportional to the resistors used around the amplifier. These noise sources are not correlated with each other and their combined noise sums in a rootsquared-sum fashion. The full equation is given as [ en _ TOTAL = en2 + 4kTrS + (in RS )2 ] 1 2 (15) Where: en = the input voltage noise density of the amplifier. in = the input current noise of the amplifier. RS = source resistance connected to the noninverting terminal. k = Boltzmann’s constant (1.38 × 10−23 J/K). T = ambient temperature in Kelvin (K = 273.15 + °C). The input voltage noise density (en) of the AD8551/AD8552/ AD8554 is 42 nV/√Hz, and the input noise, in, is 2 fA/√Hz. The en, TOTAL is dominated by the input voltage noise, provided the source resistance is less than 106 kΩ. With source resistance greater than 106 kΩ, the overall noise of the system is dominated by the Johnson noise of the resistor itself. Because the input current noise of the AD8551/AD8552/ AD8554 is very small, it does not become a dominant term unless RS is greater than 4 GΩ, which is an impractical value of source resistance. The total noise (en, TOTAL) is expressed in volts per square root Hertz, and the equivalent rms noise over a certain bandwidth can be found as en = en,TOTAL × BW (16) where BW is the bandwidth of interest in Hertz. OUTPUT OVERDRIVE RECOVERY The AD8551/AD8552/AD8554 amplifiers have an excellent overdrive recovery of only 200 μs from either supply rail. This characteristic is particularly difficult for autocorrection amplifiers because the nulling amplifier requires a nontrivial amount of time to error correct the main amplifier back to a valid output. Figure 29 and Figure 30 show the positive and negative overdrive recovery times for the AD8551/AD8552/ AD8554. The output overdrive recovery for an autocorrection amplifier is defined as the time it takes for the output to correct to its final voltage from an overload state. It is measured by placing the amplifier in a high gain configuration with an input signal that forces the output voltage to the supply rail. The input voltage is then stepped down to the linear region of the amplifier, usually to halfway between the supplies. The time from the input signal stepdown to the output settling to within 100 μV of its final value is the overdrive recovery time. INPUT OVERVOLTAGE PROTECTION Although the AD8551/AD8552/AD8554 are rail-to-rail input amplifiers, exercise care to ensure that the potential difference between the inputs does not exceed 5 V. Under normal operating conditions, the amplifier corrects its output to ensure the two inputs are at the same voltage. However, if the device is configured as a comparator, or is under some unusual operating condition, the input voltages may be forced to different potentials. This can cause excessive current to flow through internal diodes in the AD8551/AD8552/AD8554 used to protect the input stage against overvoltage. If either input exceeds either supply rail by more than 0.3 V, large amounts of current begin to flow through the ESD protection diodes in the amplifier. These diodes connect between the inputs and each supply rail to protect the input transistors against an electrostatic discharge event and are normally reverse-biased. However, if the input voltage exceeds the supply voltage, these ESD diodes become forward-biased. Without current limiting, excessive amounts of current can flow through these diodes, causing permanent damage to the device. If inputs are subjected to overvoltage, appropriate series resistors should be inserted to limit the diode current to less than 2 mA maximum. OUTPUT PHASE REVERSAL Output phase reversal occurs in some amplifiers when the input common-mode voltage range is exceeded. As common-mode voltage moves outside of the common-mode range, the outputs of these amplifiers suddenly jump in the opposite direction to the supply rail. This is the result of the differential input pair shutting down and causing a radical shifting of internal voltages, resulting in the erratic output behavior. The AD8551/AD8552/AD8554 amplifiers have been carefully designed to prevent any output phase reversal, provided both inputs are maintained within the supply voltages. If there is the potential of one or both inputs exceeding either supply voltage, place a resistor in series with the input to limit the current to less than 2 mA to ensure the output does not reverse its phase. Rev. F | Page 18 of 24 Data Sheet AD8551/AD8552/AD8554 CAPACITIVE LOAD DRIVE Table 5. Snubber Network Values for Driving Capacitive Loads The AD8551/AD8552/AD8554 have excellent capacitive load driving capabilities and can safely drive up to 10 nF from a single 5 V supply. Although the device is stable, capacitive loading limits the bandwidth of the amplifier. Capacitive loads also increase the amount of overshoot and ringing at the output. An R-C snubber network, shown in Figure 61, can be used to compensate the amplifier against capacitive load ringing and overshoot. CLOAD 1 nF 4.7 nF 10 nF 5V AD8551/ AD8552/ AD8554 RX 60Ω CX 0.47µF CL 4.7nF CX 1 nF 0.47 μF 10 μF POWER-UP BEHAVIOR At power-up, the AD8551/AD8552/AD8554 settle to a valid output within 5 μs. Figure 63 shows an oscilloscope photo of the output of the amplifier with the power supply voltage, and Figure 64 shows the test circuit. With the amplifier configured for unity gain, the device takes approximately 5 μs to settle to its final output voltage. This turn-on response time is much faster than most other autocorrection amplifiers, which can take hundreds of microseconds or longer for their output to settle. 01101-061 VOUT VIN 200mV p-p RX 200 Ω 60 Ω 20 Ω Figure 61. Snubber Network Configuration for Driving Capacitive Loads Although the snubber does not recover the loss of amplifier bandwidth from the load capacitance, it does allow the amplifier to drive larger values of capacitance while maintaining a minimum of overshoot and ringing. Figure 62 shows the output of an AD8551/ AD8552/AD8554 driving a 1 nF capacitor with and without a snubber network. VOUT 0V V+ 0V 10µs 1V WITH SNUBBER 01101-063 5µs BOTTOM TRACE = 2V/DIV TOP TRACE = 1V/DIV Figure 63. AD8551/AD8552/AD8554 Output Behavior on Power-Up 100kΩ VSY = 0V TO 5V WITHOUT SNUBBER 100mV 01101-062 VSY = 5V CLOAD = 4.7nF AD8551/ AD8552/ AD8554 01101-064 VOUT 100kΩ Figure 64. AD8551/AD8552/AD8554 Test Circuit for Turn-On Time Figure 62. Overshoot and Ringing are Substantially Reduced Using a Snubber Network The optimum value for the resistor and capacitor is a function of the load capacitance and is best determined empirically because actual CLOAD (CL) includes stray capacitances and may differ substantially from the nominal capacitive load. Table 5 shows some snubber network values that can be used as starting points. Rev. F | Page 19 of 24 AD8551/AD8552/AD8554 Data Sheet APPLICATIONS INFORMATION A 5 V PRECISION STRAIN GAGE CIRCUIT R2 The extremely low offset voltage of the AD8552 makes it an ideal amplifier for any application requiring accuracy with high gains, such as a weigh scale or strain gage. Figure 65 shows a configuration for a single-supply, precision, strain gage measurement system. (17) R3 IF 2 5V AV = 3 AD8552-A R4 100Ω (19) The high common-mode rejection, high open-loop gain, and operation down to 3 V of supply voltage makes the AD8551/ AD8552/AD8554 an excellent choice of op amp for discrete single-supply instrumentation amplifiers. The common-mode rejection ratio of the AD8551/AD8552/AD8554 is greater than 120 dB, but the CMRR of the system is also a function of the external resistor tolerances. The gain of the difference amplifier shown in Figure 66 is given as R2 − V 2 R 1 (20) (21) (18) 1 2δ (22) AD8554-A R 3 V INSTRUMENTATION AMPLIFIER VOUT × (V1 – V2) R1R4 + 2R2 R4 + R2 R3 2R1R4 − 2R2 R3 CMRRMIN = V2 Figure 65. A 5 V Precision Strain Gage Amplifier R 1 + 1 R 2 R1 In the three-op amp, instrumentation amplifier configuration shown in Figure 67, the output difference amplifier is set to unity gain with all four resistors equal in value. If the tolerance of the resistors used in the circuit is given as δ, the worst-case CMRR of the instrumentation amplifier is VOUT 0V TO 4.0V A1 NOTES 1. USE 0.1% TOLERANCE RESISTORS. R4 = V 1 R3 + R 4 R2 R2 R4 = R1 R3 CMRR = R2 100Ω R3 17.4kΩ , THEN VOUT = Due to finite component tolerance, the ratio between the four resistors is not exactly equal, and any mismatch results in a reduction of common-mode rejection from the system. Referring to Figure 66, the exact common-mode rejection ratio can be expressed as 20kΩ 40mV FULL-SCALE R1 Which sets the output voltage of the system to 4 AD8552-B R1 17.4kΩ 350Ω LOAD CELL REF192 6 12.0kΩ R2 In an ideal difference amplifier, the ratio of the resistors are set exactly equal to 01101-065 4.0V 2.5V A2 = VOUT = AV (V1 − V2) Using the values given in Figure 65, the output voltage linearly varies from 0 V with no strain to 4.0 V under full strain. 1kΩ R3 R4 Figure 66. Using the AD8551/AD8552/AD8554 as a Difference Amplifier where RB is the resistance of the load cell. Q1 2N2222 OR EQUIVALENT R4 AD8551/ AD8552/ AD8554 R R R R RG V1 VOUT R AD8554-C RTRIM AD8554-B VOUT = 1 + 2R (V1 – V2) RG 01101-067 2 × (R1 + R2 ) RB VOUT V1 01101-066 A REF192 provides a 2.5 V precision reference voltage for A2. The A2 amplifier boosts this voltage to provide a 4.0 V reference for the top of the strain gage resistor bridge. Q1 provides the current drive for the 350 Ω bridge network. A1 is used to amplify the output of the bridge with the full-scale output voltage equal to R1 V2 Figure 67. A Discrete Instrumentation Amplifier Configuration Consequently, using 1% tolerance resistors results in a worst-case system CMRR of 0.02, or 34 dB. Therefore, either high precision resistors or an additional trimming resistor, as shown in Figure 67, must be used to achieve high common-mode rejection. The value of this trimming resistor must be equal to the value of R multiplied by its tolerance. For example, using 10 kΩ resistors with 1% tolerance requires a series trimming resistor equal to 100 Ω. Rev. F | Page 20 of 24 Data Sheet AD8551/AD8552/AD8554 A HIGH ACCURACY THERMOCOUPLE AMPLIFIER R Monitor Output = R2 × SENSE R1 Figure 68 shows a K-type thermocouple amplifier configuration with cold junction compensation. Even from a 5 V supply, the AD8551 can provide enough accuracy to achieve a resolution of better than 0.02°C from 0°C to 500°C. D1 is used as a temperature measuring device to correct the cold junction error from the thermocouple and should be placed as close as possible to the two terminating junctions. With the thermocouple measuring tip immersed in a 0°C ice bath, R6 should be adjusted until the output is at 0 V. Figure 70 shows the low-side monitor equivalent. In this circuit, the input common-mode voltage to the AD8552 is at or near ground. Again, a 0.1 Ω resistor provides a voltage drop proportional to the return current. The output voltage is given as R VOUT = (V + ) − 2 × RSENSE × I L R1 5.000V R1 10.7kΩ R5 40.2kΩ R1 100Ω 5V 10µF + D1 3 2 0.1µF R2 2.74kΩ R6 200Ω R4 5.62kΩ V+ 3V R8 124kΩ 1N4148 K-TYPE THERMOCOUPLE 40.7µV/°C IL R3 53.6Ω R7 453Ω 2 – 3 + 8 1/2 AD8552 1 4 S 7 AD8551 0.1µF G M1 Si9433 1 D MONITOR OUTPUT 4 0V TO 5.00V (0°C TO 500°C) R2 2.49kΩ 01101-069 4 RSENSE 0.1Ω 3V Figure 69. A High-Side Load Current Monitor Figure 68. A Precision K-Type Thermocouple Amplifier with Cold Junction Compensation V+ PRECISION CURRENT METER R2 2.49kΩ Because of its low input bias current and superb offset voltage at single supply voltages, the AD8551/AD8552/AD8554 are excellent amplifiers for precision current monitoring. Its rail-torail input allows the amplifier to be used as either a high-side or low-side current monitor. Using both amplifiers in the AD8552 provides a simple method to monitor both current supply and return paths for load or fault detection. Figure 69 shows a high-side current monitor configuration. In this configuration, the input common-mode voltage of the amplifier is at or near the positive supply voltage. The rail-torail input of the amplifier provides a precise measurement even with the input common-mode voltage at the supply voltage. The CMOS input structure does not draw any input bias current, ensuring a minimum of measurement error. The 0.1 Ω resistor creates a voltage drop to the noninverting input of the AD8551/AD8552/AD8554. The output of the amplifier is corrected until this voltage appears at the inverting input. This creates a current through R1, which in turn flows through R2. The monitor output is given by VOUT Q1 V+ R1 100Ω RSENSE 0.1Ω 1/2 AD8552 RETURN TO GROUND 01101-070 REF02EZ 6 (24) For the component values shown in Figure 70, the output transfer function decreases from V+ at −2.5 V/A. 01101-068 0.1µF 2 (23) Using the components shown in Figure 69, the monitor output transfer function is 2.5 V/A. Using the values shown in Figure 68, the output voltage tracks temperature at 10 mV/°C. For a wider range of temperature measurement, R9 can be decreased to 62 kΩ. This creates a 5 mV/°C change at the output, allowing measurements of up to 1000°C. 12V × IL Figure 70. A Low-Side Load Current Monitor PRECISION VOLTAGE COMPARATOR The AD8551/AD8552/AD8554 can be operated open-loop and used as a precision comparator. The AD8551/AD8552/AD8554 have less than 50 μV of offset voltage when run in this configuration. The slight increase of offset voltage stems from the fact that the autocorrection architecture operates with lowest offset in a closed-loop configuration, that is, one with negative feedback. With 50 mV of overdrive, the device has a propagation delay of 15 μs on the rising edge and 8 μs on the falling edge. Ensure the maximum differential voltage of the device is not exceeded. For more information, refer to the Input Overvoltage Protection section. Rev. F | Page 21 of 24 AD8551/AD8552/AD8554 Data Sheet OUTLINE DIMENSIONS 3.20 3.00 2.80 8 3.20 3.00 2.80 5.15 4.90 4.65 5 1 4 PIN 1 IDENTIFIER 0.65 BSC 0.95 0.85 0.75 15° MAX 1.10 MAX 0.80 0.55 0.40 0.23 0.09 6° 0° 0.40 0.25 10-07-2009-B 0.15 0.05 COPLANARITY 0.10 COMPLIANT TO JEDEC STANDARDS MO-187-AA Figure 71. 8-Lead Mini Small Outline Package [MSOP] (RM-8) Dimensions shown in millimeters 3.10 3.00 2.90 8 5 4.50 4.40 4.30 1 6.40 BSC 4 PIN 1 0.65 BSC 0.15 0.05 1.20 MAX COPLANARITY 0.10 0.30 0.19 SEATING 0.20 PLANE 0.09 8° 0° 0.75 0.60 0.45 COMPLIANT TO JEDEC STANDARDS MO-153-AA Figure 72. 8-Lead Thin Shrink Small Outline Package [TSSOP] (RU-8) Dimensions shown in millimeters Rev. F | Page 22 of 24 Data Sheet AD8551/AD8552/AD8554 5.00 (0.1968) 4.80 (0.1890) 5 1 6.20 (0.2441) 5.80 (0.2284) 4 1.27 (0.0500) BSC 1.75 (0.0688) 1.35 (0.0532) 0.25 (0.0098) 0.10 (0.0040) 0.51 (0.0201) 0.31 (0.0122) COPLANARITY 0.10 SEATING PLANE 0.50 (0.0196) 0.25 (0.0099) 45° 8° 0° 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-AA CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 012407-A 8 4.00 (0.1574) 3.80 (0.1497) Figure 73. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches) 5.10 5.00 4.90 14 8 4.50 4.40 4.30 6.40 BSC 1 7 PIN 1 0.65 BSC 1.20 MAX 0.15 0.05 COPLANARITY 0.10 0.30 0.19 0.20 0.09 SEATING PLANE 8° 0° COMPLIANT TO JEDEC STANDARDS MO-153-AB-1 Figure 74. 14-Lead Thin Shrink Small Outline Package [TSSOP] (RU-14) Dimensions shown in millimeters Rev. F | Page 23 of 24 0.75 0.60 0.45 061908-A 1.05 1.00 0.80 AD8551/AD8552/AD8554 Data Sheet 8.75 (0.3445) 8.55 (0.3366) 8 14 1 7 1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0039) COPLANARITY 0.10 0.51 (0.0201) 0.31 (0.0122) 6.20 (0.2441) 5.80 (0.2283) 0.50 (0.0197) 0.25 (0.0098) 1.75 (0.0689) 1.35 (0.0531) SEATING PLANE 45° 8° 0° 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-AB CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. 060606-A 4.00 (0.1575) 3.80 (0.1496) Figure 75. 14-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-14) Dimensions shown in millimeters and (inches) ORDERING GUIDE Model1 AD8551ARZ AD8551ARZ-REEL AD8551ARZ-REEL7 AD8551ARM-REEL AD8551ARMZ AD8551ARMZ-REEL AD8552AR AD8552AR-REEL AD8552AR-REEL7 AD8552ARZ AD8552ARZ-REEL AD8552ARZ-REEL7 AD8552ARU AD8552ARU-REEL AD8552ARUZ AD8552ARUZ-REEL AD8554ARZ AD8554ARZ-REEL AD8554ARZ-REEL7 AD8554ARUZ AD8554ARUZ-REEL 1 Temperature Range −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C −40°C to +125°C Package Description 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead MSOP 8-Lead MSOP 8-Lead MSOP 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead SOIC_N 8-Lead TSSOP 8-Lead TSSOP 8-Lead TSSOP 8-Lead TSSOP 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead SOIC_N 14-Lead TSSOP 14-Lead TSSOP Z = RoHS Compliant Part, # denotes RoHS compliant part may be top or bottom marked. ©1999–2015 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D01101-0-6/15(F) Rev. F | Page 24 of 24 Package Option R-8 R-8 R-8 RM-8 RM-8 RM-8 R-8 R-8 R-8 R-8 R-8 R-8 RU-8 RU-8 RU-8 RU-8 R-14 R-14 R-14 RU-14 RU-14 Branding AHA AHA# AHA#