design ideas Edited by Bill Travis Make noise with a PIC Peter Guettler, APS Software Engineering GmbH, Cologne, Germany uilding a stable noise generator for audio-frequency purposes requires only a few components. The circuit in Figure 1 relies on linear-feedback shift registers and some simple software. An eight-pin Microchip (www. microchip.com) PIC12C508 controller (IC2) with a short program generates pseudorandom noise at its output pin, GP0. A single controller is sufficient for simple applications. To obtain Gaussiandistributed noise, you can use a number of identical PIC controllers in parallel in a true realization of the central-limit theorem. (The central-limit theorem states that the sum of an infinite number of noise sources has Gaussian distribution, regardless of the individual noise distribution of each generator.) Using an infinite number of noise generators is impractical, but 10 to 16 are sufficient in most cases. And, because the smallest member of the PIC family is an inexpensive chip with low current consumption, the circuit is easy to realize. All noise generators run the same program (Listing 1 on the Web version of this Design Idea at www.edn.com). Perfectionists might program each PIC with an individual initial value for the shift register, but, because all controllers run uncorrelated with their own internal oscillators and start out of reset at different times, this measure is unnecessary. Op amp IC1A sums and level-shifts the noise signals. Summing resistors R1 and R2 must have a value of 10 k times the number of noise generators you use. The output signal of IC1A feeds a 3-dB/octave filter to obtain pink noise. Buffer IC1B decouples the filter and provides low output impedance. The signal amplitude is B 5V approximately 400 mV p-p with a flat spectral distribution of 20 Hz to 20 kHz. Closing S1 or applying a low level at pin GP4 immediately stops all noise generators and freezes the prevailing signal amplitude. You can download the PIC software from the Web version of this Design Idea at www.edn.com.왏 Make noise with a PIC ..................................77 Circuit provides linear resistance-to-time conversion ....................78 PC-configurable RLC resonator yields single-output filter..........................................82 Single IC provides gains of 10 and 10 ..............................................86 Publish your Design Idea in EDN. See the What’s Up section at www.edn.com. 5V 10k 1 VDD GP5/CIN GP4/COUT GP3/MC IC2 GP2 PIC12C508P GP1 8 VSS GP0 2 3 4 5 6 7 R1 nR R 5V 5V ADDITIONAL PICs 8 10k 1 GP5/CIN GP4/COUT GP3/MC IC3 GP2 PIC12C508P GP1 8 GP0 VSS VDD 6 _ 2 _ 2 3 4 5 6 7 IC1A TL072P 3 + 1 2 0.1 µF DISABLE 220 VOUT PINK NOISE 400 mV P-P 300 1k 5V 3.3k nR S1 7 4 3k R2 IC1B TL072P 5 + 6.8k 1 µF 0.27 µF 47 nF 47 nF 33 nF 1 NOISE GENERATORS Figure 1 www.edn.com SUMMING AMPLIFIER –3-dB/OCTAVE FILTER OUTPUT BUFFER This simple circuit generates Gaussian-distributed noise for audio applications. August 7, 2003 | edn 77 design ideas Circuit provides linear resistance-to-time conversion S Kaliyugavaradan and D Arul Raj, Anna University, Chennai, India esistance-based transducers, such as strain gauges and piezoresistive devices, find common use in the measurement of several physical parameters. For applications in which digital processors or microcontrollers serve for data acquisition and signal processing, the transducer’s response must assume a form suitable for conversion to digital format. It is desirable to convert the resistance change of such sensors into a proportional frequency or a time interval so that you can easily obtain an output in digital form, using a counter/timer. The circuit of Figure1 linearly converts the sensor resistance, RS, into a proportional time period. The circuit is essentially a relaxation oscillator, comprising a current source, a bridge amplifier, a comparator, and a flip-flop. The current, IS, divides in the paths of R1 and R2 as if the two resistors were connected in parallel. Assuming ideal op amps, the circuit functions as an oscillator when RX (R4RS) is greater than R1R3/R2. The circuit produces waveforms at the input and output of the comparator, IC2 (Figure 2). T1 and T2 are the time intervals for which the comparator’s output assumes levels VS1 and VS2, respectively. The output voltage from IC2, with its lev- els changed via a zener-diode circuit, serves as clock input to a D flip-flop. From the 7474 VS1 flip-flop, you obtain a squarewave output that is high and COMPARATOR t low alternately for a time peOUTPUT VOLTAGE 0 T1T2 T1 riod T4C(R2RXR1R3)/R1. This equation indicates that VS2 the circuit converts a change in sensor resistance into a proportional time period T with sensitivity Figure 2 T/RS4C(R2/R1). The following salient features of Figure 1 merit mention: kIS ● The sensor is grounded; you can easily vary the conCOMPARATOR t INPUT VOLTAGE 0 version sensitivity by varying T1 T1T2 either R1 or R2. ● You can adjust the offset kIS value, T0 (about which changes in T occur because of a change in the sensor’s resistance), by changing These waveforms represent the input and output of either R3 or R4 without af- comparator IC2. fecting the conversion sensi● Thanks to the current source, the tivity. ● The offset voltages of the op amps al- output is largely insensitive to noise voltter T1 and T2 in opposite ways, such that ages in the line of the current source and their overall effect on T(T1T2) is not ap- to changes that occur in VS1 and VS2. preciable. Consider the example of converting a R OFFSET ADJUST R3 D2 D1 34.3k SENSITIVITY ADJUST IS 65.4k R1 2 3 D 6 4 5 OUTPUT 7474 7 3 + LF 411 + Q V0UT VCC 15V IC1 2k 2 _ 6 IC2 LF411 R5 Q 3 6 CLK 10k 4 D4 R4 1k RX Figure 1 1 µF 7 2 _ D3 FF VCC 15V R2 IN5287 C VEE –15V RS (SENSOR) Pt 100) VEE –15V 5V ZENER NOTE: D1, D2, D3, AND D4 ARE IN4002s. This simple circuit converts a resistance reading to a time period. 78 edn | August 7, 2003 www.edn.com design ideas Pt-100 (platinum RTD) sensor in the range of 119.4 to 138.51, which corresponds to a temperature range of 50 to 100C, into time periods of 10 to 12.5 msec. The design is simple. Because the current through the sensor is a fraction of IS, IS should be low enough to keep the self-heating error to an acceptably low level. This design uses an IN5287 current regulator; it provides an IS of approximately 0.33 mA and has a dynamic impedance better than 1.35 M. For a better current source, you could use a circuit based on a voltage-regulator IC. In the next step, with suitable and practical fixed values for R1 and C, you adjust R2 until you obtain the needed sensitivity: 130.82 sec/. Following this step, with a fixed R4, you adjust R3 to obtain the offset required in the output (T). Figure 1 shows the values of components for this example. The resistors all have 1% tolerance and 0.25W rating, and C is a polycarbonate capacitor.왏 PC-configurable RLC resonator yields single-output filter Saurav Gupta, New Delhi, India OUTPUT his Design Idea presTRANSCONDUCTANCE S5 ents a versatile filter cirAMPLIFIER D6 TO D9 S6 cuit for low-powerS7 consumption instrumenPC VOUT D2 TO D5 PORT S8 tation that you can proR gram from your PC using the parallel port. The cirD2 cuit uses analog switches 1k S1 and latches instead of digiD3 R tal potentiometers for the 1 2.2k S2 D4 digital control (figures 1 10k S3 and 2). By running simple C1 D5 VIN 100k S4 software code on the PC, 10V R you can configure a single robust design to work as a _ lowpass, highpass, or bandR2 + pass filter, and you can also select the desired center fre10V 10V quency, 0 (Listing 1). UnR3 like a similarly controllable design (Reference 1), this LP C2 design is a single-outFigure 1 10V put-at-a-time filter. PCMany power-sensitive sys- A PC-configurable filter uses a synthesized R5 PROGRAMMABLE tems do not require simul- inductor and analog switches to determine INDUCTOR taneous-filter functions. filter type and center frequency. The design exploits the fact that a series RLC resonator can pro- Figure 1, the inductor, LP, is implement- upon the state of switches S1 through S4 vide various filter functions with its ele- ed as a PC-controlled synthesized induc- (determined by PC-port data bits D2 ments. Because the design is based on an tor. The value of the inductor is through D5). The expression for the freRLC section, it is trivial to convert the de- LPC2RPR3R5/R2. Here, RP can assume quency is 0(R2/C1RPR3R5)1/2. You can sign into a PC-controlled resonator. In any of 15 possible values, depending thus effectively select 15 frequency values. (This design uses 12 values of practical interest.) Data bits D6 through D9 from TABLE 1—REPRESENTATIVE PORT SETTINGS AND FILTER PARAMETERS the PC’s parallel port set the state of anaPort setting Filter type/center Hex output log switches S5 through S8. The state of frequency D9 D8 D7 D6 D5 D4 D3 D2 from PC Lowpass/9.93 kHz 0 0 1 1 0 1 0 0 X34 the switches determines the type of filHighpass/22.9 kHz 1 0 1 0 0 1 1 0 XA6 ter. Bandpass/3.16 kHz 0 1 0 1 1 0 0 0 X58 Figure 3 shows the software-generated Bandpass/37.3 kHz 0 1 0 1 0 1 1 1 X57 display for the circuit. This design uses a T P _ + 82 edn | August 7, 2003 www.edn.com design ideas 1k 14 15 16 17 18 19 20 21 22 23 24 25 1 2 3 4 5 6 7 8 9 10 11 12 13 OC 1D 2D 3D 4D 5D 6D 7D 8D GND VCC 1Q 2Q 3Q 4Q 5Q 6Q 7Q 8Q EN 74573 100k IN1 D1 S1 V GND S4 D4 IN4 IN2 D2 S2 V NC S3 D3 IN3 2.2k _ + 10k DG308 C1 0.1 F IN1 D1 S1 V GND S4 D4 IN4 IN2 D2 S2 V NC S3 D3 IN3 R2 100k R3 100 C2 0.1 F R5 1k _ R1 4.7k 1/2 OPA2241 VIN DG308 50 Figure 2 You must take the finite on-resistance value of the analog switches into account in determining the center frequency of the filter. 1/2 OPA2241 50 500 V+ V+ IN+ IOUT Z+ V+ NC ISET Z V IN NC V V + VOUT 500 MAX436 9.93-kHz bandpass filter for demonstration and testing. Increasing the number of analog switches can provide a wider range. Moreover, you could use additional switches for gain programmability. The 74573 latch provides the interface to the PC. Table 1 shows the port/switch settings for a few frequency and filtertype selections. Note that the analog switches (DG308) have a finite operating on-resistance of approximately about 110; you must take this resistance into account when you calculate the center frequency. For precision instrumentation, other switches are available with operating on-resistances as low as 30 to 50. You can download Listing 1 from the Web version of this Design Idea at www.edn.com.왏 Figure 3 This user-friendly configuration screen allows you to determine filter type and frequency. 84 edn | August 7, 2003 Reference 1. Gupta, Saurav, and Tejinder Singh, “PC-based configurable filter uses no digital potentiometers,” EDN, Jan 23, 2003, pg 76. www.edn.com design ideas Single IC provides gains of 10 and 10 Moshe Gerstenhaber and Charles Kitchin, Analog Devices, Wilmington, MA eal-world data-acquisition systems require amplifying weak signals to match the full-scale input range of an A/D converter. Unfortunately, when you configure them as gain blocks, most common amplifiers have both gain errors and offset drift. The typical two-resistor gain-setting arrangement found in many op-amp circuits has serious accuracy and drift limitations. With standard 1% resistors, the circuit gain can be off by as much as 2%. Also, the gain can vary with temperature, because each resistor drifts differently. You can use monolithic resistor networks for precise gain setting, but these components are expensive and consume valuable pc-board space. The circuits of figures 1 and 2 offer improved performance and lower cost; they are also smaller. The R 86 edn | August 7, 2003 single-SOIC approach 15V is the smallest 0.1 F Figure 1 available for this 7 VS function, and the circuits 4 10k require no external components. Figure 1 shows AD628 8 100k 10k –IN an AD628 precision gain VOUT A1 +IN 5 A2 +IN block connected to pro1 100k –IN vide a voltage gain of 10. 10k The gain block itself VREF 3 RG 6 comprises two internal –VG 2 amplifiers: a gain-of-0.1 0.1 F difference amplifier, A1, VIN followed by an uncom–15V mitted buffer amplifier, This circuit has a precise gain of 10 and uses no external A2. You can configure it components. to provide different gains by strapping or grounding the appropri- nects between the VREF pin (Pin 3) and ground, instead of to the op amp’s inputs. ate pins. For a gain of 10, the input signal con- With the input tied to the VREF pin, the www.edn.com design ideas voltage at the noninverting input of A1 equals VIN(100 k/110 k), or VIN(10/11). The inverting input of A2 (Pin 6) is grounded; therefore, feedback from the output of A2 forces the noninverting input of A2 to be 0V. The output of A1 must then also be at 0V. The voltage on the inverting input of A1 must be equal to the voltage on the noninverting input of A1, so both equal VIN(10/11). Thus, the output voltage of A2, VOUT, equals VOUT = VIN × 10 100k × 1 + 11 10k 10 × 11 = 10 VIN , 11 providing a precise gain of 10 with no external components. The companion circuit of Figure 2 provides a gain of 10. This time, the input connects between the inverting input of A2 (Pin 6) and ground. Operation is similar to that of Figure 1, but A2 now in= VIN × 88 edn | August 7, 2003 15V verts the input signal 0.1 F Figure 2 by 180. With the VREF pin grounded, the 7 VS 4 noninverting input of A1 10k is at 0V, so feedback AD628 8 100k forces the inverting input 10k –IN VOUT A1 +IN 5 of A1 to 0V as well. BeA2 +IN 1 100k –IN cause A1 operates at a 10k gain of 0.1, the output of A2 necessary to force the VREF 3 RG 6 –VG 2 inverting input of A1 to 0.1 F 0V is 10VIN. The two connections exhibit difVIN –15V ferent input impedances. When you drive the VREF This companion circuit to the one in Figure 1 delivers input (Pin 3) for a gain of an accurate gain of 10. 10, the input impedance to ground is 110 k; it is approximately efficient lower than 5 ppm/C. The 350 G when you drive the noninverting dB bandwidth is approximately 110 kHz input of A2 (Pin 6) for a gain of 10. All with a 10-mV input and 95 kHz with a resistors are internal to the gain block, so 100-mV input. Although 15V supplies both accuracy and drift are excellent. are appropriate, you may operate these These circuits have gain accuracy better circuits with dual supplies from 2.25V than 0.1%, with a gain temperature co- to 18V.왏 www.edn.com