12-Bit, 20 MSPS/40 MSPS/65 MSPS/80 MSPS, 1.8 V Dual Analog-to-Digital Converter AD9231 FEATURES FUNCTIONAL BLOCK DIAGRAM AVDD ADC VIN–A VREF SENSE VCM AD9231 REF SELECT VIN–B ADC VIN+B CLK+ CLK– D11A D0A DCOA DRVDD CMOS OUTPUT BUFFER RBIAS ORA DIVIDE 1 TO 8 DUTY CYCLE STABILIZER MODE CONTROLS SYNC DCS PDWN DFS OEB ORB D11B D0B DCOB 08121-001 PROGRAMMING DATA VIN+A CMOS OUTPUT BUFFER SPI Figure 1. PRODUCT HIGHLIGHTS 1. APPLICATIONS Communications Diversity radio systems Multimode digital receivers GSM, EDGE, W-CDMA, LTE, CDMA2000, WiMAX, TD-SCDMA I/Q demodulation systems Smart antenna systems Battery-powered instruments Hand held scope meters Portable medical imaging Ultrasound Radar/LIDAR SDIO SCLK CSB GND MUX OPTION 1.8 V analog supply operation 1.8 V to 3.3 V output supply SNR 71.3 dBFS at 9.7 MHz input 69.0 dBFS at 200 MHz input SFDR 93 dBc at 9.7 MHz input 83 dBc at 200 MHz input Low power 32 mW per channel at 20 MSPS 71 mW per channel at 80 MSPS Differential input with 700 MHz bandwidth On-chip voltage reference and sample-and-hold circuit 2 V p-p differential analog input DNL = ±0.40 LSB Serial port control options Offset binary, gray code, or twos complement data format Optional clock duty cycle stabilizer Integer 1-to-8 input clock divider Data output multiplex option Built-in selectable digital test pattern generation Energy-saving power-down modes Data clock out with programmable clock and data alignment 2. 3. 4. The AD9231 operates from a single 1.8 V analog power supply and features a separate digital output driver supply to accommodate 1.8 V to 3.3 V logic families. The patented sample-and-hold circuit maintains excellent performance for input frequencies up to 200 MHz and is designed for low cost, low power, and ease of use. A standard serial port interface supports various product features and functions, such as data output formatting, internal clock divider, power-down, DCO/DATA timing and offset adjustments, and voltage reference modes. The AD9231 is packaged in a 64-lead RoHS compliant LFCSP that is pin compatible with the AD9268 16-bit ADC, the AD9258 14-bit ADC, the AD9251 14-bit ADC, and the AD9204 10-bit ADC, enabling a simple migration path between 10-bit and 16-bit converters sampling from 20 MSPS to 125 MSPS. Rev. A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2009–2010 Analog Devices, Inc. All rights reserved. AD9231 TABLE OF CONTENTS Features .............................................................................................. 1 Voltage Reference ....................................................................... 22 Applications....................................................................................... 1 Clock Input Considerations...................................................... 23 Functional Block Diagram .............................................................. 1 Channel/Chip Synchronization................................................ 25 Product Highlights ........................................................................... 1 Power Dissipation and Standby Mode .................................... 25 Revision History ............................................................................... 2 Digital Outputs ........................................................................... 26 General Description ......................................................................... 3 Timing ......................................................................................... 26 Specifications..................................................................................... 4 Built-In Self-Test (BIST) and Output Test .................................. 27 DC Specifications ......................................................................... 4 Built-In Self-Test (BIST)............................................................ 27 AC Specifications.......................................................................... 5 Output Test Modes..................................................................... 27 Digital Specifications ................................................................... 6 Serial Port Interface (SPI).............................................................. 28 Switching Specifications .............................................................. 7 Configuration Using the SPI..................................................... 28 Timing Specifications .................................................................. 8 Hardware Interface..................................................................... 29 Absolute Maximum Ratings............................................................ 9 Configuration Without the SPI ................................................ 29 Thermal Characteristics .............................................................. 9 SPI Accessible Features.............................................................. 29 ESD Caution.................................................................................. 9 Memory Map .................................................................................. 30 Pin Configuration and Function Descriptions........................... 10 Reading the Memory Map Register Table............................... 30 Typical Performance Characteristics ........................................... 12 Open Locations .......................................................................... 30 AD9231-80 .................................................................................. 12 Default Values ............................................................................. 30 AD9231-65 .................................................................................. 14 Memory Map Register Table..................................................... 31 AD9231-40 .................................................................................. 15 Memory Map Register Descriptions........................................ 33 AD9231-20 .................................................................................. 16 Applications Information .............................................................. 34 Equivalent Circuits ......................................................................... 17 Design Guidelines ...................................................................... 34 Theory of Operation ...................................................................... 19 Outline Dimensions ....................................................................... 35 ADC Architecture ...................................................................... 19 Ordering Guide .......................................................................... 35 Analog Input Considerations.................................................... 19 REVISION HISTORY 6/10—Rev. 0 to Rev. A Changes to Features Section............................................................ 1 10/09—Revision 0: Initial Version Rev. A | Page 2 of 36 AD9231 GENERAL DESCRIPTION The AD9231 is a monolithic, dual-channel, 1.8 V supply, 12-bit, 20 MSPS/40 MSPS/65 MSPS/80 MSPS analog-to-digital converter (ADC). It features a high performance sample-and-hold circuit and on-chip voltage reference. A differential clock input controls all internal conversion cycles. An optional duty cycle stabilizer (DCS) compensates for wide variations in the clock duty cycle while maintaining excellent overall ADC performance. The product uses multistage differential pipeline architecture with output error correction logic to provide 12-bit accuracy at 80 MSPS data rates and to guarantee no missing codes over the full operating temperature range. The digital output data is presented in offset binary, gray code, or twos complement format. A data output clock (DCO) is provided for each ADC channel to ensure proper latch timing with receiving logic. Both 1.8 V and 3.3 V CMOS levels are supported, and output data can be multiplexed onto a single output bus. The ADC contains several features designed to maximize flexibility and minimize system cost, such as programmable clock and data alignment and programmable digital test pattern generation. The available digital test patterns include built-in deterministic and pseudorandom patterns, along with custom user-defined test patterns entered via the serial port interface (SPI). The AD9231 is available in a 64-lead RoHS compliant LFCSP and is specified over the industrial temperature range (−40°C to +85°C). Rev. A | Page 3 of 36 AD9231 SPECIFICATIONS DC SPECIFICATIONS AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. Table 1. Parameter RESOLUTION ACCURACY No Missing Codes Offset Error Gain Error 1 Differential Nonlinearity (DNL) 2 Integral Nonlinearity (INL)2 MATCHING CHARACTERISTICS Offset Error Gain Error1 TEMPERATURE DRIFT Offset Error INTERNAL VOLTAGE REFERENCE Output Voltage (1 V Mode) Load Regulation Error at 1.0 mA INPUT-REFERRED NOISE VREF = 1.0 V ANALOG INPUT Input Span, VREF = 1.0 V Input Capacitance 3 Input Common-Mode Voltage Input Common-Mode Range REFERENCE INPUT RESISTANCE POWER SUPPLIES Supply Voltage AVDD DRVDD Supply Current IAVDD2 IDRVDD2 (1.8 V) IDRVDD2 (3.3 V) POWER CONSUMPTION DC Input Sine Wave Input2 (DRVDD = 1.8 V) Sine Wave Input2 (DRVDD = 3.3 V) Standby Power 4 Power-Down Power Temp Full AD9231-20/AD9231-40 Min Typ Max 12 Full Full Full Full 25°C Full 25°C Guaranteed 0.05 ±0.5 −1.5 ±0.30 ±0.12 ±0.45 ±0.15 25°C 25°C ±0.0 0.3 Full ±2 Full Full 0.981 Min 12 AD9231-65 Typ Max Min 12 Guaranteed 0.05 ±0.5 −1.5 ±0.40 ±0.17 ±0.50 ±0.17 ±0.70 ±0.0 0.3 AD9231-80 Typ Max Guaranteed 0.05 ±0.5 −1.5 ±0.40 ±0.2 ±0.65 ±0.2 ±0.60 ±0.0 0.4 ±2 0.993 2 1.005 0.981 0.993 2 ±0.60 ±2 1.005 0.981 0.993 2 Unit Bits % FSR % FSR LSB LSB LSB LSB % FSR % FSR ppm/°C 1.005 V mV 25°C 0.25 0.25 0.25 LSB rms Full Full Full Full Full 2 6 0.9 2 6 0.9 2 6 0.9 V p-p pF V V kΩ Full Full 0.5 1.3 0.5 7.5 1.7 1.7 1.3 0.5 7.5 1.8 1.9 3.6 Full Full Full 35.7/49.0 3.0/5.1 5.9/10.1 37.7/52.2 Full Full Full Full Full 63.5/87.1 69.7/97.3 83.7/121.5 37/37 2.2 73.3/103.0 1 1.7 1.7 1.8 1.9 3.6 69 7.4 14.9 72.4 122.9 138.0 173.4 37 2.2 1.7 1.7 143.8 Measured with 1.0 V external reference. Measured with a 10 MHz input frequency at rated sample rate, full-scale sine wave, with approximately 5 pF loading on each output bit. Input capacitance refers to the effective capacitance between one differential input pin and AGND. 4 Standby power is measured with a dc input and the CLK active. 2 3 Rev. A | Page 4 of 36 1.3 7.5 1.8 1.9 3.6 V V 80.0 9.1 18.3 83.4 mA mA mA 141.8 160.4 204 37 2.2 166.5 mW mW mW mW mW AD9231 AC SPECIFICATIONS AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. Table 2. Parameter 1 SIGNAL-TO-NOISE RATIO (SNR) fIN = 9.7 MHz fIN = 30.5 MHz fIN = 70 MHz fIN = 200 MHz SIGNAL-TO-NOISE-AND-DISTORTION (SINAD) fIN = 9.7 MHz fIN = 30.5 MHz fIN = 70 MHz fIN = 200 MHz EFFECTIVE NUMBER OF BITS (ENOB) fIN = 9.7 MHz fIN = 30.5 MHz fIN = 70 MHz fIN = 200 MHz WORST SECOND OR THIRD HARMONIC fIN = 9.7 MHz fIN = 30.5 MHz fIN = 70 MHz fIN = 200 MHz SPURIOUS-FREE DYNAMIC RANGE (SFDR) fIN = 9.7 MHz fIN = 30.5 MHz fIN = 70 MHz fIN = 200 MHz WORST OTHER (HARMONIC OR SPUR) fIN = 9.7 MHz fIN = 30.5 MHz fIN = 70 MHz fIN = 200 MHz TWO-TONE SFDR fIN = 28.3 MHz (−7 dBFS), 30.6 MHz (−7 dBFS) CROSSTALK 2 ANALOG INPUT BANDWIDTH 1 2 Temp 25°C 25°C Full 25°C Full 25°C 25°C 25°C Full 25°C Full 25°C AD9231-20/AD9231-40 Min Typ Max AD9231-65 Min Typ Max AD9231-80 Min Typ Max 71.4 71.3 71.3 71.2 70.7/71.5 70.6/71.3 70.1/70.7 dBFS dBFS dBFS dBFS dBFS dBFS 70.5 70.5/71.0 71.0 70.9 70.1 69.0 69.0 69.0 70.6/71.4 70.6/71.2 71.3 71.2 71.2 71.1 Unit 68 68 68 dBFS dBFS dBFS dBFS dBFS dBFS 25°C 25°C 25°C 25°C 11.4/11.6 11.4/11.5 11.4/11.5 11.0 11.6 11.5 11.5 11.0 11.5 11.5 11.5 11.0 Bits Bits Bits Bits 25°C 25°C Full 25°C Full 25°C −95 −95 −95 −95 −93 −93 dBc dBc dBc dBc dBc dBc 25°C 25°C Full 25°C Full 25°C 70.1/70.6 70.0 70.4/70.9 70.9 70.8 70.0 −81 −81 −92/−94 −94 −92 −83 −83 −83 95 95 95 95 93 93 −81 81 dBc dBc dBc dBc dBc dBc 81 92/94 94 92 81 83 83 83 25°C 25°C Full 25°C Full 25°C −98 −97/−98 −98 −98 −97 −97 25°C Full 25°C −97/−98 −98 −95 −92 −92 −92 dBc dBc dBc dBc dBc dBc 90 −110 700 90 −110 700 90 −110 700 dBc dBc MHz −90 −90 −89 See the AN-835 Application Note, Understanding High Speed ADC Testing and Evaluation, for a complete set of definitions. Crosstalk is measured at 100 MHz with −1.0 dBFS on one channel and no input on the alternate channel. Rev. A | Page 5 of 36 AD9231 DIGITAL SPECIFICATIONS AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. Table 3. Parameter DIFFERENTIAL CLOCK INPUTS (CLK+, CLK−) Logic Compliance Internal Common-Mode Bias Differential Input Voltage Input Voltage Range High Level Input Current Low Level Input Current Input Resistance Input Capacitance LOGIC INPUTS (SCLK/DFS, SYNC, PDWN) 1 High Level Input Voltage Low Level Input Voltage High Level Input Current Low Level Input Current Input Resistance Input Capacitance LOGIC INPUTS (CSB) 2 High Level Input Voltage Low Level Input Voltage High Level Input Current Low Level Input Current Input Resistance Input Capacitance LOGIC INPUTS (SDIO1/DCS2) High Level Input Voltage Low Level Input Voltage High Level Input Current Low Level Input Current Input Resistance Input Capacitance DIGITAL OUTPUTS DRVDD = 3.3 V High Level Output Voltage, IOH = 50 μA High Level Output Voltage, IOH = 0.5 mA Low Level Output Voltage, IOL = 1.6 mA Low Level Output Voltage, IOL = 50 μA DRVDD = 1.8 V High Level Output Voltage, IOH = 50 μA High Level Output Voltage, IOH = 0.5 mA Low Level Output Voltage, IOL = 1.6 mA Low Level Output Voltage, IOL = 50 μA 1 2 Temp Full Full Full Full Full Full Full Min AD9231-20/AD9231-40/AD9231-65/AD9231-80 Typ Max CMOS/LVDS/LVPECL 0.9 0.2 GND − 0.3 −10 −10 8 Full Full Full Full Full Full 1.2 0 −50 −10 Full Full Full Full Full Full 1.2 0 −10 40 Full Full Full Full Full Full 1.2 0 −10 40 Full Full Full Full 3.29 3.25 Full Full Full Full 1.79 1.75 10 4 3.6 AVDD + 0.2 +10 +10 12 V V μA μA kΩ pF DRVDD + 0.3 0.8 +10 135 V V μA μA kΩ pF DRVDD + 0.3 0.8 +10 130 V V μA μA kΩ pF 26 2 26 5 Rev. A | Page 6 of 36 V V p-p V μA μA kΩ pF DRVDD + 0.3 0.8 −75 +10 30 2 Internal 30 kΩ pull-down. Internal 30 kΩ pull-up. Unit 0.2 0.05 V V V V 0.2 0.05 V V V V AD9231 SWITCHING SPECIFICATIONS AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. Table 4. Parameter CLOCK INPUT PARAMETERS Input Clock Rate Conversion Rate 1 CLK Period—Divide-by-1 Mode (tCLK) CLK Pulse Width High (tCH) Aperture Delay (tA) Aperture Uncertainty (Jitter, tJ) DATA OUTPUT PARAMETERS Data Propagation Delay (tPD) DCO Propagation Delay (tDCO) DCO to Data Skew (tSKEW) Pipeline Delay (Latency) Wake-Up Time 2 Standby OUT-OF-RANGE RECOVERY TIME 625 20/40 Min AD9231-65 Typ Max AD9231-80 Typ Max Full Full 25.0/12.5 1.0 0.1 7.69 1.0 0.1 6.25 1.0 0.1 Full Full Full Full Full Full Full 3 3 0.1 9 350 600/400 2 3 3 0.1 9 350 300 2 3 3 0.1 9 350 260 2 ns ns ns Cycles μs ns Cycles 50/25 3 15.38 625 80 Unit MHz MSPS ns ns ns ps rms 3 625 65 Min 3 12.5 Conversion rate is the clock rate after the CLK divider. Wake-up time is dependent on the value of the decoupling capacitors. N–1 N+4 tA N+5 N N+3 VIN N+1 tCH N+2 tCLK CLK+ CLK– tDCO DCOA/DCOB tSKEW CH A/CH B DATA N–9 N–8 N–7 N–6 N–5 tPD 08121-002 2 Full Full Full AD9231-20/AD9231-40 Min Typ Max Figure 2. CMOS Output Data Timing N–1 N+4 tA N+5 N N+3 VIN N+1 tCH N+2 tCLK CLK+ CLK– tDCO DCOA/DCOB tSKEW CH A N–9 CH A/CH B DATA CH B N–9 CH A N–8 CH B N–8 CH A N–7 tPD Figure 3. CMOS Interleaved Output Timing Rev. A | Page 7 of 36 CH B N–7 CH A N–6 CH B N–6 CH A N–5 08121-003 1 Temp AD9231 TIMING SPECIFICATIONS Table 5. tDIS_SDIO Conditions Min SYNC to rising edge of CLK setup time SYNC to rising edge of CLK hold time Setup time between the data and the rising edge of SCLK Hold time between the data and the rising edge of SCLK Period of the SCLK Setup time between CSB and SCLK Hold time between CSB and SCLK SCLK pulse width high SCLK pulse width low Time required for the SDIO pin to switch from an input to an output relative to the SCLK falling edge Time required for the SDIO pin to switch from an output to an input relative to the SCLK rising edge Figure 4. SYNC Input Timing Requirements Rev. A | Page 8 of 36 Unit ns ns ns ns ns ns ns ns ns ns 10 ns tHSYNC SYNC Max 2 2 40 2 2 10 10 10 CLK+ tSSYNC Typ 0.24 0.40 08121-004 Parameter SYNC TIMING REQUIREMENTS tSSYNC tHSYNC SPI TIMING REQUIREMENTS tDS tDH tCLK tS tH tHIGH tLOW tEN_SDIO AD9231 ABSOLUTE MAXIMUM RATINGS THERMAL CHARACTERISTICS Table 6. Parameter AVDD to AGND DRVDD to AGND VIN+A, VIN+B, VIN−A, VIN−B to AGND CLK+, CLK− to AGND SYNC to AGND VREF to AGND SENSE to AGND VCM to AGND RBIAS to AGND CSB to AGND SCLK/DFS to AGND SDIO/DCS to AGND OEB to AGND PDWN to AGND D0A/D0B through D11A/D11B to AGND DCOA/DCOB to AGND Operating Temperature Range (Ambient) Maximum Junction Temperature Under Bias Storage Temperature Range (Ambient) Rating −0.3 V to +2.0 V −0.3 V to +3.9 V −0.3 V to AVDD + 0.2 V −0.3 V to AVDD + 0.2 V −0.3 V to DRVDD + 0.3 V −0.3 V to AVDD + 0.2 V −0.3 V to AVDD + 0.2 V −0.3 V to AVDD + 0.2 V −0.3 V to AVDD + 0.2 V −0.3 V to DRVDD + 0.3 V −0.3 V to DRVDD + 0.3 V −0.3 V to DRVDD + 0.3 V −0.3 V to DRVDD + 0.3 V −0.3 V to DRVDD + 0.3 V −0.3 V to DRVDD + 0.3 V −0.3 V to DRVDD + 0.3 V −40°C to +85°C 150°C −65°C to +150°C Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. The exposed paddle is the only ground connection for the chip. The exposed paddle must be soldered to the AGND plane of the user’s circuit board. Soldering the exposed paddle to the user’s board also increases the reliability of the solder joints and maximizes the thermal capability of the package. Table 7. Thermal Resistance Package Type 64-Lead LFCSP 9 mm × 9 mm (CP-64-4) Airflow Velocity (m/sec) 0 1.0 2.5 θJA1, 2 23 20 18 θJC1, 3 2.0 θJB1, 4 12 Unit °C/W °C/W °C/W 1 Per JEDEC 51-7, plus JEDEC 25-5 2S2P test board. Per JEDEC JESD51-2 (still air) or JEDEC JESD51-6 (moving air). Per MIL-Std 883, Method 1012.1. 4 Per JEDEC JESD51-8 (still air). 2 3 Typical θJA is specified for a 4-layer PCB with a solid ground plane. As shown in Table 7, airflow improves heat dissipation, which reduces θJA. In addition, metal in direct contact with the package leads from metal traces, through holes, ground, and power planes, reduces the θJA. ESD CAUTION Rev. A | Page 9 of 36 AD9231 64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49 AVDD AVDD VIN+B VIN–B AVDD AVDD RBIAS VCM SENSE VREF AVDD AVDD VIN–A VIN+A AVDD AVDD PIN CONFIGURATION AND FUNCTION DESCRIPTIONS 1 2 3 4 5 6 7 8 9 10 11 12 13 14 15 16 PIN 1 INDICATOR AD9231 TOP VIEW (Not to Scale) 48 47 46 45 44 43 42 41 40 39 38 37 36 35 34 33 PDWN OEB CSB SCLK/DFS SDIO/DCS ORA D11A (MSB) D10A D9A D8A D7A DRVDD D6A D5A D4A D3A 08121-005 D8B D9B DRVDD D10B (MSB) D11B ORB DCOB DCOA NC NC NC DRVDD NC (LSB) D0A D1A D2A 17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32 CLK+ CLK– SYNC NC NC NC NC (LSB) D0B D1B DRVDD D2B D3B D4B D5B D6B D7B NOTES 1. NC = NO CONNECT. 2. THE EXPOSED PADDLE MUST BE SOLDERED TO THE PCB GROUND TO ENSURE PROPER HEAT DISSIPATION, NOISE, AND MECHANICAL STRENGTH BENEFITS. Figure 5. Pin Configuration Table 8. Pin Function Description Pin No. 0 1, 2 3 4, 5, 6, 7, 25, 26, 27, 29 8 to 9, 11 to 18, 20, 21 10, 19, 28, 37 22 23 24 30 to 36, 38 to 42 43 44 Mnemonic GND CLK+, CLK− SYNC NC D0B to D11B DRVDD ORB DCOB DCOA D0A to D11A ORA SDIO/DCS 45 SCLK/DFS 46 47 CSB OEB 48 PDWN Description Exposed paddle is the only ground connection for the chip. Must be connected to PCB AGND. Differential Encode Clock. PECL, LVDS, or 1.8 V CMOS inputs. Digital Input. SYNC input to clock divider. 30 kΩ internal pull-down. Do Not Connect. Channel B Digital Outputs. D11B = MSB. Digital Output Driver Supply (1.8 V to 3.3 V). Channel B Out-of-Range Digital Output. Channel B Data Clock Digital Output. Channel A Data Clock Digital Output. Channel A Digital Outputs. D11A = MSB. Channel A Out-of-Range Digital Output. SPI Data Input/Output (SDIO). Bidirectional SPI Data I/O in SPI mode. 30 kΩ internal pulldown in SPI mode. Duty Cycle Stabilizer (DCS). Static enable input for duty cycle stabilizer in non-SPI mode. 30 kΩ internal pull-up in non-SPI (DCS) mode. SPI Clock (SCLK) Input in SPI mode. 30 kΩ internal pull-down. Data Format Select (DFS). Static control of data output format in non-SPI mode. 30 kΩ internal pull-down. DFS high = twos complement output. DFS low = offset binary output. SPI Chip Select. Active low enable; 30 kΩ internal pull-up. Digital Input. Enable Channel A and Channel B digital outputs if low, tristate outputs if high. 30 kΩ internal pull-down. Digital Input. 30 kΩ internal pull-down. PDWN high = power-down device. PDWN low = run device, normal operation. Rev. A | Page 10 of 36 AD9231 Pin No. 49, 50, 53, 54, 59, 60, 63, 64 51, 52 55 56 57 58 61, 62 Mnemonic AVDD VIN+A, VIN−A VREF SENSE VCM RBIAS VIN−B, VIN+B Description 1.8 V Analog Supply Pins. Channel A Analog Inputs. Voltage Reference Input/Output. Reference Mode Selection. Analog output voltage at midsupply to set common mode of the analog inputs. Sets Analog Current Bias. Connect to 10 kΩ (1% tolerance) resistor to ground. Channel B Analog Inputs. Rev. A | Page 11 of 36 AD9231 TYPICAL PERFORMANCE CHARACTERISTICS AD9231-80 AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. 0 0 80MSPS 9.7MHz @ –1dBFS SNR = 70.2dB (71.2dBFS) SFDR = 93.6dBc –30 –45 –60 –75 –90 + 2 3 6 –105 5 4 8 12 16 20 24 28 FREQUENCY (MHz) 32 36 40 –135 5 0 4 8 12 16 20 24 28 FREQUENCY (MHz) 32 36 40 Figure 9. AD9231-80 Single-Tone FFT with fIN = 100.3 MHz 80MSPS 30.6MHz @ –1dBFS SNR = 70.1dB (71.1dBFS) SFDR = 94.4dBc –30 –45 –60 –75 –90 –105 + 3 5 80MSPS 210.3MHz @ –1dBFS SNR = 67.9dB (68.9dBFS) SFDR = 83.2dBc –15 AMPLITUDE (dBFS) –30 6 2 –45 –60 –75 2 3 –90 + 6 5 4 4 –105 –120 –120 4 8 12 16 20 24 28 FREQUENCY (MHz) 32 36 40 –135 08121-055 0 Figure 7. AD9231-80 Single-Tone FFT with fIN = 30.6 MHz 0 4 8 12 16 20 24 28 FREQUENCY (MHz) 32 36 40 08121-058 AMPLITUDE (dBFS) 2 6 + 4 0 –15 Figure 10. AD9231-80 Single-Tone FFT with fIN = 210.3 MHz 0 0 80MSPS 69MHz @ –1dBFS SNR = 69.9dB (70.9dBFS) SFDR = 94.3dBc –15 –30 –45 –60 –75 –90 + 6 –105 2 3 5 80MSPS 28.3 @ –7dBFS 30.6 @ –7dBFS SFDR = 90dBc –15 AMPLITUDE (dBFS) –30 4 –45 –60 –75 –90 F2 – F1 2F2 – F1 2F1 + F2 –105 –120 F1 + F2 2F2 – F1 2F1 – F2 –120 0 4 8 12 16 20 24 28 FREQUENCY (MHz) 32 36 Figure 8. AD9231-80 Single-Tone FFT with fIN = 69 MHz 40 –135 08121-056 AMPLITUDE (dBFS) 3 –90 08121-057 4 08121-054 0 0 –135 –75 –120 Figure 6. AD9231-80 Single-Tone FFT with fIN = 9.7 MHz –135 –60 –105 –120 –135 –45 0 4 8 12 16 20 24 28 FREQUENCY (MHz) 32 36 40 08121-059 AMPLITUDE (dBFS) –30 80MSPS 100.3MHz @ –1dBFS SNR = 69.5dB (70.5dBFS) SFDR = 87.7dBc –15 AMPLITUDE (dBFS) –15 Figure 11. AD9231-80 Two-Tone FFT with fIN1 = 28.3 MHz and fIN2 = 30.6 MHz Rev. A | Page 12 of 36 AD9231 AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. 100 0 90 SFDR 80 SFDR (dBc) SNRFS/SFDR (dBFS/dBc) SFDR/IMD3 (dBc/dBFS) –20 –40 IMD3 (dBc) –60 –80 SFDR (dBFS) SNRFS 70 60 50 40 30 20 –100 10 –50 –40 –30 –20 INPUT AMPLITUDE (dBFS) –10 Figure 12. Two-Tone SFDR/IMD3 vs. Input Amplitude (AIN) with fIN1 = 28.3 MHz and fIN2 = 30.6 MHz 20 30 40 50 60 70 80 Figure 15. AD9231-80 SNR/SFDR vs. Sample Rate with AIN = 9.7 MHz 100 SFDR 90 90 80 80 SNR 70 10 SAMPLE RATE (MHz) SNR/SFDR (dBc AND dBFS) 60 50 40 30 20 SFDRFS SNRFS 70 60 SFDR 50 SNR 40 30 20 10 10 0 50 100 150 INPUT FREQUENCY (MHz) 200 0 –70 08121-061 0 Figure 13. AD9231-80 SNR/SFDR vs. Input Frequency (AIN) with 2 V p-p Full Scale –60 –50 –40 –30 –20 INPUT AMPLITUDE (dBc) –10 0 08121-064 SNR/SFDR (dBFS/dBc) 100 0 08121-060 –60 08121-062 IMD3 (dBFS) –120 –70 Figure 16. AD9231-80 SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz 0.4 0.3 0.2 INL ERROR (LSB) 0 –0.1 0 –0.2 –0.3 0 500 1000 1500 2000 2500 OUTPUT CODE 3000 3500 4000 0.4 0 500 1000 1500 2000 2500 OUTPUT CODE 3000 3500 Figure 17. AD9231-80 INL with fIN = 9.7 MHz Figure 14. AD9231-80 DNL Error with fIN = 9.7 MHz Rev. A | Page 13 of 36 4000 08121-066 –0.2 08121-063 DNL ERROR (LSB) 0.2 0.1 AD9231 AD9231-65 AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. 0 120 65MSPS 9.7MHz @ –1dBFS SNR =70.3 (71.3dBFS) SFDR = 94.2dBc AMPLITUDE (dBFS) –30 –45 –60 –75 –90 + 5 6 –105 2 3 4 SFDRFS 100 SNR/SFDR (dBc AND dBFS) –15 80 SNRFS 60 SFDR 40 SNR 20 0 3 6 9 12 15 18 21 FREQUENCY (MHz) 24 27 30 33 0 –70 08121-067 Figure 18. AD9231-65 Single-Tone FFT with fIN = 9.7 MHz 0 –45 –60 –75 –90 + 2 3 5 6 4 –105 SFDR 80 SNR/SFDR (dBFS/dBc) AMPLITUDE (dBFS) 90 65MSPS 30.6MHz @ –1dBFS SNR = 70.2dB (71.2dBFS) SFDR = 94.1dBc –30 SNR 70 60 50 40 30 20 10 –120 3 6 9 12 15 18 21 FREQUENCY (MHz) 24 27 30 33 0 08121-069 0 65MSPS 69MHz @ –1dBFS SNR = 69.9dB (70.9dBFS) SFDR = 92.0dBc –30 –45 –60 –75 –90 + 2 3 –105 4 5 6 3 6 9 12 15 18 21 FREQUENCY (MHz) 24 27 30 33 08121-068 –120 0 50 100 150 200 Figure 22. AD9231-65 SNR/SFDR vs. Input Frequency (AIN) with 2 V p-p Full Scale 0 –15 0 INPUT FREQUENCY (MHz) Figure 19. AD9231-65 Single-Tone FFT with fIN = 30.6 MHz AMPLITUDE (dBFS) –10 100 –15 –135 –50 –40 –30 –20 INPUT AMPLITUDE (dBc) Figure 21. AD9231-65 SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz 0 –135 –60 Figure 20. AD9231-65 Single-Tone FFT with fIN = 69 MHz Rev. A | Page 14 of 36 08121-071 –135 08121-070 –120 AD9231 AD9231-40 AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. 0 120 40MSPS 9.7MHz @ –1dBFS SNR = 70.3dB (71.3dBFS) SFDR = 93.8dBc AMPLITUDE (dBFS) –30 –45 –60 –75 –90 5 4 –105 SFDRFS 100 SNR/SFDR (dBc AND dBFS) –15 + 3 2 6 80 SNRFS 60 SFDR 40 SNR 20 0 2 4 6 8 10 12 14 FREQUENCY (MHz) 16 18 20 0 –70 08121-072 –135 Figure 23. AD9231-40 Single-Tone FFT with fIN = 9.7 MHz 40MSPS 30.6MHz @ –1dBFS SNR = 70.2dB (71.2dBFS) SFDR = 95.4dBc –45 –60 –75 –90 + 5 4 –105 3 2 6 –120 –135 0 2 4 6 8 10 12 14 FREQUENCY (MHz) 16 18 20 08121-073 AMPLITUDE (dBFS) –30 –50 –40 –30 –20 INPUT AMPLITUDE (dBc) –10 0 Figure 25. AD9231-40 SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz 0 –15 –60 08121-074 –120 Figure 24. AD9231-40 Single-Tone FFT with fIN = 30.6 MHz Rev. A | Page 15 of 36 AD9231 AD9231-20 AVDD = 1.8 V; DRVDD = 1.8 V, maximum sample rate, 2 V p-p differential input, 1.0 V internal reference; AIN = −1.0 dBFS, DCS disabled, unless otherwise noted. 0 120 20MSPS 9.7MHz @ –1dBFS SNR = 70.3dB (71.3dBFS) SFDR = 94.1dBc AMPLITUDE (dBFS) –30 –45 –60 –75 –90 + 2 4 –105 SFDRFS 100 SNR/SFDR (dBc AND dBFS) –15 5 3 6 80 SNRFS 60 SFDR 40 SNR 20 0 0.95 1.90 2.85 3.80 4.75 5.70 6.65 7.60 8.55 9.50 FREQUENCY (MHz) 0 –70 08121-075 –135 Figure 26. AD9231-20 Single-Tone FFT with fIN = 9.7 MHz 20MSPS 30.6MHz @ –1dBFS SNR = 70.2dB (71.2dBFS) SFDR = 94.6dBc –45 –60 –75 –90 2 –105 + 4 6 5 3 –120 –135 0 0.95 1.90 2.85 3.80 4.75 5.70 6.65 7.60 8.55 9.50 FREQUENCY (MHz) 08121-076 AMPLITUDE (dBFS) –30 –50 –40 –30 –20 INPUT AMPLITUDE (dBc) –10 0 Figure 28. AD9231-20 SNR/SFDR vs. Input Amplitude (AIN) with fIN = 9.7 MHz 0 –15 –60 08121-077 –120 Figure 27. AD9231-20 Single-Tone FFT with fIN = 30.6 MHz Rev. A | Page 16 of 36 AD9231 EQUIVALENT CIRCUITS DRVDD AVDD 08121-042 08121-039 VIN±x Figure 29. Equivalent Analog Input Circuit Figure 32. Equivalent Digital Output Circuit 5Ω CLK+ 15kΩ 0.9V 15kΩ DRVDD 5Ω CLK– 350Ω 30kΩ 08121-040 08121-043 SCLK/DFS, SYNC, OEB, AND PDWN Figure 33. Equivalent SCLK/DFS, SYNC, OEB, and PDWN Input Circuit Figure 30. Equivalent Clock Input Circuit AVDD AVDD DRVDD 30kΩ SDIO/DCS 350Ω RBIAS AND VCM 375Ω 08121-041 08121-044 30kΩ Figure 34. Equivalent RBIAS and VCM Circuit Figure 31. Equivalent SDIO/DCS Input Circuit Rev. A | Page 17 of 36 AD9231 DRVDD AVDD AVDD 30kΩ 375Ω VREF 08121-045 7.5kΩ Figure 35. Equivalent CSB Input Circuit Figure 37. Equivalent VREF Circuit AVDD 375Ω 08121-046 SENSE Figure 36. Equivalent SENSE Circuit Rev. A | Page 18 of 36 08121-047 350Ω CSB AD9231 THEORY OF OPERATION ANALOG INPUT CONSIDERATIONS The analog input to the AD9231 is a differential switchedcapacitor circuit designed for processing differential input signals. This circuit can support a wide common-mode range while maintaining excellent performance. By using an input common-mode voltage of midsupply, users can minimize signal-dependent errors and achieve optimum performance. H In nondiversity applications, the AD9231 can be used as a baseband or direct downconversion receiver, where one ADC is used for I input data and the other is used for Q input data. CPAR H VIN+x CSAMPLE Synchronization capability is provided to allow synchronized timing between multiple channels or multiple devices. S S S S CSAMPLE VIN–x Programming and control of the AD9231 is accomplished using a 3-bit SPI-compatible serial interface. H CPAR H ADC ARCHITECTURE The AD9231 architecture consists of a multistage, pipelined ADC. Each stage provides sufficient overlap to correct for flash errors in the preceding stage. The quantized outputs from each stage are combined into a final 12-bit result in the digital correction logic. The pipelined architecture permits the first stage to operate with a new input sample while the remaining stages operate with preceding samples. Sampling occurs on the rising edge of the clock. Each stage of the pipeline, excluding the last, consists of a low resolution flash ADC connected to a switched-capacitor DAC and an interstage residue amplifier (for example, a multiplying digital-to-analog converter (MDAC)). The residue amplifier magnifies the difference between the reconstructed DAC output and the flash input for the next stage in the pipeline. One bit of redundancy is used in each stage to facilitate digital correction of flash errors. The last stage simply consists of a flash ADC. The output staging block aligns the data, corrects errors, and passes the data to the CMOS output buffers. The output buffers are powered from a separate (DRVDD) supply, allowing adjustment of the output voltage swing. During power-down, the output buffers go into a high impedance state. 08121-006 The AD9231 dual ADC design can be used for diversity reception of signals, where the ADCs are operating identically on the same carrier but from two separate antennae. The ADCs can also be operated with independent analog inputs. The user can sample any fS/2 frequency segment from dc to 200 MHz, using appropriate low-pass or band-pass filtering at the ADC inputs with little loss in ADC performance. Operation to 300 MHz analog input is permitted but occurs at the expense of increased ADC noise and distortion. Figure 38. Switched-Capacitor Input Circuit The clock signal alternately switches the input circuit between sample-and-hold mode (see Figure 38). When the input circuit is switched to sample mode, the signal source must be capable of charging the sample capacitors and settling within one-half of a clock cycle. A small resistor in series with each input can help reduce the peak transient current injected from the output stage of the driving source. In addition, low Q inductors or ferrite beads can be placed on each leg of the input to reduce high differential capacitance at the analog inputs and, therefore, achieve the maximum bandwidth of the ADC. Such use of low Q inductors or ferrite beads is required when driving the converter front end at high IF frequencies. Either a shunt capacitor or two single-ended capacitors can be placed on the inputs to provide a matching passive network. This ultimately creates a low-pass filter at the input to limit unwanted broadband noise. See the AN-742 Application Note, the AN-827 Application Note, and the Analog Dialogue article “Transformer-Coupled Front-End for Wideband A/D Converters” (Volume 39, April 2005) for more information. In general, the precise values depend on the application. Rev. A | Page 19 of 36 AD9231 The analog inputs of the AD9231 are not internally dc-biased. Therefore, in ac-coupled applications, the user must provide a dc bias externally. Setting the device so that VCM = AVDD/2 is recommended for optimum performance, but the device can function over a wider range with reasonable performance, as shown in Figure 39 and Figure 40. The output common-mode voltage of the ADA4938-2 is easily set with the VCM pin of the AD9231 (see Figure 41), and the driver can be configured in a Sallen-Key filter topology to provide band limiting of the input signal. 200Ω VIN 33Ω VIN–x AVDD 90Ω 10pF ADA4938 0.1µF An on-board, common-mode voltage reference is included in the design and is available from the VCM pin. The VCM pin must be decoupled to ground by a 0.1 μF capacitor, as described in the Applications Information section. 120Ω 33Ω ADC VCM VIN+x 200Ω Figure 41. Differential Input Configuration Using the ADA4938-2 For baseband applications below ~10 MHz where SNR is a key parameter, differential transformer-coupling is the recommended input configuration. An example is shown in Figure 42. To bias the analog input, the VCM voltage can be connected to the center tap of the secondary winding of the transformer. 100 SFDR (dBc) 90 80 VIN+x R SNR (dBFS) 2V p-p 49.9Ω C 70 R 60 ADC VIN–x VCM 08121-008 SNR/SFDR (dBFS/dBc) 76.8Ω 08121-007 Input Common Mode 0.1µF 0.6 0.7 0.8 0.9 1.0 1.1 INPUT COMMON-MODE VOLTAGE (V) 1.2 1.3 08121-149 Figure 42. Differential Transformer-Coupled Configuration 50 0.5 The signal characteristics must be considered when selecting a transformer. Most RF transformers saturate at frequencies below a few megahertz (MHz). Excessive signal power can also cause core saturation, which leads to distortion. Figure 39. SNR/SFDR vs. Input Common-Mode Voltage, fIN = 32.1 MHz, fS = 80 MSPS 100 At input frequencies in the second Nyquist zone and above, the noise performance of most amplifiers is not adequate to achieve the true SNR performance of the AD9231. For applications above ~10 MHz where SNR is a key parameter, differential double balun coupling is the recommended input configuration (see Figure 44). SFDR (dBc) SNR/SFDR (dBFS/dBc) 90 80 An alternative to using a transformer-coupled input at frequencies in the second Nyquist zone is to use the AD8352 differential driver. An example is shown in Figure 45. See the AD8352 data sheet for more information. SNR (dBFS) 70 60 0.6 0.7 0.8 0.9 1.0 1.1 INPUT COMMON-MODE VOLTAGE (V) 1.2 1.3 In any configuration, the value of Shunt Capacitor C is dependent on the input frequency and source impedance and may need to be reduced or removed. Table 9 displays the suggested values to set the RC network. However, these values are dependent on the input signal and should be used only as a starting guide. 08121-150 50 0.5 Figure 40. SNR/SFDR vs. Input Common-Mode Voltage, fIN = 10.3 MHz, fS = 20 MSPS Differential Input Configurations Table 9. Example RC Network Optimum performance is achieved while driving the AD9231 in a differential input configuration. For baseband applications, the AD8138, ADA4937-2, and ADA4938-2 differential drivers provide excellent performance and a flexible interface to the ADC. Frequency Range (MHz) 0 to 70 70 to 200 Rev. A | Page 20 of 36 R Series (Ω Each) 33 125 C Differential (pF) 22 Open AD9231 10µF AVDD 1kΩ A single-ended input can provide adequate performance in cost-sensitive applications. In this configuration, SFDR and distortion performance degrade due to the large input commonmode swing. If the source impedances on each input are matched, there should be little effect on SNR performance. Figure 43 shows a typical single-ended input configuration. 1V p-p 0.1µF 49.9Ω R AVDD 0.1µF ADC C 1kΩ 10µF VIN+x 1kΩ R VIN–x 1kΩ Figure 43. Single-Ended Input Configuration 0.1µF 0.1µF R VIN+x 2V p-p 25Ω S S P 0.1µF 25Ω ADC C 0.1µF R VIN–x VCM 08121-010 PA Figure 44. Differential Double Balun Input Configuration VCC ANALOG INPUT 0Ω 16 1 8, 13 11 2 CD RD RG 3 5 0.1µF 0Ω R VIN+x 200Ω 10 ADC C AD8352 4 ANALOG INPUT 0.1µF 0.1µF 0.1µF 200Ω R 14 0.1µF 0.1µF Figure 45. Differential Input Configuration Using the AD8352 Rev. A | Page 21 of 36 VIN–x VCM 08121-011 0.1µF 08121-009 Single-Ended Input Configuration AD9231 Internal Reference Connection A comparator within the AD9231 detects the potential at the SENSE pin and configures the reference into two possible modes, which are summarized in Table 10. If SENSE is grounded, the reference amplifier switch is connected to the internal resistor divider (see Figure 46), setting VREF to 1.0 V. 0 VIN+A/VIN+B –0.5 –1.0 INTERNAL VREF = 0.993V –1.5 –2.0 –2.5 –3.0 VIN–A/VIN–B 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LOAD CURRENT (mA) Figure 47. VREF Accuracy vs. Load Current ADC CORE VREF 1.0µF 0.1µF SELECT LOGIC SENSE ADC 08121-012 0.5V Figure 46. Internal Reference Configuration Table 10. Reference Configuration Summary Selected Mode Fixed Internal Reference Fixed External Reference SENSE Voltage (V) AGND to 0.2 AVDD Resulting VREF (V) 1.0 internal 1.0 applied to external VREF pin Rev. A | Page 22 of 36 Resulting Differential Span (V p-p) 2.0 2.0 08121-014 A stable and accurate 1.0 V voltage reference is built into the AD9231. The VREF can be configured using either the internal 1.0 V reference or an externally applied 1.0 V reference voltage. The various reference modes are summarized in the sections that follow. The Reference Decoupling section describes the best practices PCB layout of the reference. If the internal reference of the AD9231 is used to drive multiple converters to improve gain matching, the loading of the reference by the other converters must be considered. Figure 47 shows how the internal reference voltage is affected by loading. REFERENCE VOLTAGE ERROR (%) VOLTAGE REFERENCE AD9231 External Reference Operation Clock Input Options The use of an external reference may be necessary to enhance the gain accuracy of the ADC or improve thermal drift characteristics. Figure 48 shows the typical drift characteristics of the internal reference in 1.0 V mode. The AD9231 has a very flexible clock input structure. The clock input can be a CMOS, LVDS, LVPECL, or sine wave signal. Regardless of the type of signal being used, clock source jitter is of the most concern, as described in the Jitter Considerations section. 4 Figure 50 and Figure 51 show two preferred methods for clocking the AD9231 (at clock rates up to 625 MHz). A low jitter clock source is converted from a single-ended signal to a differential signal using either an RF transformer or an RF balun. 3 VREF ERROR (mV) 1 0 The RF balun configuration is recommended for clock frequencies between 125 MHz and 625 MHz, and the RF transformer is recommended for clock frequencies from 10 MHz to 200 MHz. The back-to-back Schottky diodes across the transformer/balun secondary limit clock excursions into the AD9231 to approximately 0.8 V p-p differential. –2 –3 –4 –6 –40 –20 0 20 40 TEMPERATURE (°C) 60 80 08121-052 –5 Figure 48. Typical VREF Drift When the SENSE pin is tied to AVDD, the internal reference is disabled, allowing the use of an external reference. An internal reference buffer loads the external reference with an equivalent 7.5 kΩ load (see Figure 37). The internal buffer generates the positive and negative full-scale references for the ADC core. Therefore, the external reference must be limited to a maximum of 1.0 V. This limit helps prevent the large voltage swings of the clock from feeding through to other portions of the AD9231 while preserving the fast rise and fall times of the signal that are critical to a low jitter performance. Mini-Circuits® ADT1-1WT, 1:1 Z 0.1µF CLK+ 100Ω 50Ω ADC 0.1µF SCHOTTKY DIODES: HSMS2822 0.1µF Figure 50. Transformer-Coupled Differential Clock (Up to 200 MHz) 1nF CLOCK INPUT AVDD 0.1µF CLK+ 50Ω ADC 0.1µF 1nF CLK– SCHOTTKY DIODES: HSMS2822 0.9V CLK+ Figure 51. Balun-Coupled Differential Clock (Up to 625 MHz) CLK– 2pF 08121-016 2pF XFMR CLK– CLOCK INPUT CONSIDERATIONS For optimum performance, clock the AD9231 sample clock inputs, CLK+ and CLK−, with a differential signal. The signal is typically ac-coupled into the CLK+ and CLK− pins via a transformer or capacitors. These pins are biased internally (see Figure 49) and require no external bias. 0.1µF CLOCK INPUT 08121-017 –1 Figure 49. Equivalent Clock Input Circuit Rev. A | Page 23 of 36 08121-018 VREF ERROR (mV) 2 AD9231 If a low jitter clock source is not available, another option is to ac couple a differential PECL signal to the sample clock input pins, as shown in Figure 52. The AD9510/AD9511/AD9512/ AD9513/AD9514/AD9515/AD9516/AD9517 clock drivers offer excellent jitter performance. CLK+ 0.1µF 50kΩ AD951x PECL DRIVER 240Ω 50kΩ 100Ω 0.1µF ADC CLK– 08121-019 CLOCK INPUT 0.1µF 240Ω Figure 52. Differential PECL Sample Clock (Up to 625 MHz) A third option is to ac couple a differential LVDS signal to the sample clock input pins, as shown in Figure 53. The AD9510/ AD9511/AD9512/AD9513/AD9514/AD9515/AD9516/AD9517 clock drivers offer excellent jitter performance. 0.1µF CLOCK INPUT CLK+ 0.1µF 50kΩ AD951x LVDS DRIVER 100Ω 0.1µF ADC CLK– 08121-020 CLOCK INPUT 0.1µF 50kΩ Figure 53. Differential LVDS Sample Clock (Up to 625 MHz) In some applications, it may be acceptable to drive the sample clock inputs with a single-ended 1.8 V CMOS signal. In such applications, drive the CLK+ pin directly from a CMOS gate, and bypass the CLK− pin to ground with a 0.1 μF capacitor (see Figure 54). CLOCK INPUT 50Ω 1 1kΩ AD951x CMOS DRIVER OPTIONAL 0.1µF 100Ω 1kΩ The AD9231 clock divider can be synchronized using the external SYNC input. Bit 1 and Bit 2 of Register 0x100 allow the clock divider to be resynchronized on every SYNC signal or only on the first SYNC signal after the register is written. A valid SYNC causes the clock divider to reset to its initial state. This synchronization feature allows multiple parts to have their clock dividers aligned to guarantee simultaneous input sampling. Clock Duty Cycle Typical high speed ADCs use both clock edges to generate a variety of internal timing signals and, as a result, may be sensitive to clock duty cycle. Commonly, a ±5% tolerance is required on the clock duty cycle to maintain dynamic performance characteristics. The AD9231 contains a duty cycle stabilizer (DCS) that retimes the nonsampling (falling) edge, providing an internal clock signal with a nominal 50% duty cycle. This allows the user to provide a wide range of clock input duty cycles without affecting the performance of the AD9231. Noise and distortion performance are nearly flat for a wide range of duty cycles with the DCS on, as shown in Figure 55. Jitter in the rising edge of the input is still of concern and is not easily reduced by the internal stabilization circuit. The duty cycle control loop does not function for clock rates less than 20 MHz nominally. The loop has a time constant associated with it that must be considered in applications in which the clock rate can change dynamically. A wait time of 1.5 μs to 5 μs is required after a dynamic clock frequency increase or decrease before the DCS loop is relocked to the input signal. VCC 0.1µF The AD9231 contains an input clock divider with the ability to divide the input clock by integer values between 1 and 8. Optimum performance is obtained by enabling the internal duty cycle stabilizer (DCS) when using divide ratios other than 1, 2, or 4. CLK+ ADC CLK– 80 RESISTOR IS OPTIONAL. 75 DCS ON 70 Figure 54. Single-Ended 1.8 V CMOS Input Clock (Up to 200 MHz) SNR (dBFS) 150Ω 08121-021 0.1µF 65 60 DCS OFF 55 50 45 08121-078 0.1µF CLOCK INPUT Input Clock Divider 40 10 20 30 40 50 60 POSITIVE DUTY CYCLE (%) Figure 55. SNR vs. DCS On/Off Rev. A | Page 24 of 36 70 80 AD9231 Jitter Considerations POWER DISSIPATION AND STANDBY MODE High speed, high resolution ADCs are sensitive to the quality of the clock input. The degradation in SNR from the low frequency SNR (SNRLF) at a given input frequency (fINPUT) due to jitter (tJRMS) can be calculated by As shown in Figure 57, the analog core power dissipated by the AD9231 is proportional to its sample rate. The digital power dissipation of the CMOS outputs are determined primarily by the strength of the digital drivers and the load on each output bit. SNRHF = −10 log[(2π × fINPUT × tJRMS)2 + 10 ( − SNRLF /10) ] In the previous equation, the rms aperture jitter represents the clock input jitter specification. IF undersampling applications are particularly sensitive to jitter, as illustrated in Figure 56. 80 75 70 0.5ps 1.5ps 50 45 3.0ps 1 10 2.0ps 2.5ps 100 FREQUENCY (MHz) where N is the number of output bits (26, in the case of the AD9231). Reducing the capacitive load presented to the output drivers can minimize digital power consumption. The data in Figure 57 was taken using the same operating conditions as those used for the Typical Performance Characteristics, with a 5 pF load on each output driver. 1.0ps 1k 08121-022 SNR (dBFS) 0.2ps 65 55 IDRVDD = VDRVDD × CLOAD × fCLK × N This maximum current occurs when every output bit switches on every clock cycle, that is, a full-scale square wave at the Nyquist frequency of fCLK/2. In practice, the DRVDD current is established by the average number of output bits switching, which is determined by the sample rate and the characteristics of the analog input signal. 0.05ps 60 The maximum DRVDD current (IDRVDD) can be calculated as 150 For more information, see the AN-501 Application Note and the AN-756 Application Note available on www.analog.com. CHANNEL/CHIP SYNCHRONIZATION 130 AD9231-80 110 AD9231-65 90 AD9231-40 70 08121-079 The clock input should be treated as an analog signal in cases in which aperture jitter may affect the dynamic range of the AD9231. To avoid modulating the clock signal with digital noise, keep power supplies for clock drivers separate from the ADC output driver supplies. Low jitter, crystal-controlled oscillators make the best clock sources. If the clock is generated from another type of source (by gating, dividing, or another method), it should be retimed by the original clock at the last step. ANALOG CORE POWER (mW) Figure 56. SNR vs. Input Frequency and Jitter AD9231-20 50 0 10 20 30 40 50 CLOCK RATE (MSPS) 60 Figure 57. Analog Core Power vs. Clock Rate The AD9231 has a SYNC input that offers the user flexible synchronization options for synchronizing sample clocks across multiple ADCs. The input clock divider can be enabled to synchronize on a single occurrence of the SYNC signal or on every occurrence. The SYNC input is internally synchronized to the sample clock; however, to ensure there is no timing uncertainty between multiple parts, the SYNC input signal should be externally synchronized to the input clock signal, meeting the setup and hold times shown in Table 5. Drive the SYNC input using a single-ended CMOS-type signal. Rev. A | Page 25 of 36 70 80 AD9231 The AD9231 is placed in power-down mode either by the SPI port or by asserting the PDWN pin high. In this state, the ADC typically dissipates 2.2 mW. During power-down, the output drivers are placed in a high impedance state. Asserting the PDWN pin low returns the AD9231 to its normal operating mode. Note that PDWN is referenced to the digital output driver supply (DRVDD) and should not exceed that supply voltage. As detailed in the AN-877 Application Note, Interfacing to High Speed ADCs via SPI, the data format can be selected for offset binary, twos complement, or gray code when using the SPI control. Low power dissipation in power-down mode is achieved by shutting down the reference, reference buffer, biasing networks, and clock. Internal capacitors are discharged when entering powerdown mode and then must be recharged when returning to normal operation. As a result, wake-up time is related to the time spent in power-down mode, and shorter power-down cycles result in proportionally shorter wake-up times. Digital Output Enable Function (OEB) When using the SPI port interface, the user can place the ADC in power-down mode or standby mode. Standby mode allows the user to keep the internal reference circuitry powered when faster wake-up times are required. See the Memory Map section for more details. DIGITAL OUTPUTS The AD9231 output drivers can be configured to interface with 1.8 V to 3.3 V CMOS logic families. Output data can also be multiplexed onto a single output bus to reduce the total number of traces required. The CMOS output drivers are sized to provide sufficient output current to drive a wide variety of logic families. However, large drive currents tend to cause current glitches on the supplies and may affect converter performance. Applications requiring the ADC to drive large capacitive loads or large fanouts may require external buffers or latches. The output data format can be selected to be either offset binary or twos complement by setting the SCLK/DFS pin when operating in the external pin mode (see Table 11). Table 11. SCLK/DFS Mode Selection (External Pin Mode) Voltage at Pin AGND DRVDD SCLK/DFS Offset binary (default) Twos complement SDIO/DCS DCS disabled(default) DCS enabled The AD9231 has a flexible three-state ability for the digital output pins. The three-state mode is enabled using the OEB pin or through the SPI interface. If the OEB pin is low, the output data drivers and DCOs are enabled. If the OEB pin is high, the output data drivers and DCOs are placed in a high impedance state. This OEB function is not intended for rapid access to the data bus. Note that OEB is referenced to the digital output driver supply (DRVDD) and should not exceed that supply voltage. When using the SPI interface, the data outputs and DCO of each channel can be independently three-stated by using the output disable (OEB) bit (Bit 4) in Register 0x14. TIMING The AD9231 provides latched data with a pipeline delay of 9 clock cycles. Data outputs are available one propagation delay (tPD) after the rising edge of the clock signal. Minimize the length of the output data lines and loads placed on them to reduce transients within the AD9231. These transients can degrade converter dynamic performance. The lowest typical conversion rate of the AD9231 is 3 MSPS. At clock rates below 3 MSPS, dynamic performance can degrade. Data Clock Output (DCO) The AD9231 provides two data clock output (DCO) signals intended for capturing the data in an external register. The CMOS data outputs are valid on the rising edge of DCO, unless the DCO clock polarity has been changed via the SPI. See Figure 2 and Figure 3 for a graphical timing description. Table 12. Output Data Format Input (V) VIN+ − VIN− VIN+ − VIN− VIN+ − VIN− VIN+ − VIN− VIN+ − VIN− Condition (V) < −VREF − 0.5 LSB = −VREF =0 = +VREF − 1.0 LSB > +VREF − 0.5 LSB Offset Binary Output Mode 0000 0000 0000 0000 0000 0000 1000 0000 0000 1111 1111 1111 1111 1111 1111 Rev. A | Page 26 of 36 Twos Complement Mode 1000 0000 0000 1000 0000 0000 0000 0000 0000 0111 1111 1111 0111 1111 1111 OR 1 0 0 0 1 AD9231 BUILT-IN SELF-TEST (BIST) AND OUTPUT TEST The AD9231 includes a built-in self-test feature designed to enable verification of the integrity of each channel as well as to facilitate board level debugging. A built-in self-test (BIST) feature that verifies the integrity of the digital datapath of the AD9231 is included. Various output test options are also provided to place predictable values on the outputs of the AD9231. generator, Bit 2 (BIST INIT) of Register 0x0E. At the completion of the BIST, Bit 0 of Register 0x24 is automatically cleared. The PN sequence can be continued from its last value by writing a 0 in Bit 2 of Register 0x0E. However, if the PN sequence is not reset, the signature calculation does not equal the predetermined value at the end of the test. At that point, the user needs to rely on verifying the output data. BUILT-IN SELF-TEST (BIST) OUTPUT TEST MODES The BIST is a thorough test of the digital portion of the selected AD9231 signal path. Perform the BIST test after a reset to ensure the part is in a known state. During BIST, data from an internal pseudorandom noise (PN) source is driven through the digital datapath of both channels, starting at the ADC block output. At the datapath output, CRC logic calculates a signature from the data. The BIST sequence runs for 512 cycles and then stops. Once completed, the BIST compares the signature results with a pre-determined value. If the signatures match, the BIST sets Bit 0 of Register 0x24, signifying the test passed. If the BIST test failed, Bit 0 of Register 0x24 is cleared. The outputs are connected during this test, so the PN sequence can be observed as it runs. Writing the value 0x05 to Register 0x0E runs the BIST. This enables the Bit 0 (BIST enable) of Register 0x0E and resets the PN sequence The output test options are described in Table 16 at Address 0x0D. When an output test mode is enabled, the analog section of the ADC is disconnected from the digital back-end blocks and the test pattern is run through the output formatting block. Some of the test patterns are subject to output formatting, and some are not. The PN generators from the PN sequence tests can be reset by setting Bit 4 or Bit 5 of Register 0x0D. These tests can be performed with or without an analog signal (if present, the analog signal is ignored), but they do require an encode clock. For more information, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. Rev. A | Page 27 of 36 AD9231 SERIAL PORT INTERFACE (SPI) The AD9231 serial port interface (SPI) allows the user to configure the converter for specific functions or operations through a structured register space provided inside the ADC. The SPI gives the user added flexibility and customization, depending on the application. Addresses are accessed via the serial port and can be written to or read from via the port. Memory is organized into bytes that can be further divided into fields, which are documented in the Memory Map section. For detailed operational information, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. The falling edge of CSB, in conjunction with the rising edge of SCLK, determines the start of the framing. An example of the serial timing and its definitions can be found in Figure 58 and Table 5. CONFIGURATION USING THE SPI During an instruction phase, a 16-bit instruction is transmitted. Data follows the instruction phase, and its length is determined by the W0 and W1 bits as shown in Figure 58. Other modes involving the CSB are available. The CSB can be held low indefinitely, which permanently enables the device; this is called streaming. The CSB can stall high between bytes to allow for additional external timing. When CSB is tied high, SPI functions are placed in high impedance mode. This mode turns on any SPI pin secondary functions. Three pins define the SPI of this ADC: the SCLK, the SDIO, and the CSB (see Table 13). The SCLK (a serial clock) is used to synchronize the read and write data presented from and to the ADC. The SDIO (serial data input/output) is a dual-purpose pin that allows data to be sent and read from the internal ADC memory map registers. The CSB (chip select bar) is an activelow control that enables or disables the read and write cycles. All data is composed of 8-bit words. The first bit of the first byte in a multibyte serial data transfer frame indicates whether a read command or a write command is issued. This allows the serial data input/output (SDIO) pin to change direction from an input to an output at the appropriate point in the serial frame. In addition to word length, the instruction phase determines whether the serial frame is a read or write operation, allowing the serial port to be used both to program the chip and to read the contents of the on-chip memory. If the instruction is a readback operation, performing a readback causes the serial data input/ output (SDIO) pin to change direction from an input to an output at the appropriate point in the serial frame. Table 13. Serial Port Interface Pins Pin SCLK SDIO CSB Function Serial Clock. The serial shift clock input, which is used to synchronize serial interface reads and writes. Serial Data Input/Output. A dual-purpose pin that typically serves as an input or an output, depending on the instruction being sent and the relative position in the timing frame. Chip Select Bar. An active-low control that gates the read and write cycles. tHIGH tDS tS tDH Data can be sent in MSB-first mode or in LSB-first mode. MSB first is the default on power-up and can be changed via the SPI port configuration register. For more information about this and other features, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. tCLK tH tLOW CSB SDIO DON’T CARE DON’T CARE R/W W1 W0 A12 A11 A10 A9 A8 A7 D5 Figure 58. Serial Port Interface Timing Diagram Rev. A | Page 28 of 36 D4 D3 D2 D1 D0 DON’T CARE 08121-023 SCLK DON’T CARE AD9231 HARDWARE INTERFACE Table 14. Mode Selection The pins described in Table 13 constitute the physical interface between the programming device of the user and the serial port of the AD9231. The SCLK pin and the CSB pin function as inputs when using the SPI interface. The SDIO pin is bidirectional, functioning as an input during write phases and as an output during readback. Pin SDIO/DCS SCLK/DFS External Voltage DRVDD AGND(default) DRVDD AGND (default) DRVDD AGND (default) DRVDD AGND (default) Configuration Duty cycle stabilizer enabled Duty cycle stabilizer disabled Twos complement enabled Offset binary enabled Outputs in high impedance Outputs enabled Chip in power-down or standby Normal operation The SPI interface is flexible enough to be controlled by either FPGAs or microcontrollers. One method for SPI configuration is described in detail in the AN-812 Application Note, Microcontroller-Based Serial Port Interface (SPI) Boot Circuit. OEB The SPI port should not be active during periods when the full dynamic performance of the converter is required. Because the SCLK signal, the CSB signal, and the SDIO signal are typically asynchronous to the ADC clock, noise from these signals can degrade converter performance. If the on-board SPI bus is used for other devices, it may be necessary to provide buffers between this bus and the AD9231 to prevent these signals from transitioning at the converter inputs during critical sampling periods. Table 15 provides a brief description of the general features that are accessible via the SPI. These features are described in detail in the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. The AD9231 part-specific features are described in detail in Table 16. PDWN SPI ACCESSIBLE FEATURES SDIO/DCS and SCLK/DFS serve a dual function when the SPI interface is not being used. When the pins are strapped to DRVDD or ground during device power-on, they are associated with a specific function. The Digital Outputs section describes the strappable functions supported on the AD9231. Table 15. Features Accessible Using the SPI Feature Mode Clock Offset Test I/O CONFIGURATION WITHOUT THE SPI In applications that do not interface to the SPI control registers, the SDIO/DCS pin, the SCLK/DFS pin, the OEB pin, and the PDWN pin serve as standalone CMOS-compatible control pins. When the device is powered up, it is assumed that the user intends to use the pins as static control lines for the duty cycle stabilizer, output data format, output enable, and powerdown feature control. In this mode, connect the CSB chip select to DRVDD, which disables the serial port interface. Output Mode Output Phase Output Delay Rev. A | Page 29 of 36 Description Allows the user to set either power-down mode or standby mode Allows the user to access the DCS via the SPI Allows the user to digitally adjust the converter offset Allows the user to set test modes to have known data on output bits Allows the user to set up outputs Allows the user to set the output clock polarity Allows the user to vary the DCO delay AD9231 MEMORY MAP READING THE MEMORY MAP REGISTER TABLE Logic Levels Each row in the memory map register table (see Table 16) has eight bit locations. The memory map is roughly divided into four sections: the chip configuration registers (Address 0x00 to Address 0x02); the device index and transfer registers (Address 0x05 and Address 0xFF); the program registers, including setup, control, and test (Address 0x08 to Address 0x2E); and the digital feature control registers (Address 0x100 and Address 0x101). An explanation of logic level terminology follows: Table 16 documents the default hexadecimal value for each hexadecimal address shown. The column with the heading Bit 7 (MSB) is the start of the default hexadecimal value given. For example, Address 0x05, the channel index register, has a hexadecimal default value of 0x03. This means that in Address 0x05 Bit[7:2] = 0, and the remaining Bits[1:0] = 1. This setting is the default channel index setting. The default value results in both ADC channels receiving the next write command. For more information on this function and others, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. This application note details the functions controlled by Register 0x00 to Register 0xFF. The remaining registers, Register 0x100 and Register 0x101, are documented in the Memory Map Register Descriptions section following Table 16. OPEN LOCATIONS All address and bit locations that are not included in the SPI map are not currently supported for this device. Unused bits of a valid address location should be written with 0s. Writing to these locations is required only when part of an address location is open (for example, Address 0x05). If the entire address location is open, it is omitted from the SPI map (for example, Address 0x13) and should not be written. • • “Bit is set” is synonymous with “bit is set to Logic 1” or “writing Logic 1 for the bit.” “Clear a bit” is synonymous with “bit is set to Logic 0” or “writing Logic 0 for the bit.” Transfer Register Map Address 0x08 to Address 0x18 are shadowed. Writes to these addresses do not affect part operation until a transfer command is issued by writing 0x01 to Address 0xFF, setting the transfer bit. This allows these registers to be updated internally and simultaneously when the transfer bit is set. The internal update takes place when the transfer bit is set, and then the bit autoclears. Channel-Specific Registers Some channel setup functions can be programmed differently for each channel. In these cases, channel address locations are internally duplicated for each channel. These registers and bits are designated in the memory map register table as local. These local registers and bits can be accessed by setting the appropriate Channel A (Bit 0) or Channel B (Bit 1) bits in Register 0x05. If both bits are set, the subsequent write affects the registers of both channels. In a read cycle, set only Channel A or Channel B to read one of the two registers. If both bits are set during an SPI read cycle, the part returns the value for Channel A. Registers and bits designated as global in the memory map register table affect the entire part or the channel features for which independent settings are not allowed between channels. The settings in Register 0x05 do not affect the global registers and bits. DEFAULT VALUES After the AD9231 is reset, critical registers are loaded with default values. The default values for the registers are given in the memory map register table (see Table 16). Rev. A | Page 30 of 36 AD9231 MEMORY MAP REGISTER TABLE All address and bit locations that are not included in Table 16 are not currently supported for this device. Table 16. Addr Register (Hex) Name Chip Configuration Registers 0x00 SPI port configuration (global) Bit 7 (MSB) 0x01 Chip ID (global) 8-bit chip ID bits [7:0] AD9231 = 0x24 0x02 Chip grade (global) Open 0 Bit 0 (LSB) Default Value (Hex) Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 LSB first Soft reset 1 1 Soft reset LSB first 0 0x18 The nibbles are mirrored so that LSB- or MSB-first mode registers correctly, regardless of shift mode Unique chip ID used to differentiate devices; read only Unique speed grade ID used to differentiate devices; read only Bits are set to determine which device on chip receives the next write command; the default is all devices on chip Synchronously transfers data from the master shift register to the slave Open Speed grade ID [6:4] 20 MSPS = 000 40 MSPS = 001 65 MSPS = 010 80 MSPS = 011 Device Index and Transfer Registers 0x05 Channel index Open Open Open Open Open Open ADC B default ADC A default 0x03 0xFF Open Open Open Open Open Open Transfer 0x00 Open Open 00 = chip run 01 = full powerdown 10 = standby 11 = chip wide digital reset (local) Open Duty cycle stabilize Clock divider [2:0] Clock divide ratio 000 = divide by 1 001 = divide by 2 010 = divide by 3 011 = divide by 4 100 = divide by 5 101 = divide by 6 110 = divide by 7 111 = divide by 8 Output test mode [3:0] (local) 0000 = off (default) 0001 = midscale short 0010 = positive FS 0011 = negative FS 0100 = alternating checkerboard 0101 = PN 23 sequence 0110 = PN 9 sequence 0111 = one/zero word toggle 1000 = user input 1001 = 1-/0-bit toggle 1010 = 1x sync 1011 = one bit high 1100 = mixed bit frequency 0x80 Transfer Open Comments Bit 6 Program Registers (May or May Not Be Indexed by Device Index) External pin function 0x08 Modes External 0x00 full power-down power0x01 standby down (local) enable (local) 0x09 Clock (global) Open Open 0x0B Clock divide (global) Open 0x0D Test mode (local) User test mode (local) 00 = single 01 = alternate 10 = single once 11 = alternate once Open Reset PN long gen Open Reset PN short gen Rev. A | Page 31 of 36 Determines various generic modes of chip operation 0x00 0x00 The divide ratio is the value plus 1 0x00 When set, the test data is placed on the output pins in place of normal data AD9231 Addr (Hex) 0x0E Register Name BIST enable Bit 7 (MSB) Open 0x10 0x14 Offset adjust (local) Output mode 8-bit device offset adjustment [7:0] (local) Offset adjust in LSBs from +127 to −128 (twos complement format) 00 = 3.3 V CMOS Output mux Output Open Output 10 = 1.8 V CMOS enable disable invert (interleaved) (local) (local) 0x15 OUTPUT_ADJUST 0x16 OUTPUT_PHASE 3.3 V DCO drive strength 00 = 1 stripe (default) 01 = 2 stripes 10 = 3 stripes 11 = 4 stripes Open DCO output polarity 0= normal 1= inverted (local) Open Open 0x17 OUTPUT_DELAY Enable DCO delay Open Enable data delay Open Bit 6 Open Bit 5 Open Bit 4 Open 1.8 V DCO drive strength 00 = 1 stripe 01 = 2 stripes 10 = 3 stripes (default) 11 = 4 stripes Bit 3 Open Bit 2 BIST INIT Bit 1 Open Bit 0 (LSB) BIST enable Default Value (Hex) 0x00 0x00 00 = offset binary 01 = twos complement 10 = gray code 11 = offset binary (local) 1.8 V data 3.3 V data drive strength drive strength 00 = 1 stripe 00 = 1 stripe 01 = 2 stripes (default) 10 = 3 stripes 01 = 2 stripes (default) 10 = 3 stripes 11 = 4 stripes 11 = 4 stripes Open Input clock phase adjust [2:0] (Value is number of input clock cycles of phase delay) 000 = no delay 001 = 1 input clock cycle 010 = 2 input clock cycles 011 = 3 input clock cycles 100 = 4 input clock cycles 101 = 5 input clock cycles 110 = 6 input clock cycles 111 = 7 input clock cycles Open DCO/Data delay[2:0] 0x00 Configures the outputs and the format of the data 0x22 Determines CMOS output drive strength properties 0x00 On devices that utilize global clock divide, this register determines which phase of the divider output is used to supply the output clock; internal latching is unaffected 0x00 This sets the fine output delay of the output clock but does not change internal timing User-defined pattern, 1 LSB User-defined pattern, 1 MSB User-defined pattern, 2 LSB User-defined pattern, 2 MSB Least significant byte of BIST signature, read only Disable the OR pin for the indexed channel Assign an ADC to an output channel 000 = 0.56 ns 001 = 1.12 ns 010 = 1.68 ns 011 = 2.24 ns 100 = 2.80 ns 101 = 3.36 ns 110 = 3.92 ns 111 = 4.48 ns 0x19 USER_PATT1_LSB B7 B6 B5 B4 B3 B2 B1 B0 0x00 0x1A USER_PATT1_MSB B15 B14 B13 B12 B11 B10 B9 B8 0x00 0x1B USER_PATT2_LSB B7 B6 B5 B4 B3 B2 B1 B0 0x00 0x1C USER_PATT2_MSB B15 B14 B13 B12 B11 B10 B9 B8 0x00 0x24 BIST signature LSB 0x2A Features Open Open Open Open Open Open Open OR OE (local) 0x01 0x2E Output assign Open Open Open Open Open Open Open 0= ADC A 1= ADC B (local) Ch A = 0x00 Ch B = 0x01 BIST signature [7:0] Rev. A | Page 32 of 36 Comments When Bit 0 is set, the BIST function is initiated Device offset trim 0x00 AD9231 Default Value (Hex) Addr Register (Hex) Name Digital Feature Control 0x100 Sync control (global) Bit 7 (MSB) Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Open Open Open Open Open Clock divider sync enable Master sync enable 0x01 0x101 Enable OEB Pin 47 (local) Open Open Open Enable GCLK detect Clock divider next sync only Run GCLK Open Disable SDIO pulldown 0x88 USR2 Bit 0 (LSB) MEMORY MAP REGISTER DESCRIPTIONS USR2 (Register 0x101) For additional information about functions controlled in Register 0x00 to Register 0xFF, see the AN-877 Application Note, Interfacing to High Speed ADCs via SPI. Bit 7—Enable OEB Pin 47 Sync Control (Register 0x100) Bits[7:3]—Reserved Bit 3—Enable GCLK Detect Comments Enables internal oscillator for clock rates < 5 MHz Normally set high, this bit allows Pin 47 to function as the output enable. If it is set low, it disables Pin 47. Bit 2—Clock Divider Next Sync Only If the master sync enable bit (Address 0x100, Bit 0) and the clock divider sync enable bit (Address 0x100, Bit 1) are high, Bit 2 allows the clock divider to sync to the first sync pulse it receives and to ignore the rest. The clock divider sync enable bit (Address 0x100, Bit 1) resets after it syncs. Bit 1—Clock Divider Sync Enable Bit 1 gates the sync pulse to the clock divider. The sync signal is enabled when Bit 1 and Bit 0 are high and the device is operating in continuous sync mode as long as Bit 2 of the sync control is low. Normally set high, this bit enables a circuit that detects encode rates below about 5 MSPS. When a low encode rate is detected an internal oscillator, GCLK, is enabled ensuring the proper operation of several circuits. If set low the detector is disabled. Bit 2—Run GCLK This bit enables the GCLK oscillator. For some applications with encode rates below 10 MSPS, it may be preferable to set this bit high to supersede the GCLK detector. Bit 0—Disable SDIO Pull-Down This bit can be set high to disable the internal 30 kΩ pull-down on the SDIO pin, which can be used to limit the loading when many devices are connected to the SPI bus. Bit 0—Master Sync Enable Bit 0 must be high to enable any of the sync functions. Rev. A | Page 33 of 36 AD9231 APPLICATIONS INFORMATION DESIGN GUIDELINES Before starting design and layout of the AD9231 as a system, it is recommended that the designer become familiar with these guidelines, which discuss the special circuit connections and layout requirements needed for certain pins. Power and Ground Recommendations When connecting power to the AD9231, it is strongly recommended that two separate supplies be used. Use one 1.8 V supply for analog (AVDD); use a separate 1.8 V to 3.3 V supply for the digital output supply (DRVDD). If a common 1.8 V AVDD and DRVDD supply must be used, the AVDD and DRVDD domains must be isolated with a ferrite bead or filter choke and separate decoupling capacitors. Several different decoupling capacitors can be used to cover both high and low frequencies. Locate these capacitors close to the point of entry at the PCB level and close to the pins of the part, with minimal trace length. A single PCB ground plane should be sufficient when using the AD9231. With proper decoupling and smart partitioning of the PCB analog, digital, and clock sections, optimum performance is easily achieved. Exposed Paddle Thermal Heat Sink Recommendations The exposed paddle (Pin 0) is the only ground connection for the AD9231; therefore, it must be connected to analog ground (AGND) on the customer’s PCB. To achieve the best electrical and thermal performance, mate an exposed (no solder mask) continuous copper plane on the PCB to the AD9231 exposed paddle, Pin 0. The copper plane should have several vias to achieve the lowest possible resistive thermal path for heat dissipation to flow through the bottom of the PCB. Fill or plug these vias with nonconductive epoxy. To maximize the coverage and adhesion between the ADC and the PCB, a silkscreen should be overlaid to partition the continuous plane on the PCB into several uniform sections. This provides several tie points between the ADC and the PCB during the reflow process. Using one continuous plane with no partitions guarantees only one tie point between the ADC and the PCB. For detailed information about packaging and PCB layout of chip scale packages, see the AN-772 Application Note, A Design and Manufacturing Guide for the Lead Frame Chip Scale Package (LFCSP), at www.analog.com. VCM The VCM pin should be decoupled to ground with a 0.1 μF capacitor, as shown in Figure 42. RBIAS The AD9231 requires that a 10 kΩ resistor be placed between the RBIAS pin and ground. This resistor sets the master current reference of the ADC core and should have at least a 1% tolerance. Reference Decoupling Externally decoupled the VREF pin to ground with a low ESR, 1.0 μF capacitor in parallel with a low ESR, 0.1 μF ceramic capacitor. SPI Port The SPI port should not be active during periods when the full dynamic performance of the converter is required. Because the SCLK, CSB, and SDIO signals are typically asynchronous to the ADC clock, noise from these signals can degrade converter performance. If the on-board SPI bus is used for other devices, it may be necessary to provide buffers between this bus and the AD9231 to keep these signals from transitioning at the converter inputs during critical sampling periods. Rev. A | Page 34 of 36 AD9231 OUTLINE DIMENSIONS 0.60 MAX 9.00 BSC SQ 0.60 MAX 48 64 49 1 PIN 1 INDICATOR PIN 1 INDICATOR 0.50 BSC 0.50 0.40 0.30 33 32 16 17 0.05 MAX 0.02 NOM SEATING PLANE 0.30 0.23 0.18 0.25 MIN 7.50 REF 0.80 MAX 0.65 TYP 12° MAX 1.00 0.85 0.80 6.35 6.20 SQ 6.05 EXPOSED PAD (BOTTOM VIEW) 0.20 REF FOR PROPER CONNECTION OF THE EXPOSED PAD, REFER TO THE PIN CONFIGURATION AND FUNCTION DESCRIPTIONS SECTION OF THIS DATA SHEET. COMPLIANT TO JEDEC STANDARDS MO-220-VMMD-4 091707-C 8.75 BSC SQ TOP VIEW Figure 59. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 9 mm × 9 mm Body, Very Thin Quad (CP-64-4) Dimensions shown in millimeters ORDERING GUIDE Model 1 AD9231BCPZ-80 AD9231BCPZRL7-80 AD9231BCPZ-65 AD9231BCPZRL7-65 AD9231BCPZ-40 AD9231BCPZRL7-40 AD9231BCPZ-20 AD9231BCPZRL7-20 AD9231-80EBZ AD9231-65EBZ AD9231-40EBZ AD9231-20EBZ 1 2 Notes 2 2 2 2 2 2 2 2 Temperature Range −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C Package Description 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) Evaluation Board Evaluation Board Evaluation Board Evaluation Board Z = RoHS Compliant Part. The exposed paddle (Pin 0) is the only GND connection on the chip and must be connected to the PCB AGND. Rev. A | Page 35 of 36 Package Option CP-64-4 CP-64-4 CP-64-4 CP-64-4 CP-64-4 CP-64-4 CP-64-4 CP-64-4 AD9231 NOTES ©2009–2010 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D08121-0-6/10(A) Rev. A | Page 36 of 36