cd00282033

AN3257
Application note
STEVAL-ISA080V1 90 W-HB LLC resonant converter
based on the L6585DE combo IC
Introduction
This application note describes the performance of a 90 W, wide range mains, power factor
corrected AC-DC power supply demonstration board.
The architecture is based on a two-stage approach: a front-end PFC pre-regulator and a
downstream multi-resonant half bridge converter. Both stages are controlled by the new IC L6585DE- which integrates PFC and half bridge control circuits and the relevant drivers.
Although this new device is dedicated to managing electronic ballast, it's possible to use it
also for a HB-LLC resonant converter.
The PFC section achieves current mode control operating in transition mode, offering a
highly linear multiplier including a THD optimizer which allows for an extremely low THD,
even over a large range of input voltages and loading conditions.
The HB controller offers the designer a very precise oscillator, a logic that manages all the
operating steps and a full set of protection features dedicated to lighting applications but
useful also for the resonant converter.
Figure 1.
March 2011
90 W LCC resonant converter driven by L6585DE demonstration board
Doc ID 17803 Rev 1
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www.st.com
Contents
AN3257
Contents
1
Basis of the HB-LLC resonant converter . . . . . . . . . . . . . . . . . . . . . . . . 5
2
Main characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3
SMPS design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
3.1
L6585DE biasing circuitry (pin-by-pin) . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
3.2
PFC power section design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
3.3
LLC resonant circuit design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
4
Experimental results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
5
Conducted emission pre-compliance test . . . . . . . . . . . . . . . . . . . . . . 21
6
BOM list . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
7
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
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AN3257
List of tables
List of tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
Table 8.
Electrical specifications. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Board performance . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Thermal results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Efficiency measurements @ Vin=115 Vac . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Efficiency measurements @ Vin=230 Vac . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
No load consumption . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
BOM list . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
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List of figures
AN3257
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
4/28
90 W LCC resonant converter driven by L6585DE demonstration board. . . . . . . . . . . . . . . 1
LLC resonant half bridge topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Efficiency vs. input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Efficiency vs. Pout . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Resonant circuit primary side waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Resonant circuit secondary side waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Load transition 0 % ==> 100 %. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Load transition 100 % ==> 0 %. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Short-circuit during run mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Startup [email protected] V - full load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
CE at 115 Vac 50 Hz - line 1 peak detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
CE at 115 Vac 50 Hz - line 2 peak detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
CE at 230 Vac 50 Hz - line 1 peak detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
CE at 230 Vac 50 Hz - line 2 peak detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Doc ID 17803 Rev 1
AN3257
1
Basis of the HB-LLC resonant converter
Basis of the HB-LLC resonant converter
The LLC resonant half bridge belongs to the family of multi-resonant converters. Actually, as
the resonant tank includes three reactive elements (Cr, Ls and Lp, shown in Figure 2), there
are two resonant frequencies associated with this circuit. One is related to the condition of
the secondary winding(s) conducting, where the inductance Lp disappears because it is
dynamically shorted out by the low-pass filter and the load (there is a constant Vout voltage
across it):
Equation 1
fR1 =
1
2π L S ⋅ CR
The other resonant frequency is relevant to the condition of the secondary winding(s) open,
where the tank circuit turns from LLC to LC because Ls and Lp can be unified in a single
inductor:
Equation 2
fR2 =
1
2π (L S + LP ) ⋅ CR
It is possible to show that for frequencies f > fR1, the input impedance of the loaded resonant
tank is inductive and that for frequencies f < fR2, the input impedance is capacitive. In the
frequency region fR2 < f < fR1, the impedance can be either inductive or capacitive
depending on the load resistance Rout. The LLC resonant converter is normally operated in
the region where the input impedance of the resonant tank has an inductive nature, i.e. it
increases with frequency. This implies that power flow can be controlled by changing the
operating frequency of the converter in such a way that a reduced power demand from the
load produces a frequency rise, while an increased power demand causes a frequency
reduction.
Figure 2.
LLC resonant half bridge topology
4
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Basis of the HB-LLC resonant converter
AN3257
The most significant advantages of this topology are:
●
Soft-switching of all semiconductor devices: ZVS (zero-voltage switching) at turn on for
the MOSFETs and ZCS (zero-current switching) at both turn on and turn off for the
secondary rectifiers. The first property results from the correct design of the resonant
tank. The second one is a natural feature of the topology.
●
Ability to accommodate an extremely broad load range, including zero load, with an
acceptable frequency variation. Also this property results from the correct design of the
resonant tank.
●
Magnetic integration, which allows the combination of different magnetic devices into a
single physical device.
As a result of all the above factors, high efficiency, high switching frequency capability, and
high power density are typical characteristics of the converters based on this topology.
Operation at resonance is the preferred operating point, where load regulation is ideally
zero, where tank current is maximally sinusoidal, and where peak tank current is minimized
for a given power throughput.
6/28
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AN3257
2
Main characteristics
Main characteristics
The SMPS electrical specifications are shown in Table 1 and the schematic is presented in
Figure 3.
Table 1.
Electrical specifications
Input parameters
VIN
Input voltage range
90 to 265 VRMS
fline
Line frequency
50/60 Hz
Output parameters
VOUT
Output voltage
19 V
IOUT
Output current
4.7 A
Doc ID 17803 Rev 1
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Main characteristics
AN3257
Schematic
!-V
AN3257
3
SMPS design
SMPS design
The design of the major circuit parts is described in this section.
3.1
L6585DE biasing circuitry (pin-by-pin)
●
Pin 1 OSC: is one of the two oscillator inputs. The value of the capacitor connected to
ground defines the half bridge switching frequency in each operating state. C4=470 pF
value was chosen.
●
Pin 2 RF: the component choice with oscillator capacitance defines the half bridge
switching frequency in each operating state. A resistor R11 connected to ground sets
the run frequency that, for resonant converters, represents the minimum frequency. For
a lamp ballast, during the preheating time (startup time) the switching frequency is set
by the parallel of the above resistance with the one R4 connected between pins RF and
EOI; in this case (resonant converter design) it is true until the control loop take over by
means and the optocoupler. Choosing the following frequencies:
Equation 3
frun = fmin = 62.5kHz fpre = fstart −up 70kHz
t ign = 46ms
it is possible calculate R11 by following the formula:
Equation 4
e = 1−
1.33
(C4 )
0.581
k=
499.6 ⋅ 10 3
(C 4 )0.872
1/ e
⎛ k ⎞
⎟
R11 = ⎜⎜
⎟
⎝ frun ⎠
= 43kΩ
and so the R4 value is:
Equation 5
⎛ k
R11 // R 4 = ⎜
⎜ fpre
⎝
●
1/ e
⎞
⎟
⎟
⎠
⇒ R 4 = 330kΩ
Pin 3 EOI: for lamp ballast applications, the net C5-R4 on the EOI pin is needed to set
the frequency change from preheating mode to run mode. In the resonant converter
this change, similar to the one between startup frequency and steady-state, is
influenced at the start by the feedback response. However, the C5-R4 choice was made
as it would be an electronic ballast; as R4 value has already been calculated and tign at
start fixed, C5 value is calculated by the following formula:
Equation 6
C5 =
●
t ign
3 ⋅ R4
= 47nF
Pin 4 TCH: is the time counter and it is activated at startup as well as after a protection
- HBCS crossing during run mode - triggering. To achieve this, an R9C6 parallel
network is connected between this pin and ground. Fixing a protection time
tTch,reduced=0.27 sec (needed for overcurrent protection) it is possible to calculate C6:
Doc ID 17803 Rev 1
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SMPS design
AN3257
Equation 7
C 6 ≅ t Tch ,reduced ⋅
1
0.26974 ⋅ 10 6
= 1.03μF ⇒ C 6 = 1μF
Choosing tpre=1.25 sec and considering the internal current generator ICH=31 µA, it's
possible to calculate R9:
Equation 8
C6
⋅ 4.63
ICH
= 2.24MΩ ⇒ 2.2MΩ
4.63
C 6 ⋅ ln
1.5
t pre −
R9 =
●
●
●
●
Pin 5 EOLP: is a 2 V reference and allows programming of the window comparator of
pin6 (EOL) according to table 5 of the L6585DE datasheet. Working in CTR tracking
configuration and choosing a window voltage amplitude ± 240 mV, R8=240 kΩ was
inserted.
Pin 6 EOL: is the input for the window comparator dedicated to lighting application
protection (lamp end of life detection). For the HB-LLC resonant converter it was
disabled; to do this, the EOL pin was directly connected to the CTR pin.
Pin 7 CTR: this is a multifunction pin (PFC overvoltage, feedback disconnection,
reference for EOL pin in case of tracking reading), connected to a resistive divider to
the PFC output bus. Establishing the maximum PFC overvoltage PFC output overshoot
(e.g. at startup) of VOVPBUSpfc=456 V and considering that the corresponding threshold
on the CTR pin must be VthrCTR=3.4 V it's possible to calculate R3+R7=2 MΩ and
R15=15 kΩ.
Pin 8 MULT: first, the maximum peak value for VMULT, VMULTmax, is selected. This
value, which occurs at maximum mains voltage, should be 3 V (linearity limit) or nearly
so in wide range mains and less in case of single mains. The PFC sense resistor
selected is RS = R21=0.18 Ω and it is described in the information relating to pin12.
Considering the maximum slope of multiplier maxslope=0.75, the maximum peak
value, occurring at maximum mains voltage is:
Equation 9
VMULT max
ILpk ⋅ R 21
V
=
⋅ AC max =
max slope VAC min
2⋅ 2 ⋅
Pout
⋅ R 21
η ⋅ VAC min ⋅ PF
V
⋅ AC max = 2.46
max slope
VAC min
it's possible to calculate the multiplier divider ε:
Equation 10
ε=
R12
V
R12
= MULT max =
+ (R 5 + R1 )
2 ⋅VAC max
2.46
2 ⋅ 265
= 6.58 ⋅ 10 − 3
Supposing 165 µA is the current flowing into the divider, the lower resistor R12 value can be
calculated and so the upper resistance R5 + R1 value is:
10/28
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AN3257
SMPS design
Equation 11
R12 =
R 5 + R1 =
VMULT max
= 15.2kΩ ⇒ R12 = 15kΩ
165μA
1− ε
⋅ R12 = 2.264MΩ ⇒ R 5 + R1 = 2.2MΩ
ε
The voltage on the multiplier pin with the selected component values is recalculated at
minimum line voltage, 0.86 V, and at maximum line voltage, 2.53 V. So the multiplier works
correctly within its linear region.
●
Pin 9 COMP: is the output of the E/A and also one of the two inputs of the multiplier.
The feedback compensation network, placed between this pin and INV (10), is just the
C19 network:
Equation 12
C19 = 147nF
●
Pin 10 INV: to implement the voltage control loop, a resistive divider is connected
between the regulated output voltage VBUSpfc=400 V of the boost and the pin. The
internal reference on the non inverting input of the E/A is 2.5 V, so R6 and R2 (Figure 3)
is then selected (fixing R14=27 kΩ, R19=360 kΩ, ⇒ R14//R19=25.11 kΩ), as follows:
Equation 13
VBUSpfc
R6 + R2
=
−1
R14 // R19
2. 5
⇒ R 6 + R 2 = 4MΩ
●
Pin 11 ZCD: is the input to the zero-current detector circuit. The ZCD pin is connected
to the auxiliary winding of the boost inductor through a limiting resistor R13. The ZCD
circuit is negative-going edge triggered: when the voltage on the pin falls below 0.7 V
the PWM latch is set and the MOSFET is turned on. To do so, however, the circuit must
be first armed: prior to falling below 0.7 V the voltage on pin 11 must experience a
positive-going edge exceeding 1.4 V (due to MOSFET's turn-off). The maximum main
to-auxiliary winding turn ratio, m, has to ensure that the voltage delivered to the pin
during MOSFET's OFF-time is sufficient to arm the ZCD circuit. Then:
Equation 14
m≤
VBUSpfc − 2 ⋅ VinRMS(max)
1. 4
= 33.10
m=10 was chosen.
Considering the upper and lower clamp voltage of the ZCD pin, its minimum current sink
current capability, according to the max and min voltage of the PFC bus, it was possible to
calculate and to choose R13=56 kΩ.
●
Pin 12 PFCS: is the inverting input of the current sense comparator. As the voltage
across the sense resistor (proportional to the instantaneous inductor current) crosses
the threshold set by the multiplier output, the power MOSFET is turned off. By following
the indication reported below, it's possible to determinate the PFC sense resistor:
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SMPS design
AN3257
Equation 15
PoutTOT
η
= 3.48A
VAC min ⋅ PF
2⋅ 2 ⋅
IL max =
R 21 <
VCS min
= 0.28Ω
IL max
R21=180 mΩ with a power rating of 1 W was chosen.
●
Pin 13 PFG: to correctly drive the external MOSFET, a resistor R18=56 Ω was used.
●
Pin 14 HBCS: considering the tank peak current of the LLC resonant converter
(calculated as described in “Designing LLC resonant converter for optimum efficiency”
EPE2009, Barcelona, Spain, September2009, ISBN:9789075815009) IRpk=0.86 A and
considering the HBCS threshold during the run phase VHBCS=1 V, it is possible to
calculate R27 as:
Equation 16
R 27 <
VHBCS
1
=
= 1.16Ω
IRpk
0.86
R27=0.56 Ω with a power rating of 1 W was chosen.
3.2
●
Pin 15 GND: device ground
●
Pin 16 LSD: to correctly drive the external half bridge low side MOSFET, resistor
R22=56 Ω was used. To reduce the MOSFET turn-off losses, a diode D13=1N4148 was
inserted, in parallel to R22.
●
Pin 17 Vcc: this pin is externally connected to the dynamic startup circuit (by means of
R23, R24, Q4, D6, D15, and R28) and to the self supply circuit composed by the net (R32,
D8, C20, R42, D16, Q6, D7, C22, R46, R43, and Q5).
●
Pin 18 out: high side driver floating reference. This pin is connected close to the source
of the high side power MOSFET.
●
Pin 19 HSD: to correctly drive the external half bridge low side MOSFET, a resistor
R16=56 Ω was used . To reduce the power MOSFET turn-off losses, a diode
D12=1N4148 was inserted, in parallel to R16.
●
Pin 20 boot: for the high side section C11= 100 nF was selected.
PFC power section design
Input capacitor
The input high frequency filter capacitor must attenuate the switching noise due to the high
frequency inductor current ripple. The worst conditions occur on the peak of the minimum
rated input voltage VACmin=90 V. Establishing the following values:
–
the coefficient of maximum high frequency voltage ripple r= 0.12
–
total system efficiency η=0.9
It is possible, considering the minimum PFC switching frequency fminpfc=36 kHz and the
total output power PoutTOT=90 W, to determinate input capacitor C3 in the following way:
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AN3257
SMPS design
Equation 17
PoutTOT
η ⋅ VAC min
= 457nF
C3 =
2 ⋅ π ⋅ fmin pfc ⋅ VAC min ⋅ r
C3=470 nF was inserted.
Output capacitor
The output bulk capacitor C1 selection depends on the DC output voltage, the admitted
overvoltage, the output power, and the desired voltage ripple. Establishing the following
values:
–
PFC output voltage VbusPFC=400 V
–
the coefficient of low frequency (twice the mains frequency fmain=50 Hz) voltage
ripple r1=0.05
It is possible to calculate the bulk capacitor in the following way:
Equation 18
PoutTOT
VbusPFC
= 17.9μF
C1 =
2π ⋅ 2fmain ⋅ VbusPFC ⋅ r1
To have a smaller ripple and good reliability, the following commercial capacitor was chosen:
C1=68 µF, 450 V.
Boost inductor
The inductance Lpfc is usually determined so that the minimum switching frequency fminpfc is
greater than the maximum frequency of the internal starter, to ensure a correct TM
operation. Considering the minimum suggested value for PFC section fminpfc=20 kHz, and
that this last one can occur at the either the maximum VACmax=265 V or the minimum
VACmin=90 V mains voltage, the inductor value is:
Equation 19
To margin from fminpfc, fpfc=36 kHz was chosen. In this condition the lower value for the
inductor is determined by VAC=VACmin and results Lpfc=0.7 mH with, as explained in the
PFCS pin description, a maximum current ILmax=3.48 A. (the inductor 1974.0002 was used,
manufactured by MAGNETICA).
Power MOSFET
The choice of the MOSFET concerns mainly its RDS(on), which depends on the output power
and its breakdown voltage; this last voltage is fixed just by the output voltage Vbuspfc=400 V,
plus the overvoltage ΔVOVPpfc=60 V admitted and a safety margin.
The MOSFET's power dissipation depends on conduction and switching losses.
Establishing a maximum total power loss, PlossesAdm=1 % Pout=0.9 W, it is easy to verify that
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AN3257
choosing MDmesh II power MOSFET STF12NM50N, the estimated total MOSFET power
loss, in the worst case, is about PlossesEst=0.75 W so this choice was the definitive one. To
dissipate the power losses in a better way, a heatsink was added.
Boost diode
The boost freewheeling diode is a fast recovery one. The breakdown voltage is fixed
following a similar criterion as that for the MOSFET 1.2* (Vout + ΔVOVP). The value of its DC
and RMS current, needed to choose the current rating of the diode, are:
Equation 20
ID1dc =
PoutTOT
= 0.225 A
VBUSpfc
ID1rms = 2 2 ⋅ IAC max ⋅
4 2 VAC min
⋅
= 0.62A
9π VBUSpfc
As the PFC works in transition mode, the Turbo 2 Ultrafast high voltage rectifier, STTH2L06,
was selected.
3.3
LLC resonant circuit design
Using what is commonly known as the “first harmonic approximation” (FHA) technique, the
LLC resonant circuit was designed. Considering the following converter specifications:
●
Nominal input DC voltage: 400 V
●
Output Voltage: 19 V @ 4.7 A
●
Resonance frequency: 90 kHz
●
Minimum operating frequency: 60 kHz
●
Maximum operating frequency: 230 kHz
●
Delay time (L6585DE datasheet): 1.2 µs
●
Foreseen capacitance at half bridge node: 120 pF.
It was possible, by means of the AN2450 application note, to calculate the resonant power
transformer and the resonant capacitor specification. The transformer 1860.0045 was used,
manufactured by MAGNETICA:
–
Lp=1200 µH
–
Lr=200 µH
and the resonant capacitor Cr=22 nF.
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AN3257
4
Experimental results
Experimental results
The schematic of the tested board is shown in Figure 3. First of all, the board was tried in
terms of efficiency, power factor, total harmonic distortion, and thermal behavior for the input
voltage range at nominal load; below, in Table 2 and Table 3., the results obtained for a 60
min test are shown.
Table 2.
Board performance
Vin (V)
Iout (A)
Vout (V)
Pout (W)
Pin (W)
Efficiency
(%)
THD (%)
PF
90
4.7
18.86
88.64
98.8
89.72
2.3
0.999
110
4.7
18.86
88.64
97.7
90.73
2.2
0.999
185
4.7
18.86
88.64
96.4
91.95
3.4
0.994
230
4.7
18.86
88.64
96.3
92.05
5.5
0.987
265
4.7
18.86
88.64
96.2
92.14
7.8
0.977
Figure 4.
Efficiency vs. input voltage
(IILFLHQF\
(II#$
,QSXW9ROWDJH9
!-V
The high efficiency of the PFC, working in transition mode, and the very high efficiency of
the resonant stage, working in ZVS, provides, on average, an efficiency better than 91 %.
For low input mains voltage the efficiency is just a little lower because PFC conduction
losses increase.
Table 3.
Thermal results
Ambient
temp (°C)
90
25
59
65
53
37
61
72
44
110
25
53
62
50
37
60.5
70
44
185
25
44.5
55
41
40
57
70
44
MOSPFC (°C) DIODEPFC (°C) LPFC (°C)
MOSHB
(°C)
LResonant DIODEoutput L6585DE
(°C)
(°C)
(°C)
VIN(V)
Doc ID 17803 Rev 1
15/28
Experimental results
Table 3.
AN3257
Thermal results (continued)
VIN(V)
Ambient
temp (°C)
230
25
40.2
52
39.8
39.7
57
70
44
265
25
40.2
50
37.8
39.7
57
70
44
MOSPFC (°C) DIODEPFC (°C) LPFC (°C)
MOSHB
(°C)
LResonant DIODEoutput L6585DE
(°C)
(°C)
(°C)
Table 4 and 5 show the output voltage and efficiency measurement at nominal mains with
different load conditions powering the board at the two nominal input mains voltages.
Table 4.
Vin(V)
Vout(V)
Iout(A)
Pout(W)
Pin(W)
Efficiency(%)
115
18.86
4.7
88.60
97.4
90.96
115
18.88
3.7
69.82
76.5
91.27
115
18.92
2.7
51.06
56
91.17
115
18.95
1.7
32.22
35.8
89.99
115
18.98
1
18.98
21.7
87.47
115
18.99
0.5
9.50
11.7
81.15
115
19
0.25
4.75
6.7
70.90
Table 5.
16/28
Efficiency measurements @ Vin=115 Vac
Efficiency measurements @ Vin=230 Vac
Vin(V)
Vout(V)
Iout(A)
Pout(W)
Pin(W)
Efficiency(%)
230
18.86
4.7
88.64
96.3
92.05
230
18.88
3.7
69.86
76
91.92
230
18.92
2.7
51.08
56
91.22
230
18.95
1.7
32.22
36.1
89.24
230
18.98
1
18.98
22
86.27
230
18.99
0.5
9.50
11.7
81.15
230
19
0.25
4.75
6.3
75.40
Doc ID 17803 Rev 1
AN3257
Experimental results
Figure 5.
Efficiency vs. Pout
(IILFLHQF\
(II#9DF
(II#9DF
,RXW$
!-V
At lower loads the efficiency decreases because HB works at high frequency so the
switching losses increase.
Resonant stage operating waveforms
In Figure 6 some waveforms are shown (HB middle point voltage, primary winding current,
and oscillator voltage) during steady-state operation of the circuit at full load.
Figure 6.
Resonant circuit primary side waveforms
The transformer primary current wave is almost sinusoidal because the operating frequency
is slightly above the resonance one. In this condition, the circuit has a good margin for ZVS
operations, providing good efficiency, and the almost sinusoidal wave shape provides for an
extremely low EMI generation.
In Figure 7 some waveforms relevant to the secondary side (output voltage ripple, current of
one output diode, and output diode voltage) are shown.
Doc ID 17803 Rev 1
17/28
Experimental results
Figure 7.
AN3257
Resonant circuit secondary side waveforms
The current in the diode is almost a sine wave and its average value is half of the output
current while the rectifiers reverse voltage is, according to theoretical value, 2VOUT+VF. The
high frequency ripple of the output voltage is only 180 mV (0.93 %).
No load power consumption and Load transition
In Table 6 the power consumption in no load condition is given, while Figure 8 shows the
waveforms of the output voltage and the current during a load variation from 0 % to 100 %;
the circuit response is fast enough to avoid output voltage dips. In Figure 9, the opposite
load transition is checked (100 % to 0 %). Also in this case the transition doesn't show any
problems for the output voltage regulation.
Table 6.
18/28
No load consumption
Vin (V)
Iout (A)
Vout (V)
Pout (W)
Pin (W)
90
0
19.01
0
1.3
110
0
19.01
0
1.2
185
0
19.01
0
1.1
230
0
19.01
0
1.1
265
0
19.01
0
1.1
Doc ID 17803 Rev 1
AN3257
Experimental results
Figure 8.
Load transition 0 % ==> 100 %
Figure 9.
Load transition 100 % ==> 0 %
Short-circuit protections and startup sequence
The L6585DE is equipped with, in the HB section, a current sensing input (pin14, HBCS)
and a dedicated overcurrent management system. The current flowing in the circuit is
detected and the signal is fed into the HBCS pin. It is internally connected to the input of the
first comparator referenced to 1.6 V and to that of a second comparator referenced to 2.75
V. When one of the two comparators is activated, the IC is latched in low consumption
mode.
In Figure 10 the response system in short-circuit condition during run mode is shown; as
soon as HBCS=1.6 V the IC is stopped.
Doc ID 17803 Rev 1
19/28
Experimental results
AN3257
Figure 10. Short-circuit during run mode
Figure 11 shows the waveforms during startup at 115 Vac and full load.
Figure 11. Startup [email protected] V - full load
20/28
Doc ID 17803 Rev 1
AN3257
5
Conducted emission pre-compliance test
Conducted emission pre-compliance test
The limits indicated on both diagrams at 115 Vac and 230 Vac comply with EN55022 ClassB specifications. The values are measured in peak detection mode.
Figure 12. CE at 115 Vac 50 Hz - line 1 peak detector
Figure 13. CE at 115 Vac 50 Hz - line 2 peak detector
Doc ID 17803 Rev 1
21/28
Conducted emission pre-compliance test
Figure 14. CE at 230 Vac 50 Hz - line 1 peak detector
Figure 15. CE at 230 Vac 50 Hz - line 2 peak detector
22/28
Doc ID 17803 Rev 1
AN3257
BOM list
Table 7.
BOM list
BOM list
23/28
6
Doc ID 17803 Rev 1
Reference
Part / value
Technology information
Package
Manufacturer code
C1
68 µF 450 V
Electrolytic aluminium
capacitor
TH Radial
EPCOS
B43890A5686M000
C3
470 nF 630 Vdc
Metallized polypropylene
film capacitor
TH Radial
EPCOS
B32653A6474K000
C4
470 pF, 25 V
Ceramic capacitor COG, 5 %
SMD 0805
C5
47 nF, 25 V
Ceramic capacitor
SMD 0805
C6
1 µF, 25 V
Ceramic capacitor
SMD 0805
C7
10 nF, 25 V
Ceramic capacitor
SMD 0805
C8,C9
305 V X2, 470 nF, 10 %
Polypropylene
TH Radial
C10
1 nF 250 V Y1
Ceramic capacitor
TH Radial
C11
100 nF 50 V
Ceramic capacitor
SMD 1206
C12
470 µF,
35 V
Electrolytic aluminium
capacitor
TH Radial
C13
470 µF,
35 V
Electrolytic aluminium
capacitor
TH Radial
C14
22 nF 630 V
Polypropylene
TH Radial
C15
2.2n-Y1- 250 V
Ceramic capacitor
TH Radial
C16
1 µF, 25 V
Ceramic capacitor
SMD 0805
C19
150 nF, 25 V
ceramic capacitor
SMD 0805
C20
100 µF 35 V
Electrolytic aluminium
capacitor
TH Radial
C21
220 nF, 25 V
ceramic capacitor
SMD 0805
EPCOS
B32922C3474K000
AN3257
BOM list
Doc ID 17803 Rev 1
Part / value
Technology information
Package
C22
470 µF 25 V
Electrolytic aluminium
capacitor
TH radial
D1
600 V, 3 A
Turbo 2
ultrafast high
voltage rectifier
DO41
STMicroelectronics
STTH2L06
D2+heatsink
800 V, 4 A
Bridge
TH
GBU8K
D3,D4 +heatsink
10 A, 60 V
Power schottky rectifier
TO-220
STMicroelectronics
STPS10L60FP
D6,D7,D8,
D12, D13,D14
150 mA, 100 V
Schottky rectifier
Minimelf
STMicroelectronics
TMMBAT46
D15
12 V
Zener diode
Minimelf
D16
15 V
Zener diode
Minimelf
F1
3A
Fuse
TH
ISO1
PC817
Photo Coup
TH
J1
CON3
Terminal
pin distance 7.62 mm
TH
J2
CON2
Terminal
pin distance 5 mm
TH
LC1
2x12 mH/1.8 A
Common choke
TH
TDK
HF2826-123Y1R8-T01
LPFC1
520 µH,1.4 A
PFC inductor
TH
Magnetica
1974.002
Q1,Q2,Q3+ heatsink
STF12NM50N
MDMESH™ II MOSFET
TO220-FP
STMicroelectronics
STF12NM50N
Q4
STQ1HNK60R
SuperMESH™ MOSFET
TH
STMicroelectronics
STQ1HNK60R
Q5,Q6
BC847
Small signal bipolar
SMD SOT 23
R1
1.2 MΩ, 1 %, 1/4 W
SMD 1206
Manufacturer code
BOM list
24/28
Reference
AN3257
Table 7.
BOM list
Doc ID 17803 Rev 1
Part / value
Technology information
Package
R2
1.8 MΩ, 1 %, 1/4 W
SMD 1206
R3,R5,R7
1 MΩ, 1 %, 1/4 W
SMD1206
R4
330 kΩ, 1 %, 1/8 W
SMD 0805
R6
2.2 MΩ, 1 %, 1/4 W
SMD1206
R9
2.2 MΩ, 1 %, 1/8 W
SMD 0805
R8
240 kΩ, 1 %, 1/8 W
SMD 0805
R10
Not mounted
SMD 0805
R11
43 kΩ,1 %, 1/4 W
SMD 0805
R12,R15
15 kΩ, 1 %, 1/8 W
SMD 0805
R17
15 kΩ, 1 %, 1/4 W
SMD 1206
R13
56 kΩ, 1 %, 1/4 W
SMD 1206
R14
27 kΩ, 1 %, 1/4 W
SMD 1206
R16,R18,R22
56 Ω, 1 %, 1/4 W
SMD 1206
R19
360 kΩ, 1 %, 1/4 W
SMD 1206
R20
Not mounted
TH, 1 W
R21
0.180 Ω, 1 %, 1 W
TH radial
R23,R24
2.4 MΩ, 1 %, 1/4 W
SMD 1206
R26
Not mounted
TH, 1 W
R27
0.56 Ω, 1 %, 1 W
TH Radial
R28
10 kΩ, 5 %,1/4 W
SMD 1206
R29
5.6 kΩ, 1 %, 1/8 W
SMD 0805
R32
36 Ω,1 %, 1 W
SMD 1206
R33
39 kΩ, 1 %,1/8 W
SMD 0805
R34
47 kΩ, 1 %, 1/8 W
SMD 0805
R35
6.2 kΩ, 1 %, 1/8 W
SMD 0805
Manufacturer code
AN3257
Reference
BOM list
25/28
Table 7.
BOM list
Doc ID 17803 Rev 1
Reference
Part / value
Technology information
Package
Manufacturer code
R36
120 kΩ, 1 %, 1/8 W
SMD 0805
R37,R38,R39, R44
0 Ω, 1/4 W, 5 %
SMD 1206
R40
1 kΩ, 1 %, 1/8 W
SMD 0805
R42
4.7 kΩ, 1 %, 1/8 W
SMD 0805
R43
6.8 kΩ, 1 %, 1/8 W
SMD 0805
R46
150 kΩ, 1 %, 1/8 W
SMD 0805
T1
TRAFO
TH
MAGNETICA
1860.0045
U1
L6585DE
SMD SO20
STMicroelectronics
L6585DE
U2
TL431
TH TO92
STMicroelectronics
TL431
AN3257
Table 7.
BOM list
26/28
AN3257
7
Revision history
Revision history
Table 8.
Document revision history
Date
Revision
31-Mar-2011
1
Changes
Initial release.
Doc ID 17803 Rev 1
27/28
AN3257
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