LTC3410B - 2.25MHz, 300mA Synchronous Step-Down Regulator in SC70

LTC3410B
2.25MHz, 300mA
Synchronous Step-Down
Regulator in SC70
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FEATURES
DESCRIPTIO
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The LTC ®3410B is a high efficiency monolithic synchronous buck regulator using a constant frequency, current
mode architecture. The device is available in adjustable
and fixed output voltage versions. Supply current during
operation is only 200µA, dropping to <1µA in shutdown.
The 2.5V to 5.5V input voltage range makes the LTC3410B
ideally suited for single Li-Ion battery-powered applications. 100% duty cycle provides low dropout operation,
extending battery life in portable systems. PWM pulse
skipping mode operation provides very low output ripple
voltage for noise sensitive applications.
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High Efficiency: Up to 96%
300mA Output Current at VIN = 3V
380mA Minimum Peak Switch Current
2.5V to 5.5V Input Voltage Range
2.25MHz Constant Frequency Operation
No Schottky Diode Required
Low Dropout Operation: 100% Duty Cycle
Stable with Ceramic Capacitors
0.8V Reference Allows Low Output Voltages
Shutdown Mode Draws < 1µA Supply Current
±2% Output Voltage Accuracy
Current Mode Operation for Excellent Line and
Load Transient Response
Overtemperature Protected
Available in Low Profile SC70 Package
Switching frequency is internally set at 2.25MHz, allowing
the use of small surface mount inductors and capacitors.
The LTC3410B is specifically designed to work well with
ceramic output capacitors, achieving very low output
voltage ripple and a small PCB footprint.
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APPLICATIO S
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The internal synchronous switch increases efficiency and
eliminates the need for an external Schottky diode. Low
output voltages are easily supported with the 0.8V feedback reference voltage. The LTC3410B is available in a
tiny, low profile SC70 package.
Cellular Telephones
Personal Information Appliances
Wireless and DSL Modems
Digital Still Cameras
MP3 Players
Portable Instruments
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All
other trademarks are the property of their respective owners. Protected by U.S. Patents,
including 5481178, 5994885, 6580258, 6304066, 6127815, 6498466, 6611131.
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Efficiency and Power Loss
vs Output Current
TYPICAL APPLICATIO
1
100
90
4.7µH
SW
LTC3410B
10pF
RUN
VFB
GND
232k
464k
3410 TA01
COUT
2.2µF
CER
EFFICIENCY
80
VOUT
1.2V
EFFICIENCY (%)
CIN
2.2µF
CER
VIN
0.1
70
60
50
0.01
POWER LOSS
40
30
POWER LOSS (W)
VIN
2.7V
TO 5.5V
0.001
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
20
0
1
10
100
OUTPUT CURRENT (mA)
0.0001
1000
3410 TA01b
3410bfa
1
LTC3410B
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AXI U
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ABSOLUTE
RATI GS
(Note 1)
Input Supply Voltage .................................. – 0.3V to 6V
RUN, VFB Voltages ..................................... – 0.3V to VIN
SW Voltage (DC) ......................... – 0.3V to (VIN + 0.3V)
P-Channel Switch Source Current (DC) ............. 500mA
N-Channel Switch Sink Current (DC) ................. 500mA
Peak SW Sink and Source Current .................... 630mA
Operating Temperature Range (Note 2) .. – 40°C to 85°C
Junction Temperature (Note 3) ............................ 125°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
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PACKAGE/ORDER I FOR ATIO
TOP VIEW
TOP VIEW
RUN 1
6 VFB
RUN 1
6 VOUT
GND 2
5 GND
GND 2
5 GND
SW 3
4 VIN
SW 3
4 VIN
SC6 PACKAGE
6-LEAD PLASTIC SC70
SC6 PACKAGE
6-LEAD PLASTIC SC70
TJMAX = 125°C, θJA = 250°C/ W
ORDER PART NUMBER
LTC3410BESC6
SC6 PART MARKING
LBZY
TJMAX = 125°C, θJA = 250°C/ W
ORDER PART NUMBER
LTC3410BESC6-1.2
LTC3410BESC6-1.5
LTC3410BESC6-1.8
LTC3410BESC6-1.875
SC6 PART MARKING
LCMX
LCMY
LCMZ
LCHZ
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
IVFB
Feedback Current
Adjustable Output Voltage
●
MIN
IVOUT
Output Voltage Feedback Current
Fixed Output Voltage
●
IPK
Peak Inductor Current
VIN = 3V, VFB = 0.7V or VOUT = 90%, Duty Cycle < 35%
VFB
Regulated Feedback Voltage
Adjustable Output Voltage (LTC3410BE)
●
∆VFB
Reference Voltage Line Regulation
VIN = 2.5V to 5.5V
●
VOUT
Regulated Output Voltage
LTC3410B-1.2, IOUT = 100mA
LTC3410B-1.5, IOUT = 100mA
LTC3410B-1.8, IOUT = 100mA
LTC3410B-1.875, IOUT = 100mA
●
●
●
●
∆VOUT
Output Voltage Line Regulation
VIN = 2.5V to 5.5V
●
VLOADREG
Output Voltage Load Regulation
ILOAD = 50mA to 250mA
VIN
Input Voltage Range
VUVLO
Undervoltage Lockout Threshold
3.3
MAX
UNITS
±30
nA
6
µA
mA
380
490
600
0.784
0.8
0.816
0.04
0.4
1.2
1.5
1.8
1.875
1.224
1.53
1.836
1.913
0.04
0.4
1.176
1.47
1.764
1.837
0.5
●
VIN Rising
VIN Falling
TYP
2.5
2.0
1.94
V
%/V
V
V
V
V
%/V
%
5.5
V
2.3
V
V
3410bfa
2
LTC3410B
ELECTRICAL CHARACTERISTICS
The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.
VIN = 3.6V unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
IS
Input DC Bias Current
Operating
Shutdown
(Note 4)
VFB = 0.83V or VOUT = 104%, ILOAD = 0A
VRUN = 0V
200
0.1
300
1
µA
µA
fOSC
Oscillator Frequency
VFB = 0.8V or VOUT = 100%
VFB = 0V or VOUT = 0V
2.25
310
2.7
MHz
kHz
RPFET
RDS(ON) of P-Channel FET
ISW = 100mA
0.75
0.9
Ω
RNFET
RDS(ON) of N-Channel FET
ISW = –100mA
0.55
0.7
Ω
ILSW
SW Leakage
VRUN = 0V, VSW = 0V or 5V, VIN = 5V
±0.01
±1
µA
VRUN
RUN Threshold
●
1
1.5
V
IRUN
RUN Leakage Current
●
±0.01
±1
µA
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3410BE is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
LTC3410B: TJ = TA + (PD)(250°C/W)
●
1.8
0.3
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1 Except for the Resistive Divider Resistor Values)
Efficiency vs Input Voltage
Reference Voltage vs
Temperature
Efficiency vs Output Current
0.814
100
100
VIN = 3.6V
90
90 IOUT = 100mA
0.809
70
IOUT = 10mA
60 IOUT = 1mA
50
REFERENCE VOLTAGE (V)
EFFICIENCY (%)
IOUT = 250mA
EFFICIENCY (%)
80
80
70
60
50
40
30
40
20
30
VOUT = 1.2V
20
3
3.5
4.5
2.5
4
INPUT VOLTAGE (V)
10
0
5
5.5
3410 G01
VOUT = 1.8V
1
10
100
OUTPUT CURRENT (mA)
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
1000
3410 G02
0.804
0.799
0.794
0.789
0.784
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
3410 G03
3410bfa
3
LTC3410B
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1 Except for the Resistive Divider Resistor Values)
Oscillator Frequency vs
Temperature
VIN = 3.6V
OSCILLATOR FREQUENCY (MHz)
2.5
2.4
2.3
2.2
2.1
2.0
1.9
1.8
–50
2.7
1.0
2.6
0.6
2.4
2.3
2.2
2.1
0
25
50
75
TEMPERATURE (°C)
100
6
3
5
4
SUPPLY VOLTAGE (V)
1.0
0.9
MAIN SWITCH
VIN = 3.6V
0.8
0.7
0.6
SYNCHRONOUS SWITCH
0.4
0.6
VIN = 4.2V
0.4
VIN = 2.7V
0.3
0.2
0.2
7
VIN = 3.6V
MAIN SWITCH
SYNCHRONOUS SWITCH
0.1
0
–50 –30 –10 10 30 50 70 90 110 130
TEMPERATURE (°C)
3410 G07
VOUT = 1.2V
ILOAD = 0A
260
220
180
140
100
1
2
4
3
5
6
VIN (V)
3410 G09
3410 G08
Dynamic Supply Current
vs Temperature
Switch Leakage vs Temperature
250
110
VOUT = 1.2V
ILOAD = 0A
100
VIN = 5.5V
RUN = 0V
90
230
SWITCH LEAKAGE (nA)
DYNAMIC SUPPLY CURRENT (µA)
500
300
VIN = 2.7V
RDS (ON) (Ω)
RDS (ON) (Ω)
300
400
200
LOAD CURRENT (mA)
Dynamic Supply Current vs VIN
VIN = 4.2V
1.0
6
100
3410 G06
RDS(ON) vs Temperature
5
4
3
INPUT VOLTAGE (V)
0
3410 G05
1.2
2
–1.8
–3.0
2
1.1
1
–1.4
–2.6
RDS(ON) vs Input Voltage
0
–1.0
–2.2
125
1.2
0.5
–0.6
1.9
3410 G04
0.8
–0.2
2.0
1.8
–25
VIN = 3.6V
VOUT = 1.8V
0.2
2.5
DYNAMIC SUPPLY CURRENT (µA)
OSCILLATOR FREQUENCY (MHz)
2.6
Output Voltage vs Load Current
VOUT ERROR (%)
2.7
Oscillator Frequency vs
Supply Voltage
210
190
170
80
70
SYNCHRONOUS
SWITCH
60
50
40
30
MAIN
SWITCH
20
10
150
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3410 G10
0
–50
–25
50
25
0
75
TEMPERATURE (°C)
100
125
3410 G11
3410bfa
4
LTC3410B
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TYPICAL PERFOR A CE CHARACTERISTICS
(From Figure 1 Except for the Resistive Divider Resistor Values)
Start-Up from Shutdown
Pulse Skipping
Switch Leakage vs Input Voltage
600
550
VOUT
10mV/DIV
AC COUPLED
LEAKAGE CURRENT (pA)
500
450
RUN
2V/DIV
400
350
MAIN
SWITCH
300
SW
2V/DIV
VOUT
1V/DIV
IL
100mA/DIV
IL
200mA/DIV
250
200
150
SYNCHRONOUS
SWITCH
100
50
0
0
1
4
3
2
INPUT VOLTAGE (V)
5
6
VIN = 3.6V
VOUT = 1.8V
ILOAD = 2mA
1µs/DIV
3410 G13
VIN = 3.6V
VOUT = 1.8V
ILOAD = 128mA
100µs/DIV
3410 G14
3410 G12
Load Step
Load Step
VOUT
100mV/DIV
AC COUPLED
VOUT
100mV/DIV
AC COUPLED
IL
200mA/DIV
IL
200mA/DIV
ILOAD
200mA/DIV
ILOAD
200mA/DIV
4µs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 0mA TO 300mA
3410 G15
4µs/DIV
VIN = 3.6V
VOUT = 1.8V
ILOAD = 30mA TO 300mA
3410 G16
3410bfa
5
LTC3410B
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PI FU CTIO S
RUN (Pin 1): Run Control Input. Forcing this pin above
1.5V enables the part. Forcing this pin below 0.3V shuts
down the device. In shutdown, all functions are disabled
drawing <1µA supply current. Do not leave RUN floating.
GND (Pins 2, 5): Ground Pin.
SW (Pin 3): Switch Node Connection to Inductor. This pin
connects to the drains of the internal main and synchronous power MOSFET switches.
VIN (Pin 4): Main Supply Pin. Must be closely decoupled
to GND, Pin 2, with a 2.2µF or greater ceramic capacitor.
VFB (Pin 6 on Adjustable Version): Feedback Pin. Receives the feedback voltage from an external resistive
divider across the output.
VOUT (Pin 6 on Fixed Voltage Versions): Output Voltage
Feedback Pin. An internal resistive divider divides the
output voltage down for comparison to the internal reference voltage.
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FU CTIO AL DIAGRA
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SLOPE
COMP
OSC
OSC
4 VIN
FREQ
SHIFT
–
VFB/VOUT
+
6
0.8V
R1*
+
R2
240k
RUN
0.8V REF
5Ω
+
ICOMP
S
Q
R
Q
RS LATCH
VIN
1
–
– EA
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
ANTISHOOTTHRU
3 SW
+
(
V
*R1 = 240k OUT – 1
0.8
)
IRCMP
5
–
SHUTDOWN
2 GND
3410 BD
3410bfa
6
LTC3410B
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OPERATIO (Refer to Functional Diagram)
Main Control Loop
The LTC3410B uses a constant frequency, current mode
step-down architecture. Both the main (P-channel
MOSFET) and synchronous (N-channel MOSFET) switches
are internal. During normal operation, the internal top
power MOSFET is turned on each cycle when the oscillator
sets the RS latch, and turned off when the current comparator, ICOMP, resets the RS latch. The peak inductor
current at which ICOMP resets the RS latch, is controlled by
the output of error amplifier EA. The VFB pin, described in
the Pin Functions section, allows EA to receive an output
feedback voltage from an external resistive divider. When
the load current increases, it causes a slight decrease in
the feedback voltage relative to the 0.8V reference, which
in turn, causes the EA amplifier’s output voltage to increase until the average inductor current matches the new
load current. While the top MOSFET is off, the bottom
MOSFET is turned on until either the inductor current
starts to reverse, as indicated by the current reversal
comparator IRCMP, or the beginning of the next clock cycle.
Pulse Skipping Mode Operation
At light loads, the inductor current may reach zero or reverse on each pulse. The bottom MOSFET is turned off by
the current reversal comparator, IRCMP, and the switch
voltage will ring. This is discontinuous mode operation,
and is normal behavior for the switching regulator. At very
light loads, the LTC3410B will automatically skip pulses in
pulse skipping mode operation to maintain output regulation. Refer to LTC3410 data sheet if Burst Mode® operation
is preferred.
Short-Circuit Protection
When the output is shorted to ground, the frequency of the
oscillator is reduced to about 310kHz, 1/7 the nominal
frequency. This frequency foldback ensures that the inductor current has more time to decay, thereby preventing
runaway. The oscillator’s frequency will progressively
increase to 2.25MHz when VFB rises above 0V.
Dropout Operation
As the input supply voltage decreases to a value approaching the output voltage, the duty cycle increases toward the
maximum on-time. Further reduction of the supply voltage forces the main switch to remain on for more than one
cycle until it reaches 100% duty cycle. The output voltage
will then be determined by the input voltage minus the
voltage drop across the P-channel MOSFET and the
inductor.
Another important detail to remember is that at low input
supply voltages, the RDS(ON) of the P-channel switch
increases (see Typical Performance Characteristics).
Therefore, the user should calculate the power dissipation
when the LTC3410B is used at 100% duty cycle with low
input voltage (See Thermal Considerations in the Applications Information section).
Slope Compensation and Inductor Peak Current
Slope compensation provides stability in constant frequency architectures by preventing subharmonic oscillations at high duty cycles. It is accomplished internally by
adding a compensating ramp to the inductor current
signal at duty cycles in excess of 40%. Normally, this
results in a reduction of maximum inductor peak current
for duty cycles > 40%. However, the LTC3410B uses a
patented scheme that counteracts this compensating
ramp, which allows the maximum inductor peak current
to remain unaffected throughout all duty cycles.
Burst Mode is a Registered Trademark of Linear Technology Corporation.
3410bfa
7
LTC3410B
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APPLICATIO S I FOR ATIO
VIN
2.7V
TO 5.5V
4.7µH
CIN
2.2µF
CER
VIN
SW
LTC3410B
10pF
RUN
VOUT
1.2V
COUT
2.2µF
CER
VFB
GND
232k
464k
3410 F01
Figure 1. High Efficiency Step-Down Converter
The basic LTC3410B application circuit is shown in Figure 1. External component selection is driven by the load
requirement and begins with the selection of L followed by
CIN and COUT.
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials
are small and don’t radiate much energy, but generally cost
more than powdered iron core inductors with similar
electrical characteristics. The choice of which style inductor to use often depends more on the price vs size requirements and any radiated field/EMI requirements than on
what the LTC3410B requires to operate. Table 1 shows
some typical surface mount inductors that work well in
LTC3410B applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER PART NUMBER
Inductor Selection
For most applications, the value of the inductor will fall in
the range of 2.2µH to 4.7µH. Its value is chosen based on
the desired ripple current. Large value inductors lower
ripple current and small value inductors result in higher
ripple currents. Higher VIN or VOUT also increases the ripple
current as shown in Equation 1. A reasonable starting point
for setting ripple current is ∆IL = 120mA (40% of 300mA).
⎛ V ⎞
1
VOUT ⎜ 1 − OUT ⎟
∆IL =
( f)(L) ⎝ VIN ⎠
Inductor Core Selection
(1)
The DC current rating of the inductor should be at least
equal to the maximum load current plus half the ripple
current to prevent core saturation. Thus, a 360mA rated
inductor should be enough for most applications (300mA
+ 60mA). For better efficiency, choose a low DC-resistance
inductor.
Taiyo Yuden
MAX DC
VALUE CURRENT DCR HEIGHT
CB2016T2R2M
CB2012T2R2M
LBC2016T3R3M
2.2µH
2.2µH
3.3µH
510mA
530mA
410mA
0.13Ω 1.6mm
0.33Ω 1.25mm
0.27Ω 1.6mm
Panasonic
ELT5KT4R7M
4.7µH
950mA
0.2Ω 1.2mm
Sumida
CDRH2D18/LD
4.7µH
630mA 0.086Ω 2mm
Murata
LQH32CN4R7M23 4.7µH
450mA
Taiyo Yuden
NR30102R2M
NR30104R7M
2.2µH
4.7µH
1100mA 0.1Ω 1mm
750mA 0.19Ω 1mm
FDK
FDKMIPF2520D
FDKMIPF2520D
FDKMIPF2520D
4.7µH
3.3µH
2.2µH
1100mA 0.11Ω 1mm
1200mA 0.1Ω 1mm
1300mA 0.08Ω 1mm
0.2Ω
2mm
CIN and COUT Selection
In continuous mode, the source current of the top MOSFET is a square wave of duty cycle VOUT/VIN. To prevent
large voltage transients, a low ESR input capacitor sized
for the maximum RMS current must be used. The maximum RMS capacitor current is given by:
⎡⎣ VOUT ( VIN − VOUT ) ⎤⎦
VIN
1/22
CIN required IRMS ≅ IOMAX
3410bfa
8
LTC3410B
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APPLICATIO S I FOR ATIO
This formula has a maximum at VIN = 2VOUT, where
IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that the capacitor
manufacturer’s ripple current ratings are often based on
2000 hours of life. This makes it advisable to further derate
the capacitor, or choose a capacitor rated at a higher
temperature than required. Always consult the manufacturer if there is any question.
Using Ceramic Input and Output Capacitors
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating
generally far exceeds the IRIPPLE(P-P) requirement. The
output ripple ∆VOUT is determined by:
However, care must be taken when ceramic capacitors are
used at the input and the output. When a ceramic capacitor
is used at the input and the power is supplied by a wall
adapter through long wires, a load step at the output can
induce ringing at the input, VIN. At best, this ringing can
couple to the output and be mistaken as loop instability. At
worst, a sudden inrush of current through the long wires
can potentially cause a voltage spike at VIN, large enough
to damage the part.
⎛
1 ⎞
∆VOUT ≅ ∆IL ⎜ ESR +
⎟
⎝
8fC OUT ⎠
where f = operating frequency, COUT = output capacitance
and ∆IL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ∆IL increases with input voltage.
If tantalum capacitors are used, it is critical that the
capacitors are surge tested for use in switching power
supplies. An excellent choice is the AVX TPS series of
surface mount tantalum. These are specially constructed
and tested for low ESR so they give the lowest ESR for a
given volume. Other capacitor types include Sanyo
POSCAP, Kemet T510 and T495 series, and Sprague 593D
and 595D series. Consult the manufacturer for other
specific recommendations.
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them ideal
for switching regulator applications. Because the
LTC3410B’s control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
Output Voltage Programming (LTC3410B Only)
The output voltage is set by a resistive divider according
to the following formula:
⎛ R2⎞
VOUT = 0.8V ⎜ 1 + ⎟
⎝ R1⎠
(2)
The external resistive divider is connected to the output,
allowing remote voltage sensing as shown in Figure 2.
0.8V ≤ VOUT ≤ 5.5V
R2
VFB
LTC3410B
R1
GND
3410 F02
Figure 2. Setting the LTC3410B Output Voltage
3410bfa
9
LTC3410B
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APPLICATIO S I FOR ATIO
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, two main sources usually account for most of the
losses in LTC3410B circuits: VIN quiescent current and I2R
losses. The VIN quiescent current loss dominates the
efficiency loss at very low load currents whereas the I2R
loss dominates the efficiency loss at medium to high load
currents. In a typical efficiency plot, the efficiency curve at
very low load currents can be misleading since the actual
power lost is of no consequence as illustrated in Figure 3.
1. The VIN quiescent current is due to two components:
the DC bias current as given in the electrical characteristics and the internal main switch and synchronous
switch gate charge currents. The gate charge current
results from switching the gate capacitance of the
internal power MOSFET switches. Each time the gate is
switched from high to low to high again, a packet of
charge, dQ, moves from VIN to ground. The resulting
dQ/dt is the current out of VIN that is typically larger than
the DC bias current. In continuous mode,
IGATECHG = f(QT + QB) where QT and QB are the
gate charges of the internal top and bottom
switches. Both the DC bias and gate charge
losses are proportional to VIN and thustheir effectswill
be more pronounced at higher supply voltages.
2. I2R losses are calculated from the resistances of the
internal switches, RSW, and external inductor RL. In
continuous mode, the average output current flowing
through inductor L is “chopped” between the main
switch and the synchronous switch. Thus, the series
resistance looking into the SW pin is a function of both
top and bottom MOSFET RDS(ON) and the duty cycle
(DC) as follows:
RSW = (RDS(ON)TOP)(DC) + (RDS(ON)BOT)(1 – DC)
The RDS(ON) for both the top and bottom MOSFETs can
be obtained from the Typical Performance Charateristics
curves. Thus, to obtain I2R losses, simply add RSW to
RL and multiply the result by the square of the average
output current.
Other losses including CIN and COUT ESR dissipative
losses and inductor core losses generally account for less
than 2% total additional loss.
1
POWER LOST (W)
0.1
0.01
0.001
0.0001
0.1
VOUT = 1.2V
VOUT = 1.8V
VOUT = 2.5V
1
10
100
LOAD CURRENT (mA)
1000
3410 F03
Figure 3. Power Lost vs Load Current
3410bfa
10
LTC3410B
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APPLICATIO S I FOR ATIO
Thermal Considerations
In most applications the LTC3410B does not dissipate
much heat due to its high efficiency. But, in applications
where the LTC3410B is running at high ambient
temperature with low supply voltage and high duty
cycles, such as in dropout, the heat dissipated may
exceed the maximum junction temperature of the part. If
the junction temperature reaches approximately 150°C,
both power switches will be turned off and the SW node
will become high impedance.
To avoid the LTC3410B from exceeding the maximum
junction temperature, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated exceeds the
maximum junction temperature of the part. The temperature rise is given by:
TR = (PD)(θJA)
where PD is the power dissipated by the regulator and
θJAis the thermal resistance from the junction of the die to
the ambient temperature.
The junction temperature, TJ, is given by:
T J = TA + TR
where TA is the ambient temperature.
As an example, consider the LTC3410B in dropout at an
input voltage of 2.7V, a load current of 300mA and an
ambient temperature of 70°C. From the typical performance graph of switch resistance, the RDS(ON) of the
P-channel switch at 70°C is approximately 1.0Ω.
Therefore, power dissipated by the part is:
PD = ILOAD2 • RDS(ON) = 90mW
For the SC70 package, the θJA is 250°C/ W. Thus, the
junction temperature of the regulator is:
TJ = 70°C + (90)(250) = 92.5°C
which is well below the maximum junction temperature
of 125°C.
Note that at higher supply voltages, the junction temperature is lower due to reduced switch resistance (RDS(ON)).
Checking Transient Response
The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to (∆ILOAD • ESR), where ESR is the effective series
resistance of COUT. ∆ILOAD also begins to charge or
discharge COUT, which generates a feedback error signal.
The regulator loop then acts to return VOUT to its steadystate value. During this recovery time VOUT can be monitored for overshoot or ringing that would indicate a stability
problem. For a detailed explanation of switching control
loop theory, see Application Note 76.
A second, more severe transient is caused by switching in
loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the load
switch resistance is low and it is driven quickly. The only
solution is to limit the rise time of the switch drive so that
the load rise time is limited to approximately (25 • CLOAD).
Thus, a 10µF capacitor charging to 3.3V would require a
250µs rise time, limiting the charging current to about
130mA.
3410bfa
11
LTC3410B
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APPLICATIO S I FOR ATIO
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3410B. These items are also illustrated graphically in
Figures 4 and 5. Check the following in your layout:
1. The power traces, consisting of the GND trace, the SW
trace and the VIN trace should be kept short, direct and
wide.
2. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1/R2 must be connected between the (+) plate of COUT and ground.
3. Does the (+) plate of CIN connect to VIN as closely as
possible? This capacitor provides the AC current to the
internal power MOSFETs.
4. Keep the (–) plates of CIN and COUT as close as possible.
5. Keep the switching node, SW, away from the sensitive
VFB node.
1
1
RUN
LTC3410B
2
–
VFB
GND
2
R2
3
+
LTC3410B-1.875
6
COUT
VOUT
L1
VIN
SW
5
RUN
–
R1
3
+
CFWD
6
COUT
VOUT
4
VOUT
GND
L1
CIN
VIN
SW
5
VIN
4
CIN
VIN
3410B F04a
BOLD LINES INDICATE HIGH CURRENT PATHS
BOLD LINES INDICATE HIGH CURRENT PATHS
3410B F04b
Figure 4b. LTC3410B-1.875 Layout Diagram
Figure 4a. LTC3410B Layout Diagram
VIA TO GND
R1
VOUT
VIN
VIA TO VIN
L1
PIN 1
L1
CFWD
LTC3410B
VIN
VIA TO VIN
VIA TO VOUT
R2
PIN 1
VOUT
SW
LTC3410B1.875
SW
COUT
CIN
COUT
CIN
GND
3410B F05a
Figure 5a. LTC3410B Suggested Layout
3410B F05b
Figure 5b. LTC3410B Fixed Output Voltage
Suggested Layout
3410bfa
12
LTC3410B
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APPLICATIO S I FOR ATIO
Design Example
For best efficiency choose a 300mA or greater inductor
with less than 0.3Ω series resistance.
As a design example, assume the LTC3410B is used in a
single lithium-ion battery-powered cellular phone
application. The VIN will be operating from a maximum of
4.2V down to about 2.7V. The load current requirement
is a maximum of 0.3A but most of the time it will be in
standby mode, requiring only 2mA. Efficiency at both low
and high load currents is important. Output voltage is
2.5V. With this information we can calculate L using
Equation (1),
L=
⎛ V ⎞
1
VOUT ⎜ 1− OUT ⎟
VIN ⎠
( f )( ∆IL )
⎝
CIN will require an RMS current rating of at least 0.125A ≅
ILOAD(MAX)/2 at temperature and COUT will require an ESR
of less than 0.5Ω. In most cases, a ceramic capacitor will
satisfy this requirement.
For the feedback resistors, choose R1 = 412k. R2 can
then be calculated from equation (2) to be:
⎛V
⎞
R2 = ⎜ OUT − 1⎟ R1 = 875.5k; use 887k
⎝ 0.8
⎠
(3)
Figure 6 shows the complete circuit along with its
efficiency curve.
Substituting VOUT = 2.5V, VIN = 4.2V, ∆IL = 100mA
and f = 2.25MHz in Equation (3) gives:
2.5V
⎛ 2.5V ⎞
⎜ 1−
⎟ = 4.5µH
2.25MHz(100mA) ⎝ 4.2V ⎠
VIN
2.7V
TO 4.2V
4
†
CIN
2.2µF
CER
VIN
SW
3
4.7µH*
VOUT
2.5V
10pF
LTC3410B
1
COUT†
2.2µF
CER
RUN
VFB
6
887k
GND
2, 5
412k
†
TAIYO YUDEN JMK212BJ225
*MURATA LQH32CN4R7M23
3410 F07a
Figure 6a
100
90
VOUT
100mV/DIV
AC COUPLED
80
EFFICIENCY (%)
L=
70
60
IL
200mA/DIV
50
40
ILOAD
200mA/DIV
30
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
1
10
100
OUTPUT CURRENT (mA)
1000
4µs/DIV
VIN = 3.6V
VOUT = 2.5V
ILOAD = 100mA TO 300mA
3410 F07c
3410 F07b
Figure 6b
Figure 6c
3410bfa
13
LTC3410B
U
TYPICAL APPLICATIO
VIN
2.7V
TO 4.2V
4
†
CIN
2.2µF
VIN
SW
3
VOUT
1.5V
10pF
LTC3410B
1
4.7µH*
COUT†
2.2µF
RUN
VFB
6
GND
2, 5
3410 TA02
402k
464k
†
TAIYO YUDEN JMK212BJ225
*MURATA LQH32CN4R7M23
100
90
EFFICIENCY (%)
80
70
60
50
40
30
20
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
10
0
1
10
100
OUTPUT CURRENT (mA)
1000
3410 TA03
VOUT
100mV/DIV
AC COUPLED
IL
200mA/DIV
ILOAD
200mA/DIV
4µs/DIV
VIN = 3.6V
VOUT = 1.5V
ILOAD = 100mA TO 250mA
3410 TA04
3410bfa
14
LTC3410B
U
PACKAGE DESCRIPTIO
SC6 Package
6-Lead Plastic SC70
(Reference LTC DWG # 05-08-1638)
0.47
MAX
0.65
REF
1.80 – 2.20
(NOTE 4)
1.00 REF
INDEX AREA
(NOTE 6)
1.80 – 2.40 1.15 – 1.35
(NOTE 4)
2.8 BSC 1.8 REF
PIN 1
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.10 – 0.40
0.65 BSC
0.15 – 0.30
6 PLCS (NOTE 3)
0.80 – 1.00
0.00 – 0.10
REF
1.00 MAX
GAUGE PLANE
0.15 BSC
0.26 – 0.46
0.10 – 0.18
(NOTE 3)
SC6 SC70 1205 REV B
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. DETAILS OF THE PIN 1 IDENTIFIER ARE OPTIONAL,
BUT MUST BE LOCATED WITHIN THE INDEX AREA
7. EIAJ PACKAGE REFERENCE IS EIAJ SC-70
8. JEDEC PACKAGE REFERENCE IS MO-203 VARIATION AB
3410bfa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LTC3410B
U
TYPICAL APPLICATIO
Using Low Profile Components, <1mm Height
VIN
2.7V
TO 4.2V
4
†
CIN
4.7µF
VIN
SW
3
4.7µH*
†
COUT
4.7µF
CER
LTC3410B-1.875
1
RUN
VOUT
6
VOUT
1.875V
GND
†
TAIYO YUDEN JMK212BJ475
*FDK MIPF2520D
2, 5
3410B TA06a
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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LTC3403
600mA (IOUT), 1.5MHz, Synchronous Step-Down
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600mA (IOUT), 1.4MHz, Synchronous Step-Down
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LTC3405/LTC3405A
300mA (IOUT), 1.5MHz, Synchronous Step-Down
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600mA (IOUT), 1.5MHz, Synchronous Step-Down
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96% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 20µA,
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LTC3407/LTC3407-2
Dual 600mA/800mA (IOUT), 1.5MHz/2.25MHz,
Synchronous Step-Down DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA,
ISD = <1µA, DFN, MS10E Packages
LTC3409
600mA (IOUT), 1.5MHz/2.25MHz, Synchronous
Step-Down DC/DC Converter
95% Efficiency, VIN = 1.6V to 5.5V, VOUT(MIN) = 0.613V, IQ = 65µA,
DD8 Package
LTC3410
300mA (IOUT), 2.25MHz, Synchronous Step-Down
DC/DC Converter with Burst Mode Operation
96% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 26µA,
ISD = <1µA, SC7O Package
LTC3411
1.25A (IOUT), 4MHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA,
ISD = <1µA, MS Package
LTC3412/LTC3412A
2.5A/3A (IOUT), 4MHz, Synchronous Step-Down
DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 0.8V, IQ = 60µA,
ISD = <1µA, TSSOP-16E Package
LTC3440
600mA (IOUT), 2MHz, Synchronous Buck-Boost
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95% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 2.5V to 5V,
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LTC3548
Dual 400mA/800mA (IOUT), 2.25MHz,
Synchronous Step-Down DC/DC Converter
95% Efficiency, VIN = 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA,
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3410bfa
16
Linear Technology Corporation
LT 0706 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
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