AND8154/D NCP1230 90 Watt, Universal Input Adapter Power Supply Prepared by: Terry Allinder [email protected] ON Semiconductor http://onsemi.com APPLICATION NOTE General Description The NCP1230 implements a standard current mode control architecture. It’s an ideal candidate for applications where a low parts count is a key parameter, particularly in low cost adapter power supplies. The NCP1230 combines a low standby power mode with an event management scheme that will disable a PFC circuit during Standby, thus reducing the no load power consumption. The 90 W Demo Board demonstrates the wide range of features found on the NCP1230 controller. The NCP1230 has a PFC_Vcc output pin which provides Vcc power for a PFC controller, or other circuitry. The PFC_Vcc pin is enabled when the output of the power supply is up and in regulation. In the event that there is an output fault, the PFC_Vcc pin is turned off, disabling the PFC controller, reducing the stress on the PFC semiconductors. In addition to excellent no load power consumption, the NCP1230 provides an internal latching function that can be used for over voltage protection by pulling the CS pin above 3.0 V. self−contained and includes a bias supply that operates off of the Auxiliary winding of the transformer. Table 1. Demo Board Specifications Current−Mode Control Lossless Startup Circuit Operation Over the Universal Input Range Direct Connection to PFC Controller Low Standby Overvoltage Protection June, 2004 − Rev. 0 Min Max Input Vac 85 265 Frequency Hz 47 63 Vo Vdc 18.6 19.38 Io Adc − 4.74 Output Power W − 90 efficiency 80 − Standby Power Vin 230 Vac mW − 150 Pin Short Circuit Load Vin 230 Vac mW Pin with 0.5 W Load Vin 230 Vac mW 100 − 0.8 2 · 2 · Pin max Vac Ipk Ipk 2 · 2 · 116 3.86 A 85 The MC33260 is a Critical Conduction Mode controller; as a result the switching frequency is a function of the boost inductor and the timing capacitor. In this application the minimum operating frequency is 30 kHz. Design Specification This Demo Board is configured as a two stage adapter power supply. The first stage operates off of the universal input, 85−265 Vac, 50−60 Hz, using the MC33260 Critical Conduction Mode controller, in the Boost Follower mode. The output voltage from the Boost Follower (when Vin is 85 Vac) is 200 V and as the input line increases to 230 Vac the output of the Boost Follower will ramp up to 400 Vdc. The second stage of the power supply features the NCP1230 driving a flyback power stage. The output of the second stage is 19 Vdc capable of 90 W of output power. It is fully Semiconductor Components Industries, LLC, 2004 Symbol PFC The MC33260 is configured as a Boost Follower operating from the universal input line. The PFC section was designed to provide approximately 116 W of power. Features • • • • • • Requirement Vo 2 2 · Tp Lp Vo · Vac · Ipk 2 · 33.33 Lp Vac · (Vac)2 200 2 85 · (85)2 200 · 85 · 3.86 414 H The value used is 400 H. 1 Publication Order Number: AND8154/D AND8154/D Where: · 112.5 432 H Lp 22· 0.566 0.4 265 1 Tp 1 33.33 sec 30 Freq min In this application the primary inductance used is 220 H. This takes into consideration the transformer tolerances, and to minimize the transformer size. Once the primary inductance has been calculated, the next step is to determine the peak primary current. Vomin = 200 Vdc (@ 85 Vac input) Vac = 85 Vac The oscillator timing capacitor is calculated by the following formula: CT Pin 1 · Ipk2 · Lp · f 2 4 Vo2 Kosc Lp Pin Cint Ro2 Vpk2 Ipk 2 CT 4 · 200 · 6400 · 400 · 116 15 809 pF 22 · 1202 Where: Kosc = 6400 Ro = 2.0 M (feedback resistor) The CT value used is 820 pF Refer to the ON Semiconductor website for Application Note AND8123/D for additional MC33260 application information, and the Excel based development tool DDTMC33260/D. Ipk 2220· 112.5 3.97 Apk · 65 The following calculations are used to verify that the current will be Discontinuous under all operating conditions. Tp Ton Toff 1 freq Ton Startup Circuit Description The High Voltage pin (pin 8) of the NCP1230 controller is connected directly to the high voltage DC bus. When the input power is turned on, an internal current source is turned on (typically 3.0 mA) charging up an external capacitor on the Vcc pin. When the Vcc capacitor is above VCCoff, the current source is turned off, and the controller delivers output drive pulses to an external MOSFET, Q1. The MOSFET, Q1, drives the primary of the transformer T1. The transformer has two additional windings, the auxiliary winding which provides power to the controller after the power supply is running, and the secondary winding which provided the 19 Vdc output power. Toff Tp Lp · Ipk Vin Ls · Iopk Vo Vf LpVin· Ipk LsVo·IopkVf Where: Ls Lp n2 n is the transformer turns ratio 6.77 Tp 220 · 3.97 4.8 · 27.22 10 s 200 19 0.7 With a primary inductance value of 220 H, Ton + Toff is less than the controller switching period. An Excel spreadsheet was designed using the above equation to help calculate the correct primary inductance value; visit the ON Semiconductor website for a copy of the spreadsheet. One method for calculating the transformer turns ratio is to minimize the voltage stress of the MOSFET (VDS) due to the reflected output voltage. Transformer The transformer primary inductance was selected so the current would be discontinuous under all operating conditions. As a result the total switching period, Ton + Toff, must be less than or equal to 1/frequency. The following assumptions were used in the design process: Dmax = 0.4 Duty Cycle Vdc bus = 200 Vdc input with Vin 85 Vac Efficiency = 0.80 Freq = 65 kHz Vo = 19 V Vf = 0.7 Po = 90 W VDSmax Vinmax n · (Vo Vf) Vspike In this application an 800 V MOSFET was selected. The goal, for safety purposes, is to limit VDSmax at high line (including the Vspike) to 700 V. To limit the power dissipation in the snubber clamp (refer to the section in the Applications Note titled “Snubber”.) Vspike is clamped at 167 V. 90 Pin Po 0.8 112.5 W n VDSmax Vinmax Vspike Vo Vf n 700 400 167 6.77 19.7 Iavg Pin 112.5 0.566 200 Vin Lp PinLp ·· f2 2 · Pin Iavg 2D·max · Freq http://onsemi.com 2 AND8154/D In the 90 W Adapter design four 2200 F (8800 F total) capacitors (C2, C3, C14, and C15) were required in parallel to handle the ripple current. A small LC filter has been added to the output of the power supply to help reduce the output ripple. The cut−off frequency for the filter is: The NCP1230 requires that the controller Vcc be supplied through an auxiliary winding on the transformer. The nominal supply voltage for the controller is 13 Vdc. naux Vaux(1−D max) Vin · D max 13.7(1−0.4) naux 0.128 200(0.4) fp The supply voltage to the controller may be higher than the calculated value because of the transformer leakage inductance. The leakage inductance spike on the auxiliary winding is averaged by the rectifier D2 and capacitor C5. Because of this, an 18 V Zener diode (D18 refer to the Demo Board Schematic Figure 8) is connected from the Vcc pin to ground. To limit the current into the Zener diode a 200 resistor is placed between C5 and the Vcc pin (R28). ON Semiconductor recommends that the Vcc capacitor be at least 47 F to be sure that the Vcc supply voltage does not drop below Vccmin (7.6 V typical) during standby power mode and unusual fault conditions. The transformer primary rms current is: Irms Ipk L1 = 2.2 H C8 = 47 F Output Rectifying Diode The rectifying diode was selected based upon on the peak inverse voltage and the diodes average forward current. The peak inverse voltage across the secondary of the transformer is: PIV Vin n Vo PIV 400 19 78 Vpk 6.77 The average current through the diode is: Don 3.97 0.4 1.45 Arms 3 3 Iavg Po 90 4.74 A 19 Vo An MBR20100CT Schottky diode was selected; it is rated for a VRRM of 100 V, with an average forward current of 10 A. The transformer secondary rms current is: 1−D 3 3.97 · 6.77 0.6 12.02 Arms 3 Irms_sec Ipk_prim · n Power Switch A MOSFET was selected as the power switching element. Several factors were used in selecting the MOSFET; current, voltage stress (VDS), and RDS(on). The rms current through the primary of the transformer is the same as the current in the MOSFET, which is 1.45 Arms. The MOSFET selected is manufactured by Infineon, part number SPP11N80C3. It is rated for 800 VDS and 11 Arms, with an RDS(on) of 0.45. The transformer for the Demo Board was manufactured by Cooper Electronics Technologies (www.cooperET.com) part number CTX22−16134. The designer should take precautions that under startup conditions, the transformer will not saturate at the low input ac line (85 Vac) and full load conditions. The above calculation assumed that the adapter was running and the PFC front end was enabled. Snubber The maximum voltage across the MOSFET is: Output Filter One of the disadvantages of a Flyback converter operating in the Discontinuous mode is there is a large ripple current in the output capacitor(s). As a result you may be required to use multiple capacitors in parallel to handle the ripple current. Vpk Vin max (Vo Vf)n Vpk 400 (19 0.7) 6.77 534 V This calculation neglects the voltage spike when the MOSFET turns off due to the transformer leakage inductance. The spike, due to the leakage inductance, must be clamped to a level below the MOSFETs’ maximum VDS. To clamp the voltage spike a resistive, capacitive, diode clamp network was used to prevent the drain voltage from rising above Vin + (Vo + Vf) n + Vclamp. The desired clamp voltage is 700 V; this provides a safety margin of 100 V. The first step is to calculate the snubber resistor. I_cap_ripple Iorms2 Io2 I_cap_ripple 12.022 4.742 11.04 A Ton 1 0.4 1 0.4 6.15 sec 65000 frequency Co Iorms (T−Ton) Vripple Rclamp Where Vripple = 50 mV. Co 1 1 15.6 kHz 2 2.2 · 47 2 LC 12.02 (15.38−9.23) 1, 478 F 0.05 Rclamp http://onsemi.com 3 2 · Vclamp · (Vclamp Vo · n) le · Ipk2 · Freq 2 · 700 · (700 (19.7 · 6.77)) 110 k 7 · 3.972 · 65 AND8154/D Rs 1 V 1 0.23 4.43 Ipk Where: Vo = the output voltage Vf = the forward voltage drop across the output diode n is the transformer turns ratio 6.77 Ie is the transformer turns ratio of 7 H The power dissipation in the clamp resistor is: PRclamp 0.5 · Ipk2 · le · Freq · PRclamp 0.5 · 3.972 · 7 · 65 · 0.2 was used. To reduce the power dissipation in the sense resistor, two 0.4 resistors were used in parallel. VclampVclamp (Vo · n) Overvoltage Protection The NCP1230 has a fast comparator which only monitors the current sense pin during the power switch off time. If the voltage on the current sense pin rises above 3.0 V (typical), the NCP1230 will immediately stop the output drive pulses and latch−off the controller. The NCP1230 will stay in the Latch−Off mode until Vcc has dropped below 4.0 V. This feature allows the user to implement several protection functions, for example, Overvoltage or Overtemperature Protection. The Auxiliary winding of the Flyback transformer (T5) can be used for overvoltage protection because the voltage on the Auxiliary winding is proportional to the output voltage. To implement Overvoltage Protection (OVP), a PNP transistor is used to bias up the current sense pin during the NCP1230 controller off time (refer to Figure 2). The base of the PNP transistor is driven by the NCP1230 drive output (pin 5), if the Auxiliary winding voltage increases above the Zener diode (D1) breakdown voltage, 13 V, current will flow through Q3 biasing up the voltage on the current sense pin. Using typical component values, if the voltage on the Auxiliary winding reaches 16.5 V (3.5 V above the nominal voltage) the NCP1230 will latch−off through the CS input (pin 3). 700 (700 19.7 · 6.77) 4.4 W The snubber capacitor can be calculated from the following equation. See Application Note AN1679/D for details of how the snubber equations were derived. C6 C6 Vclamp Vripple · Freq · Rclamp 700 0.005 F 20 · 65 · 110 After the initial snubber was calculated, the snubber values were tuned in the circuit to minimize ringing, and minimize the power dissipation. As a result the final circuit values are; Rclamp uses three 100 k (33 k equivalent), 2.0 W resistors used in parallel, and C6 is 0.01 F, 1000 V. Refer to Figure 1 for a scope waveform of the Drain to source voltage at full load and high line. OVPthreshold Vz(D1) VceQ3 CSlatchoff 13 V 0.5 V 3.0 V 16.5 V A 13 V Zener diode was selected to have the controller Latch−Off prior to having Vcc reach its maximum allowable voltage level, 18 V. Vaux 13 V NCP1230 MMBT2907A/SOT Figure 1. Current Sense Resistor Selection The input to the current sense amplifier is clamped to 1.0 V (typical). The current sense resistor should be calculated at 125% of the full rated load to be sure that under all operating conditions the power supply will be able to deliver the full rated power. 1 2 3 4 10 k GTS HV FB CS VCC GND DRV 8 6 5 1k Rsense 100 pF Po 90 · 1.25 112.5 W Pin Po 112.5 140.63 W 0.80 eff Ipk Figure 2. Overvoltage Protection Circuit · 140.63 4.43 Apk 2220 · 65 http://onsemi.com 4 AND8154/D Overtemperature Protection To implement Overtemperature Protection (OTP) shutdown, the Zener diode can be replaced by an NTC (refer to Figure 3), or an NTC can be placed in parallel with the Zener diode to have OVP and OTP protection. When an overtemperature condition occurs, the resistance of the NTC will decrease, allowing current to flow through the PNP transistor biasing up the Current Sense pin. Where: I is the peak primary current Lp is the transformer primary inductance F is the switching frequency of the controller Vo2 1 · Ipk2 · Lp · f 2 Ro Vo i Vaux Ro ·2f · Lp · n · d i Ip · Rs Vc 3 NTC MMBT2907A/SOT Q3 R26 10 k Where: Ip is the peak primary current Rs is the current sense resistor Vc is the control voltage 3, the feedback input voltage is divided down by a factor of three Combining equations the open loop gain is: NCP1230 1 2 3 4 GTS HV FB CS VCC GND DRV 8 6 5 1k Rsense C24 100 pF Vo i Ro ·2Lp · f · n · d i Ipk Rs Vc 3 Figure 3. Overtemperature Protection Circuit Vc 3 Rs Ipk Slope Compensation A Flyback converter operating in continuous conduction mode with a duty cycle greater than 50% requires slope compensation. In this application the power supply will always be operating in the discontinuous mode, so no slope compensation is required. The resistor R21 and capacitor C24 form a low pass filter suppressing the leading edge of the current signal. Typically, the leading edge of the current will have a large spike due to the transformer leakage inductance. If the spike is not filtered, it can prematurely turn off the MOSFET. The NCP1230 does have a leading edge blanking circuit, but it is a good design practice to add an external filter. The time constant of the filter must be significantly higher than the highest expected operating frequency, but low enough to filter the spike. Vo Vc Ro ·2Lp · f · n · d · Ipk · Rs · 3 With current mode control, there is pole associated with the output capacitor(s) and the load resistors. In this application there are four 2200 F capacitors in parallel: fp 1 1 9.3 Hz · 8800 · 3.9 CoRo The secondary filter made up of L1 and C8 does not affect the control loop because we are sensing the output voltage before the LC network. In addition to the pole, there is a zero associated with the output capacitor(s) and the capacitors esr. The esr of each capacitors is 0.022 (from the data sheet). fz Output Control Feedback theory states that for the control loop to be stable there must be at least 45° of phase margin when the loop gain crosses cross zero dB. The following equations derive the Flyback converter transfer function while operating in the discontinuous continuous mode. 1 1 2Co · esr 6.28 · 8800 · 0.022 4 3.3 kHz A small 0.47 nF capacitor (C25) is connected from the feedback pin to ground to reduce the switching noise on the feedback pin. Care must be taken not to have too large a capacitor, or a low frequency pole may be created in the feedback loop. Output Voltage Regulation The output voltage regulation is achieved by using a TL431 on the secondary side of the transformer. The output voltage is sensed and divided down to the reference level of the TL431 (2.5 V typical) by the resistive divider network consisting of R4 and R10. 2 Po Vo Ro Where: Po is the maximum output power Vo is the output voltage Ro is the output resistance P 1 · Ipk2 · Lp · f 2 http://onsemi.com 5 AND8154/D Loop Gain Plot The TL431 requires a minimum of 1.0 mA of current for regulation: Ropto(R22) 60 Vo Vfopto 19 1 18 k 1 mA 1 mA 40 20 GAIN IN dB In this application R22 was changed to 1.0 k to minimize the stand by power consumption. When the power supply is operating at no load, there may not be sufficient current through the optocoupler LED, so a resistor (R7) is placed in parallel. A 4.7 k resistor was selected. The optocoupler gain is: 0 −20 −40 −60 −80 Vfb Rfb · CTR 20 · 1.0 20 1 Ropto Vc −100 10 100 dBgain 20log20 26 dB Loop Phase Margin 180 140 100 Standby Power To minimize the standby power consumption, the output voltage sense resistor divider network was select to consume less than 10 mW. 60 PHASE 100000 Figure 4. Excel Spreadsheet Loop Gain CTR is the current transfer ratio of the opto and is nominally 1.0, but over time the CTR will degrade so analysis of the circuit with the CTR = 0.5 is recommended. Rfb is the internal pull−up resistor of the NCP1230 and it is a nominal 20 k. 1000 10000 FREQUENCY IN Hz R10 7.4 Vref Vo 19 2.5 V 7.4 50 R10 R4 20 −20 −60 −100 The Standby power consumption is: −140 2 2 P Vo 19 6.3 mW 57400 Rtotal −180 10 Standby power calculation: 100 P_R22 I2 * R22 12 ma · 2 k 2 mW 1000 FREQUENCY 10000 100000 Figure 5. Excel Spreadsheet Phase Margin P_TL431 (Vo V_R22 Vopto) · 1 ma 17 V · 1 ma 17 mW 180 Control Loop Two methods were used to verify that the Demo Board loop was stable, the results are shown below. The first method was to use an Excel Spreadsheet (using the previously derived equations) which can be down loaded from the ON Semiconductor website (www://onsemi.com). The results from the Excel Spreadsheet are shown below. At full load and 200 Vdc (200 Vdc is the minimum voltage being supplied from the PFC) the loop gain crosses zero dB at approximately 1.2 kHz with approximately 100° of phase margin. The second method was to model the NCP1230 Demo Board in PSPICE. The result can be seen in Figure 6. Because parasitic elements can be added to the PSPICE model, it was more accurate at high frequencies. The results from the PSPICE model (at low frequencies) shows similar results, the loop gain crosses zero dB at approximately 1.2 kHz with about 90° of phase margin. 100 0 −100 10 Hz DB(V(FB)) 100 Hz 1.0 kHz 10 kHz P(V(FB)) FREQUENCY Figure 6. SPICE Phase/Gain http://onsemi.com 6 100 kHz AND8154/D NCP1230 AV U7 + − Vin 200 Vdc OUT FB GND CTRL IN NCP1230 averaged XFMR1 U6 R7 0.16 D4 MUR810 out1 L1 2.2 H FS = 65 k L = 0.22 mH RI = 0.20 LoL 1k RATIO=0.1477 R3 0.022 CoL 1k ACMAG=1 R5 0.022 R8 0.022 R6 0.022 C4 C1 C5 C6 2200 F 2200F 2200F 2200 F Vstim FB Rload 3.9 C7 47 F R11 1k R2 49.9 k U10 R1 4.7 k C2 470 nF MOC8101 C3 0.47 nF U9 TL431 R4 7.4 k Figure 7. AC Frequency Response SPICE Model Demo Board Test Procedure Connect an ac power source to the J4 connector. Connect the dc load to the J2 connector. Place a Digital Voltage Meter (DVM) directly across the J2 output terminals. Set the ac power source to 115 Vac, and turn on the ac source. The NCP1230 controller will turn−on and supply 19 Vdc to the load (refer to Table 1 for load regulation). Vary the load from 0 to 4.73 Adc and monitor the output voltage. Adjust the ac power source from 85−265 Vac and monitor the output voltage. Set the ac power source to 230 Vac, and disconnect the dc load and monitor the standby power (refer to Table 4 for the standby power limits). http://onsemi.com 7 D12 400 H MUR460 +VDC + F1 1 D8 D9 1N5406 1N5406 L2 2 1 2 100 H J4 1 2 C11 0.1 F C27 0.22 F L3 1 1 2 L5 C23 150 F C17 0.1 F R19 1M 3 4 4.5 mH C22 0.47 F R20 1M 2 PFC_Vcc 100 H D11 D10 1N5406 1N5406 U1 MC33260 R17 8.06k R16 1.3 R5 1.3 1 OSC VC 8 2 CS FB 7 3 SYNC VCC 6 4 GND DRV 5 R6 1.3 R27 SPP07N60C3 Q2 4.7 C18 680 pF D17 18 V C19 0.1 F C12 0.68 F C13 470 pF GND AND8154/D 8 http://onsemi.com Figure 8. NCP1230 Demo Board Schematic − PFC section L4 +VDC R24 0 R2 C6 0.01 F R29 100k R18 100k 100k T5 2 D4 220 H 3 MUR160 R25 200 D2 R28 4 7 L1 2.2 H R4 49.9k D19 8 3 6 4 R26 10k CS VCC GND DRV D16 BAL19LT1 R13 20 5 C10 2.2nF Q1 SPP11N80C3 NCP1230 R21 1k R1 R3 C24 C25 1.0 nF + 100pF C7 47 F C1 0.1 F D18 18 V 0.4 0.4 4 U13 1 R7 4.7k C20 3 C3 2200 F J2 R10 7.4k C8 47F 1 2 AND8154/D MMBT2907A/SOT Q3 1 GTS HV 2 FB C14 2200 F + Figure 9. DC−DC section 9 http://onsemi.com PFC_Vcc C15 2200 F + 1k D13 1N4006 C2 2200 F + R22 + D15 1N4006 6 + BAL19LT1 C5 100 F + 200 D1 13 V MBR20100CT 5 2 SFH615AA−X007 100 nF U12 TL431 AND8154/D Table 2. Voltage Regulation and Efficiency Vin (Vac) Pin (W) Vo (Vdc) Io (Adc) Po (W) Eff (%) 85 54.50 19.02 2.36 45 82.57 115 54.40 19.03 2.36 45 82.72 230 53.21 19.06 2.36 45 84.54 265 53.1 19.06 2.36 45 84.71 85 112.00 18.82 4.77 90 80.36 115 110.84 18.88 4.77 90 81.2 230 109.42 18.88 4.77 90 82.25 265 109.01 18.89 4.77 90 82.56 Table 3. Power Factor and Distortion Vin (Vac) Pin (W) PF THD (%) Vo (Vdc) Po (W) 85 112.00 0.996 6.5 18.82 90 115 110.84 0.996 7.7 18.88 90 230 109.42 0.972 19.01 18.88 90 265 109.01 0.965 23.0 18.89 90 Table 4. Standby Power Test Condition (Vac input) Requirement Pin (mW) Pin Measured (mW) Standby Power 230 150 120 Pin Short Circuit 230 100 100 Pin with 0.5 W Load 230 800 600 Table 5. Vendor Contact List ON Semiconductor www.onsemi.com 1−800−282−9855 TDK www.component.tdk.com 1−847−803−6100 Infineon www.infineon.com Coilcraft www.coilcraft.com Vishay www.vishay.com Coiltronics www.cooperet.com 1−888−414−2645 Bussman (Cooper Ind.) www.cooperet.com 1−888−414−2645 Panasonic www.eddieray.com/panasonic.com Weidmuller www.weidmuller.com Keystone www.keyelco.com 1−800−221−5510 HH Smith www.hhsmith.com 1−888−847−6484 Aavid Thermalloy www.aavid.com http://onsemi.com 10 AND8154/D Table 6. NCP1230 Demo Board Bill of Materials Ref Des Description Part Number Manufacturer C1 Cap. Ceramic, chip, 0.1 F, 50 V VJ0805Y104KXA Vishay C2 Cap. Aluminum Elec., 2200 F, 25 V EEUFC1E222 EKB00JG422F00 Panasonic VISHAY C3 Cap. Aluminum Elec., 2200 F, 25 V EEUFC1E222 EKB00JG422F00 Panasonic VISHAY C5 Cap. Aluminum Elec., 100 F, 35 V EKB00BA310F00 Vishay C6 Cap, Ceramic, 0.01 F, 1000 V 225261148036 Vishay C7 Cap. Aluminum Elec., 47 F, 25 V Cap. Aluminum Elec., 47 F, 35 V EEUFC1E470 EKB00AA247F00 Panasonic Vishay C8 Cap. Aluminum Elec., 47 F, 25 V Cap. Aluminum Elec., 47 F, 35 V EEUFC1E470 EKB00AA247F00 Panasonic Vishay C10 Cap, Y2 class, 2.2 nF, 250 Vac F1710−222−1000 Vishay Roederstein C11 Cap, X2 0.01 F, 300 Vac F1772−410−3000 Vishay Roederstein C12 Cap. Ceramic, chip, 0.068 F, 50 V VJ0805Y683MXA Vishay C13 Cap. Ceramic, chip, 470 pF, 50 V VJ0805471KXA Vishay C14 Cap. Aluminum Elec., 2200 F, 25 V EEUFC1E222 EKB00JG422F00 Panasonic VISHAY C15 Cap. Aluminum Elec., 2200 F, 25 V EEUFC1E222 EKB00JG422F00 Panasonic VISHAY C17 Cap, Film, 0.47 F, 300 V F1772−410−3000 Vishay Roederstein C18 Cap. Ceramic, chip, 680 pF, 50 V VJ0805Y681KXA Vishay C19 Cap. Ceramic, chip, 0.1 F, 50 V VJ0805Y104KXA Vishay C20 Cap. Ceramic, chip, 100 nF, 50 V VJ0805Y103KXA Vishay C22 Cap, X2, 0.47 F, 300 Vac F1772−447−3000 Vishay Roederstein C23 Cap. Aluminum, 150 F, 450 Vdc ECOS2WP151CA Panasonic C24 Cap. Ceramic, chip, 100 pF, 50 V VJ0805A100KXA Vishay C25 Cap. Ceramic, chip, 1.0 nF, 50 V VJ0805Y102KXA Vishay C27 Cap, X2 0.22 F, 300 Vac F1772−422−3000 Vishay Roederstein D1 Diode, Zener, 13 V, SM, 0.3 W AZ23C13 VISHAY D2 Diode, Signal, 75 V, 100 mA, SM BAL19LT1 ON Semiconductor D4 Diode, Ultra Fast, 800 V, 1.0 A MUR160 ON Semiconductor D8 Diode, Rectifier, 600 V, 3.0 A 1N5408 ON Semiconductor D9 Diode, Rectifier, 600 V, 3.0 A 1N5408 ON Semiconductor D10 Diode, Rectifier, 600 V, 3.0 A 1N5408 ON Semiconductor D11 Diode, Rectifier, 600 V, 3.0 A 1N5408 ON Semiconductor D12 Diode, Ultra−Fast, 600 V, 4.0 A MUR460 ON Semiconductor D13 Diode, Rectifier, 800 V, 1.0 A 1N4006 ON Semiconductor D15 Diode, Rectifier, 800 V, 1.0 A 1N4006 ON Semiconductor D16 Diode, Signal, 75 V, 100 mA, SM BAL19LT1 ON Semiconductor D17 Diode, Zener, 18 V, SM, 0.3 W AZ23C18 VISHAY D18 Diode, Zener, 18 V, SM, 0.3 W AZ23C18 VISHAY D19 Diode, Schottky, 100 V, 10 A MBR20100CT ON Semiconductor F1 Fuse, 2.0 A, 250 Vac 1025TD2A Bussman http://onsemi.com 11 AND8154/D Table 6. NCP1230 Demo Board Bill of Materials (continued) J2 Connector 171602 Weidmuller J4 Connector 171602 Weidmuller L1 Inductor, 2.2 H, 7.5 A DO33316P−222 Coilcraft L2 Inductor, 100 H, 2.5 A TSL1315−101K2R5 TDK L3 Inductor, 100 H, 2.5 A TSL1315−101K2R5 TDK L4 PFC Inductor, 400 H, 7.1 A CTX22−16708 Cooper Electronics L5 Common Mode Inductor, 4.5 mH E3506−A Coilcraft Q1 MOSFET, 11 A, 800 V, 0.8 SPP11N80C3 Infineon Q2 MOSFET, 7.0 A, 600 V, 0.8 SPP07N60C3 Infineon Q3 Bipolar transistor, 50 V MMBT2907A ON Semiconductor R1 Resistor, SM, 0.4 , 1% WSL−2512.4 1% VISHAY R2 Resistor, 100 k, 3 W, 5% CFP−3104JT−00 VISHAY R3 Resistor, SM, 0.4 , 1% WSL−2512.4 1% VISHAY R4 Resistor, SM, 49.9 k, 1% CRCW12064992F VISHAY R5 Resistor, SM, 1.3 , 1% CRCW25121R30F VISHAY R6 Resistor, SM, 1.3 , 1% CRCW25121R30F VISHAY R7 Resistor, SM, 4.7 k, 5% CRCW0805472JNTA VISHAY R10 Resistor, SM, 7.42 k, 1% CRCW12067422F VISHAY R13 Resistor, SM, 20 , 5% CRCW805020RJNTA VISHAY R16 Resistor, SM, 1.3 , 1% CRCW25121R30F VISHAY R17 Resistor, SM, 8.06 k, 1% CRCW8058K06FKTA VISHAY R18 Resistor, 100 k, 3 W, 5% CFP−3104JT−00 VISHAY R19 Resistor, 1.0 M, 1% CMF−55−1004FKRE VISHAY R20 Resistor, 1.0 M, 1% CMF−55−1004FKRE VISHAY R21 Resistor, SM, 1.0 k, 1% CRCW8051K00FKTA VISHAY R22 Resistor, SM, 1.0 k, 1% CRCW8051K00FKTA VISHAY R24 Jumper, 22 AWG R25 Resistor, 200 , 1/4 W, 5% CRCW805200RJNTA VISHAY R26 Resistor, 10 k, 1/4 W, 5% CRCW80510K0JNTA VISHAY R27 Resistor, SM, 4.7 , 5% CRCW8054R7JNTA VISHAY R28 Resistor, 200 , 1/4 W, 5% CRCW805200RJNTA VISHAY R29 Resistor, 100 k, 3 W, 5% CFP−3104JT−00 VISHAY U1 Flyback Controller NCP1230D65 ON Semiconductor U2 PFC Controller MC33260D ON Semiconductor U12 2.5 V Programmable Reference TL431ACD ON Semiconductor U4 Opto Coupler SFH615AA−X007 Infineon T1 Flyback Transformer CTX22−16134 Cooper Electronics H1 Shoulder Washer 3049K−ND Digi−key H2 Insulator 4672 Keystone H3 Heatsink, TO−220 590302B03600 Aavid H4 Heatsink, TO−220 590302B03600 Aavid H5 Heatsink, TO−220 590302B03600 Aavid http://onsemi.com 12 AND8154/D Notes http://onsemi.com 13 AND8154/D ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). 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