AN-732 APPLICATION NOTE One Technology Way • P.O. Box 9106 • Norwood, MA 02062-9106 • Tel: 781/329-4700 • Fax: 781/326-8703 • www.analog.com Universal Precision Op Amp Evaluation Board in SOIC Package by Giampaolo Marino, Soufiane Bendaoud, and Steve Ranta INTRODUCTION The EVAL-PRAOPAMP-1R is an evaluation board which accommodates single op amps in SOIC packages. It is meant to provide the user with multiple choices and extensive flexibility for different applications circuits and configurations. This board is not intended to be used with high frequency components or high speed amplifiers. However, it provides the user with many combinations for various circuit types including active filters, differential amplifiers, and external frequency compensation circuits. A few examples of application circuits are given in this application note. LOW-PASS FILTER Figure 1 is a typical representation of a first-order lowpass filter. This circuit has a 6 dB per octave roll-off after a close-loop –3 dB point defined by f C. Gain below this frequency is defined as the magnitude of R7 to R2. The circuit might be considered as an ac integrator for frequencies well above f C ; however, the time domain response is that of a single RC, rather than an integral. f C = 1/(2 R7 C7); –3 dB frequency fL = 1/(2 R2 C7); unity gain frequency Acl = –(R7/R2); close loop gain R6 should be chosen equal to the parallel combination between R7 and R2 in order to minimize errors due to bias currents. C7 R7 R7 VIN R2 VOUT VIN1 R6 VIN2 R2 R4 VOUT R6 60 Figure 2. Difference Amplifier fC GAIN (dB) 40 20 fL 0 R7/R2 = 100 –20 f 10f 100f 1000f RELATIVE FREQUENCY Figure 1. Simple Low-Pass Filter 10000f DIFFERENCE AMPLIFIER AND PERFORMANCE OPTIMIZATION Figure 2 shows an op amp configured as a difference amplifier. The difference amplifier is the complement of the summing amplifier, and allows the subtraction of two voltages or the cancellation of a signal common to both inputs. The circuit shown in Figure 2 is useful as a computational amplifier in making a differential to single-ended conversion or in rejecting a commonmode signal. The output voltage VOUT is comprised of two separate components: 1. A component VOUT1 due to VIN1 acting alone (VIN2 short-circuited to ground.) 2. A component VOUT 2 due to VIN2 acting alone (VIN1 short-circuited to ground.) REV. A AN-732 CURRENT-TO-VOLTAGE CONVERTER Current may be measured in two ways with an operational amplifier. Current can be converted to a voltage with a resistor and then amplified or injected directly into a summing node. The algebraic sum of these two components should be equal to VOUT. By applying the principles expressed in the output voltage VOUT components, and by letting R4 = R2 and R7 = R6, then: VOUT1 = VIN1 R7/R2 R7 VOUT2 = –VIN2 R7/R2 VOUT = VOUT1 + VOUT2 = ( VIN1 – VIN2) R7/R1 IIN1 Difference amplifiers are commonly used in high accuracy circuits to improve the common-mode rejection ratio, typically known as CMRR. R6 VOUT = IIN1 ⴛ R7 For this type of application, CMRR depends upon how tightly matched resistors are used; poorly matched resistors result in a low value of CMRR. Figure 3. Current-to-Voltage Converter Figure 3 is a typical representation of a current-to-voltage transducer. The input current is fed directly into the summing node and the amplifier output voltage changes to exactly the same current from the summing node through R7. The scale factor of this circuit is R7 volts per amps. The only conversion error in this circuit is IBIAS, which is summed algebraically with IIN1. To see how this works, consider a hypothetical source of error for resistor R7 (1 – error). Using the superposition principle and letting R4 = R2 and R7 = R6, the output voltage would be as follows: VOUT VOUT R 7 R 2 + 2R 7 error 1 − R 2 + R 7 × 2 R 2 = 7 R VD + × error R2 + R7 R7 VDD = VIN 2 − VIN 1 C9 V– R4 V+ From this equation, ACM and A DM can be defined as follows: VOUT R9 ACM = R7/(R7 – R2) error Figure 4. Bistable Multivibrator ADM = R7/R2 {1 – [(R2+2R7/R2+R7) error/2]} These equations demonstrate that when there is not an error in the resistor values, the ACM = 0 and the amplifier responds only to the differential voltage being applied to its inputs; under these conditions, the CMRR of the circuit becomes highly dependent on the CMRR of the amplifier selected for this job. L+ As mentioned above, errors introduced by resistor mismatch can be a big drawback of discrete differential amplifiers, but there are different ways to optimize this circuit configuration: BL+ = VTH L– 1. The differential gain is directly related to the ratio R7/ R2; therefore, one way to optimize the performance of this circuit is to place the amplifier in a high gain configuration. When larger values for resistors R7 and R6 and smaller values for resistors R2 and R4 are selected, the higher the gain, the higher the CMRR. For example, when R7 = R6 = 10 k, and R2 = R4 = 1 k, and error = 0.1%, CMRR improves to better than 80 dB. For high gain configuration, select amplifiers with very low IBIAS and very high gain (such as the AD8551, AD8571, AD8603, and AD8605) to reduce errors. BL– = VTL Figure 5. Output Response GENERATION OF SQUARE WAVEFORMS USING A BISTABLE MULTIVIBRATOR A square waveform can be simply generated by arranging the amplifier for a bistable multivibrator to switch states periodically as Figure 5 shows. Once the output of the amplifier reaches one of two possible levels, such as L+, capacitor C9 charges toward this level through resistor R7. The voltage across C9, which is applied to the negative input terminal of the amplifier denoted as V–, then rises exponentially toward L+ with a time constant = C9R7. Meanwhile, the voltage 2. Select resistors that have much tighter tolerance and accuracy. The more closely they are matched, the better the CMRR. For example, if a CMRR of 90 dB is needed, then match resistors to approximately 0.02%. –2– REV. A AN-732 at the positive input terminal of the amplifier, denoted as V+ = BL+. This continues until the capacitor voltage reaches the positive threshold V TH , at which point the bistable multivibrator switches to the other stable state in which VO = L– and V+ = BL–. This is shown in Figure 5. VOLTAGE (200mV/DIV) RL = 10k⍀ CL = 2nF The capacitor then begins to discharge, and its voltage, V–, decreases exponentially toward L–. This continues until V– reaches the negative threshold V TL, at which time the bistable multivibrator switches to the positive output state, and the cycle repeats itself. It is important to note that the frequency of the square wave being generated, f O, depends only on the external components being used. Any variation in L+ will cause V+ to vary in proportion, ensuring the same transition time and the same oscillation frequency. The maximum operating frequency is determined by the amplifier speed, which can be increased significantly by using faster devices. TIME (10s/DIV) Figure 8. Capacitive Load Drive with Resistor EXTERNAL COMPENSATION TECHNIQUES Series Resistor Compensation The use of external compensation networks may be required to optimize certain applications. Figure 6 is a typical representation of a series resistor compensation for stabilizing an op amp driving capacitive load. The stabilizing effect of the series resistor isolates the op amp output and the feedback network from the capacitive load. The required amount of series resistance depends on the part used, but values of 5 to 50 are usually sufficient to prevent local resonance. The disadvantages of this technique are a reduction in gain accuracy and extra distortion when driving nonlinear loads. The lowest operating frequency depends on the practical upper limits set by R7 and C9. Using the name convention outlined on the PRA OPAMP evaluation board, the circuit should be connected as follows: B = R4/(R4 + R9); feedback factor (noninverting input) T = 2R7 C9 ln((1 + B)/(1 – B)); period of oscillation f O = 1/T; oscillation frequency R02 VIN CL GND VOUT RL VIN Figure 6. Series Resistor Compensation RS VOUT CL RL CS Figure 9. Snubber Network VOLTAGE (200mV/DIV) VOLTAGE (200mV/DIV) RL = 10k⍀ CL = 2nF GND TIME (10s/DIV) TIME (10s/DIV) Figure 7. Capacitive Load Drive Without Resistor REV. A GND Figure 10. Capacitive Load Drive Without Snubber –3– Snubber Network Another way to stabilize an op amp driving a capacitive load is with the use of a snubber, as shown in Figure 9. This method presents the significant advantage of not reducing the output swing because there is not any isolation resistor in the signal path. Also, the use of the snubber does not degrade the gain accuracy or cause extra distortion when driving a nonlinear load. The exact RS and CS combinations can be determined experimentally. VOLTAGE (200mV/DIV) RL = 10k⍀ CL = 500pF RS = 100⍀ CS = 1nF TIME (10s/DIV) Figure 11. Capacitive Load Drive with the Snubber C2 *R1 *C8 R7 VCC C9 *R2 ALL PASSIVE COMPONENT MOUNTING LOCATIONS CAN ACCOMMODATE 0805 TO 2512 SIZED SURFACE MOUNT PACKAGES, 1/2" SPACED LEADED RESISTORS AND 0.3" SPACED LEADED CAPACITORS C7 *THESE COMPONENTS ARE BY DEFAULT SHORTED WITH A SMALL COPPER STRAP BETWEEN THE MOUNTING PADS. TO USE THE COMPONENT MOUNTING LOCATION, SIMPLY REMOVE THE STRAP WITH A CUTTING UTENSIL (DREMEL CUTTING TIP, EXACTO BLADE, ETC.) *R11 R8 C10 G3 C1 R10 V1 RT1 G1 DUT 0 2 R5 7 V+ 6 R3 C5 3 *RO *RO1 *RO2 VO V– RS 4 CS R9 V2 RT2 G2 *R4 R6 CL RL G5 C11 C6 G4 0 *R12 VEE Figure 12. EVAL-PRAOPAMP-1R Electrical Schematic Figure 13. EVAL-PRAOPAMP-1R Board Layout Patterns © 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. –4– REV. A AN04894–0–8/04(A) AN-732

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