IR3506 DATA SHEET XPHASE3TM DDR & VTT PHASE IC DESCRIPTION TM The IR3506 Phase IC combined with IR3522 xPHASE3 Control ICs implements a full featured DDR3 power solution. The IR3522 provides control functions for both the VDDR and VTT power rails and interfaces with any number of IR3506 Phase ICs each driving and monitoring a single phase to power any number of DDR3 TM DIMMs. The xPHASE3 architecture delivers a power supply that is smaller, more flexible, and easier to design while providing higher efficiency than conventional approaches. FEATURES • • • • • • • • • • • • • • • • Power State Indicator (PSI) interface provides the capability to maximize efficiency at light loads Anti-bias circuitry 7V/2A gate drivers (4A GATEL sink current) Support loss-less inductor current sensing Phase delay DFF bypassed during PSI assertion mode to improve output ripple performance Over-current protection during PSI assertion mode operation Integrated boot-strap synchronous PFET Only four IC related external components per phase 3 wire analog bus connects Control and Phase ICs (VDAC, Error Amp, IOUT) 3 wire digital bus for accurate daisy-chain phase timing control without external components Debugging function isolates phase IC from the converter Self-calibration of PWM ramp, current sense amplifier, and current share amplifier Single-wire bidirectional average current sharing Soft-stop turn-off to insure VDDR and Vtt tracking Small thermally enhanced 16L 3 x 3mm MLPQ package RoHS Compliant APPLICATION CIRCUIT 12V 15 13 CSIN+ PSI BOOST GATEL PGND GATEH VCCL 12 CIN 11 CBST L 10 VOUT+ 9 U12 COUT VOUT- 8 PHSIN CLKIN LGND 5 4 IR3506 PHASE IC DACIN 7 3 SW 6 2 RCS U11 IOUT PHSOUT 1 7 Wire Bus to Control IC CSIN- 16 EAIN U1 14 CCS CVCCL VCCL Page 1 of 21 V3.02 IR3506 ORDERING INFORMATION Part Number IR3506MTRPBF Package 16 Lead MLPQ (3 x 3 mm body) 16 Lead MLPQ (3 x 3 mm body) * IR3506MPBF Order Quantity 3000 per reel 100 piece strips * Samples only ABSOLUTE MAXIMUM RATINGS Stresses beyond those listed below may cause permanent damage to the device. These are stress ratings only and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications are not implied. o o Operating Junction Temperature…………….. 0 C to 150 C o o Storage Temperature Range………………….-65 C to 150 C MSL Rating………………………………………2 o Reflow Temperature…………………………….260 C PIN # PIN NAME VMAX VMIN ISOURCE ISINK 1 2 3 4 5 6 IOUT DACIN LGND PHSIN PHSOUT CLKIN 8V 3.3V n/a 8V 8V 8V -0.3V -0.3V n/a -0.3V -0.3V -0.3V 1mA 1mA n/a 1mA 2mA 1mA 1mA 1mA n/a 1mA 2mA 1mA 7 PGND 0.3V -0.3V n/a 8 GATEL 8V 9 VCCL 8V -0.3V DC, -5V for 100ns -0.3V 5A for 100ns, 200mA DC 5A for 100ns, 200mA DC n/a 10 BOOST 40V -0.3V 1A for 100ns, 100mA DC 5A for 100ns, 200mA DC 5A for 100ns, 200mA DC 3A for 100ns, 100mA DC 11 GATEH 40V -0.3V DC, -5V for 100ns 3A for 100ns, 100mA DC 3A for 100ns, 100mA DC 12 SW 34V -0.3V DC, -5V for 100ns -0.3V -0.3V -0.3V -0.3V 3A for 100ns, 100mA DC 1mA 1mA 1mA 1mA n/a 13 PSI 8V 14 CSIN+ 8V 15 CSIN8V 16 EAIN 8V Note: 1. Maximum GATEH – SW = 8V 2. Maximum BOOST – GATEH = 8V Page 2 of 21 1mA 1mA 1mA 1mA V3.02 IR3506 RECOMMENDED OPERATING CONDITIONS FOR RELIABLE OPERATION WITH MARGIN o 4.75V ≤ VCCL ≤ 7.5V, 0.5V ≤ V(DACIN) ≤ 1.6V, 250kHz ≤ CLKIN ≤ 9MHz, 250kHz ≤ PHSIN ≤1.5MHz, 0 C ≤ TJ ≤ o 125 C ELECTRICAL CHARACTERISTICS The electrical characteristics table list the spread of critical values that are guaranteed to be within the recommended operating conditions (unless otherwise specified). Typical values represent the median values, which are related to 25°C. CGATEH = 3.3nF, CGATEL = 6.8nF (unless otherwise specified). PARAMETER Gate Drivers GATEH Source Resistance GATEH Sink Resistance GATEL Source Resistance GATEL Sink Resistance GATEH Source Current GATEH Sink Current GATEL Source Current GATEL Sink Current GATEH Rise Time GATEH Fall Time GATEL Rise Time GATEL Fall Time GATEL low to GATEH high delay GATEH low to GATEL high delay Disable Pull-Down Resistance Clock CLKIN Threshold CLKIN Bias Current CLKIN Phase Delay PHSIN Threshold PHSOUT Propagation Delay PHSIN Pull-Down Resistance PHSOUT High Voltage PHSOUT Low Voltage Page 3 of 21 TEST CONDITION BOOST – SW = 7V. Note 1 BOOST – SW = 7V. Note 1 VCCL – PGND = 7V. Note 1 VCCL – PGND = 7V. Note 1 BOOST=7V, GATEH=2.5V, SW=0V. BOOST=7V, GATEH=2.5V, SW=0V. VCCL=7V, GATEL=2.5V, PGND=0V. VCCL=7V, GATEL=2.5V, PGND=0V. BOOST – SW = 7V, measure 1V to 4V transition time BOOST - SW = 7V, measure 4V to 1V transition time VCCL – PGND = 7V, Measure 1V to 4V transition time VCCL – PGND = 7V, Measure 4V to 1V transition time BOOST = VCCL = 7V, SW = PGND = 0V, measure time from GATEL falling to 1V to GATEH rising to 1V BOOST = VCCL = 7V, SW = PGND = 0V, measure time from GATEH falling to 1V to GATEL rising to 1V Note 1 Compare to V(VCCL) CLKIN = V(VCCL) Measure time from CLKIN<1V to GATEH>1V Compare to V(VCCL) Measure time from CLKIN > (VCCL * 50% ) o to PHSOUT > (VCCL *50%). 10pF @125 C I(PHSOUT) = -10mA, measure VCCL – PHSOUT I(PHSOUT) = 10mA MIN TYP MAX UNIT 1.0 1.0 1.0 0.4 2.0 2.0 2.0 4.0 5 2.5 2.5 2.5 1.0 10 Ω Ω Ω Ω A A A A ns 5 10 ns 10 20 ns 5 10 ns 10 20 40 ns 10 20 40 ns 30 80 130 kΩ -0.5 40 45 0.0 75 0.5 125 µA ns 35 4 50 15 55 35 % ns 30 100 170 kΩ 1 0.6 0.4 % V 1 V3.02 V IR3506 PARAMETER PWM Comparator TEST CONDITION PWM Ramp Slope V(VCCL) = 6.8 V EAIN Bias Current Minimum Pulse Width Current Sense Amplifier CSIN+/- Bias Current CSIN+/- Bias Current Mismatch Input Offset Voltage 0 ≤ EAIN ≤ 3V Note 1 Gain Unity Gain Bandwidth Slew Rate Differential Input Range Differential Input Range Common Mode Input Range o Rout at TJ = 25 C o Rout at TJ = 125 C IOUT Source Current IOUT Sink Current Share Adjust Amplifier Input Offset Voltage Gain Unity Gain Bandwidth PWM Ramp Floor Voltage Maximum PWM Ramp Floor Voltage Minimum PWM Ramp Floor Voltage PSI Comparator Rising Threshold Voltage Falling Threshold Voltage Hysteresis Resistance Floating Voltage Page 4 of 21 Note 1 CSIN+ = CSIN- = DACIN. Measure input referred offset from DACIN 0.5V ≤ V(DACIN) < 1.6V C(IOUT)=10pF. Measure at IOUT. Note 1 MIN TYP MAX UNIT 46 -5 53 -0.3 55 63 5 70 mV/ %DC µA ns -200 -50 0 0 200 50 nA nA -1 0 1 mV 30 4.8 32.5 6.8 35 8.8 V/V MHz 6 0.8V ≤ V(DACIN) ≤ 1.6V, Note 1 0.5V ≤ V(DACIN) < 0.8V, Note 1 Note 1 Note 1 2.3 3.6 0.500 0.500 3.0 4.7 1.6 1.4 50 50 VCCL2.5V 3.7 5.4 2.9 2.9 Note 1 CSIN+ = CSIN- = DACIN. Note 1 Note 1 IOUT unconnected Measured Relative to DACIN IOUT = DACIN - 200mV Measured relative to FLOOR voltage. IOUT = DACIN + 200mV Measured relative to FLOOR voltage -3 3.6 4 -116 0 4.7 8.5 0 3 6.0 17 +116 120 180 240 -220 -160 -100 520 400 50 200 800 620 550 70 500 700 650 120 850 1150 Note 1 Note 1 Note 1 -10 -5 0 V/µs mV mV V kΩ kΩ mA mA mV V/V kHz mV mV mV V3.02 mV mV mV kΩ mV IR3506 PARAMETER TEST CONDITION Diode Emulation Preset Comparator Threshold Voltage Step V(IOUT) up until GATEL drives high. Compare to V(VCCL) 75 % Regulation Comparator Threshold Voltage The ratio of V(CSIN-) / V(DACIN), above which diode emulation cannot occur. Negative Current Comparator Input Offset Voltage Note 1 Propagation Delay Time Step V(CSIN+) – V(CSIN-) from positive to negative while V(IOUT) = VCCL. Measure time to V(GATEL) < 1V. Bootstrap Diode Forward Voltage I(BOOST) = 30mA, VCCL=6.8V Debug Comparator Threshold Voltage Compare to V(VCCL) General VCCL Supply Current BOOST Supply Current 4.75V ≤ V(BOOST)-V(SW) ≤ 8V DACIN Bias Current SW Floating Voltage MIN TYP MAX UNIT -1.0 -0.8 -0.4 V 63 74 85 % -16 30 0 200 16 380 mV ns 360 520 960 mV -1.6 -1.4 -1.2 V 4.2 0.5 -1.5 0.1 10 1.5 -0.75 0.3 16.1 3 1 0.4 mA mA µA mA Note 1: Guaranteed by design, but not tested in production Page 5 of 21 V3.02 IR3506 PIN DESCRIPTION PIN# 1 PIN SYMBOL IOUT 2 DACIN 3 4 5 LGND PHSIN PHSOUT 6 7 8 9 CLKIN PGND GATEL VCCL 10 BOOST 11 12 13 14 15 GATEH SW PSI CSIN+ CSIN- 16 EAIN Page 6 of 21 PIN DESCRIPTION Output of the Current Sense Amplifier is connected to this pin through a 3kΩ resistor. Voltage on this pin is equal to V(DACIN) + 32.5 [V(CSIN+) – V(CSIN-)]. Connecting all IOUT pins together, a share bus is implemented, which provides an indication of the average current being supplied by all the phases. The Control IC, for voltage positioning and over-current protection, uses this signal. OVP mode is initiated if the voltage on this pin rises above V(VCCL)- 0.8V. Reference voltage input from the Control IC. The Current Sense signal and PWM ramp are referenced to the voltage on this pin. Ground for internal IC circuits. IC substrate is connected to this pin. Phase clock input. Phase clock output. Clock input. Return for low side driver and reference for GATEH non-overlap comparator. Low-side driver output and input to GATEH non-overlap comparator. Supply for low-side driver. Internal bootstrap synchronous PFET is connected from this pin to the BOOST pin. Supply for high-side driver. Internal bootstrap synchronous PFET is connected between this pin and the VCCL pin. High-side driver output and input to GATEL non-overlap comparator. Return for high-side driver and reference for GATEL non-overlap comparator. Logic low is an active low (i.e. low = low power state). Non-Inverting input to the current sense amplifier and input to debug comparator. Inverting input to the current sense amplifier and input to synchronous rectification disable comparator. PWM comparator input from the error amplifier output of Control IC. V3.02 IR3506 SYSTEM THEORY OF OPERATION System Description The system consists of one control IC (IR3522) and a scalable array of phase converters, each requiring one phase IC. The control IC communicates with the phase ICs using three digital buses, i.e., CLOCK, PHSIN, PHSOUT and three analog buses, i.e., DAC, EA, IOUT. The digital buses are responsible for switching frequency determination and accurate phase timing control without any external component. The analog buses are used for PWM control and current sharing among interleaved phases. The control IC incorporates all the system functions, i.e., VID, CLOCK signals, error amplifier, fault protections, current monitor, etc. The Phase IC implements the functions required by the converter of each phase, i.e., the gate drivers, PWM comparator and latch, over-voltage protection, current sensing and sharing, etc. PWM Control Method TM The PWM block diagram of the XPhase architecture is shown in Figure 1. Voltage mode control with trailing edge modulation is used. A high-gain wide-bandwidth voltage type error amplifier in the Control IC is used for the voltage control loop. The PWM ramp slope will change with the input voltage and automatically compensate for changes in the input voltage. The input voltage can change due to variations in the silver box output voltage or due to the wire and PCB-trace voltage drop related to changes in load current. GATE DRIVE VOLTAGE VIN IR3522 CONTROL IC PHSOUT CLOCK GENERATOR CLKOUT IR3506 PHASE IC CLKIN VCCH CLK Q PWM LATCH D PHSOUT PHSIN CBST VOSNS1+ VCCL SW RESET DOMINANT VOUT1 COUT - EAIN R VCCL + GND GATEL ENABLE + PGND + VID6 - REMOTE SENSE AMPLIFIER GATEH S PWM COMPARATOR PHSIN VOSNS1- - RAMP DISCHARGE CLAMP VOUT1 VREF1 LGND - EAOUT1 IOUT - CURRENT SENSE AMPLIFIER VID6 VID6 + + - 3K RCP1 VID6 VID6 + RFB12 RFB11 CCP11 FB1 IFB1 + CFB1 IROSC CDRP1 VDRP1 AMP VDRP1 CSIN+ + CCP12 CCS RCS - + SHARE ADJUST ERROR AMPLIFIER - VDAC + ERROR AMPLIFIER CSIN- DACIN RDRP1 PHSOUT IR3506 PHASE IC + - CLKIN VCCH CLK Q Output 1 Only IIN1 PWM LATCH D PHSIN GATEH CBST S PWM COMPARATOR - EAIN VCCL SW RESET DOMINANT R VCCL + GATEL ENABLE + PGND VID6 - RAMP DISCHARGE CLAMP SHARE ADJUST ERROR AMPLIFIER CURRENT SENSE AMPLIFIER + IOUT - 3K - VID6 VID6 + CSIN+ + + CCS RCS - VID6 VID6 + CSIN- DACIN Figure 1: PWM Block Diagram Page 7 of 21 V3.02 IR3506 Frequency and Phase Timing Control The oscillator is located in the Control IC and the system clock frequency is programmable from 250kHz to 9MHZ by an external resistor. The control IC system clock signal (CLKOUT) is connected to CLKIN of all the phase ICs. The phase timing of the phase ICs is controlled by the daisy chain loop, where control IC phase clock output (PHSOUT) is connected to the phase clock input (PHSIN) of the first phase IC, and PHSOUT of the first phase IC is connected to PHSIN of the second phase IC, etc. and PHSOUT of the last phase IC is connected back to PHSIN of the control IC. During power up, the control IC sends out clock signals from both CLKOUT and PHSOUT pins and detects the feedback at PHSIN pin to determine the phase number and monitor any fault in the daisy chain loop. Figure 2 shows the phase timing for a four-phase converter. The switching frequency is set by the resistor ROSC as shown in Figure 9. The clock frequency equals the number of phase times the switching frequency. Control IC CLKOUT (Phase IC CLKIN) Control IC PHSOUT (Phase IC1 PHSIN) Phase IC1 PWM Latch SET Phase IC 1 PHSOUT (Phase IC2 PHSIN) Phase IC 2 PHSOUT (Phase IC3 PHSIN) Phase IC 3 PHSOUT (Phase IC4 PHSIN) Phase IC4 PHSOUT (Control IC PHSIN) Figure 2: Four Phase Oscillator Waveforms PWM Operation The PWM comparator is located in the phase IC. Upon receiving the falling edge of a clock pulse, the PWM latch is sets and the PWM ramp voltage begins to increase. In conjunction, the low side driver is turned off and the high side driver is turned on after a non-overlap time. When the PWM ramp voltage exceeds the error amplifier’s output voltage, the PWM latch is reset. This turns off the high side driver, turns on the low side driver after the non-overlap time, and activates the ramp discharge clamp. The clamp drives the PWM ramp voltage to a level set by the share adjust amplifier until the next clock pulse. The PWM latch is reset dominant allowing all phases to go to zero duty cycle within a few tens of nanoseconds in response to a load step decrease. Phases can overlap and go to a 100% duty cycle in response to a load step increase with turn-on gated by the clock pulses. An error amplifier output voltage greater than the common mode input range of the PWM comparator results in 100% duty cycle regardless of the voltage of the PWM ramp. This arrangement guarantees the error amplifier is always in control and can demand 0 to 100% duty cycle as required. It also favors response to a load step decrease which is appropriate given the low output to input voltage ratio of most systems. The inductor current will increase much more rapidly than decrease in response to load transients. An additional advantage of this PWM modulator is that differences in ground or input voltage at the phases have no effect on operation since the PWM ramps are referenced to VDAC. Figure 3 depicts PWM operating waveforms under various conditions. Page 8 of 21 V3.02 IR3506 PHASE IC CLOCK PULSE EAIN PWMRMP VDAC GATEH GATEL STEADY-STATE OPERATION DUTY CYCLE INCREASE DUE TO LOAD INCREASE DUTY CYCLE DECREASE DUE TO VIN INCREASE (FEED-FORWARD) DUTY CYCLE DECREASE DUE TO LOAD DECREASE (BODY BRAKING) OR FAULT (VCCLUV, OCP, VID=11111X) STEADY-STATE OPERATION Figure 3 PWM Operating Waveforms Lossless Average Inductor Current Sensing Inductor current can be sensed by connecting a series resistor and a capacitor network in parallel with the inductor and measuring the voltage across the capacitor, as shown in Figure 4. The equation of the sensing network is, vC ( s ) = vL ( s ) 1 RL + sL = iL ( s ) 1 + sRCS CCS 1 + sRCS CCS Usually the resistor Rcs and capacitor Ccs are chosen so that the time constant of Rcs and Ccs equals the time constant of the inductor which is the inductance L over the inductor DCR (RL). If the two time constants match, the voltage across Ccs is proportional to the current through L, and the sense circuit can be treated as if only a sense resistor with the value of RL was used. The mismatch of the time constants does not affect the measurement of inductor DC current, but affects the AC component of the inductor current. vL iL Current Sense Amp L RL RCS CCS VO CO vCS CSOUT Figure 4 Inductor Current Sensing and Current Sense Amplifier Page 9 of 21 V3.02 IR3506 The advantage of sensing the inductor current versus high side or low side sensing is that actual output current being delivered to the load is obtained rather than peak or sampled information about the switch currents. The output voltage can be positioned to meet a load line based on real time information. Except for a sense resistor in series with the inductor, this is the only sense method that can support a single cycle transient response. Other methods provide no information during either load increase (low side sensing) or load decrease (high side sensing). An additional problem associated with peak or valley current mode control for voltage positioning is that they suffer from peak-to-average errors. These errors will show in many ways but one example is the effect of frequency variation. If the frequency of a particular unit is 10% low, the peak to peak inductor current will be 10% larger and the output impedance of the converter will drop by about 10%. Variations in inductance, current sense amplifier bandwidth, PWM prop delay, any added slope compensation, input voltage, and output voltage are all additional sources of peak-to-average errors. Current Sense Amplifier A high speed differential current sense amplifier is located in the phase IC, as shown in Figure 5. Its gain is nominally 32.5 and the 3850 ppm/ºC increase in inductor DCR should be compensated in the voltage loop feedback path. The current sense amplifier can accept positive differential input up to 50mV and negative up to -10mV before clipping. The output of the current sense amplifier is summed with the DAC voltage and sent to the control IC and other phases through an on-chip 3KΩ resistor connected to the IOUT pin. The IOUT pins of all the phases are tied together and the voltage on the share bus represents the average current through all the inductors and is used by the control IC for voltage positioning and current limit protection. The input offset of this amplifier is calibrated to +/- 1mV in order to reduce the current sense error. The input offset voltage is the primary source of error for the current share loop. In order to achieve very small input offset error and superior current sharing performance, the current sense amplifier continuously calibrates itself. This calibration algorithm creates ripple on IOUT bus with a frequency of fsw/(32*28) in a multiphase architecture. Average Current Share Loop Current sharing between phases of the converter is achieved by the average current share loop in each phase IC. The output of the current sense amplifier is compared with the average current at the share bus. If current in a phase is smaller than the average current, the share adjust amplifier of the phase will pull down the starting point of the PWM ramp, thereby increasing its duty cycle and output current. In contrary, if current in a phase is larger than the average current, the share adjust amplifier of the phase will pull up the starting point of the PWM ramp thereby decreasing its duty cycle and output current. The current share amplifier is internally compensated so that the crossover frequency of the current share loop is much slower than that of the voltage loop and the two loops do not interact. Page 10 of 21 V3.02 IR3506 IR3506 THEORY OF OPERATION Block Diagram The Block diagram of the IR3506 is shown in Figure 5, and specific features are discussed in the following sections. CAL 1 15 CLKIN PHSOUT PHASE DELAY PHSIN PHSIN VCCL BOOST PWM COMPARATOR EAIN EAIN PWM PWM - GATE DRIVERS GATEH + SW NONOVERLAP CIRCUIT VCCL DACIN + PWM RESET VCCL PWM RAMP GENERATOR GATEL DACIN-SHARE_ADJ GATEL Disable SHARE_ADJ DIODE EMULATION PRESET COMPARATOR VCCL - 0.8V DE_PRESET + DEBUG OFF (LOW=OPEN) VCCL GATEs Disable PGND 0.15V DEBUG COMPARATOR DIODE EMULATION PRESET ENABLE LATCH - PHSIN CALIBRATION VCCL + RMPOUT IOUT NEGATIVE CURRENT COMPARATOR 75% REGULATION COMPARATOR - 75PCT_REG CURRENT SENSE AMPLIFIER - 3K + SHARE ADJUST AMPLIFIER + + - CAL DACIN CSAOUT IROSC DACIN + + LGND X 0.75 DACIN IROSC CSIN- X32.5 + CSIN+ CALIBRATION 1V PHSIN PSI PSI COMPARATOR - 500K PSI + 610mV 510mV Figure 5 Block diagram Tri-State Gate Drivers The gate drivers can deliver up to 2A peak current (4A sink current for bottom driver). An adaptive non-overlap circuit monitors the voltage on the GATEH and GATEL pins to prevent MOSFET shoot-through current while minimizing body diode conduction. The non-overlap latch is added to eliminate the error triggering caused by the switching noise. An enable signal is provided by the control IC to the phase IC without the addition of a dedicated signal line. The error amplifier output of the control IC drives low in response to any fault condition such as VCCL under voltage or output overload. Page 11 of 21 V3.02 IR3506 A synchronous rectification disable comparator is used to detect converter CSIN- pin voltage, which represents local converter output voltage. If the voltage is below 75% of VOUT1 and negative current is detected during startup, GATEL drives low, which disables synchronous rectification and eliminates negative current during powerup. Once VOUT1 reaches approximately 75 % of its nominal value, synchronous rectification is regain and can not be disable again until the startup. The gate drivers pull low if the supply voltages are below the normal operating range. An 80kΩ resistor is connected across the GATEH/GATEL and PGND pins to prevent the GATEH/GATEL voltage from rising due to leakage or other causes under these conditions. Over Voltage Protection (OVP) Output over-voltage might occur due to a high side MOSFET short or if the output voltage sense path is compromised. If the over-voltage protection comparators sense that either VOUT1 pin voltage exceeds VREF1 by 260mV or VOUT2 exceeds VREF1, the over voltage fault latch is set which pulls the error amplifier output low to turn off the converter power stage. The IR3522 communicates an OVP condition to the system by raising the CROWBAR pin voltage to within V(VCCL) – 0.2 V. With the error amplifiers outputs low, the low-side MOSFET turn-on within approximately 150ns. The low side MOSFET will remain low until the over voltage fault condition latch cleared. Page 12 of 21 V3.02 IR3506 PWM Ramp Every time the phase IC is powered up, PWM ramp magnitude is calibrated to generate a 53 mV/%DC. For example, for a 15 % duty ratio the ramp amplitude is 795mV. In response to a load step-up, the error amplifier can demand 100 % duty cycle. In order to avoid pulse skipping under this scenario and allow the BOOST cap to replenish, a minimum off-time is allowed in this mode of operation. As shown in Figure 6, 100 % duty is detected by comparing the PWM latch output (PWMQ) and its input clock (PWM_CLK). If the PWMQ is high when the PWM_CLK is asserted, the TopFET turnoff is initiated. The TopFET is again turned on once the RMPOUT drops within 200 mV of the VDAC. NORMAL OPERATION 100 % DUTY OPERATION CLKIN PHOUT (2 Phase Design) EAIN RMPOUT PWMQ Figure 1: PWM Operation during normal and 100 % duty mode. Power State Indicator (PSI) function From a system perspective, the PSI input is controlled by the system and is forced low when the load current is lower than a preset limit and forced high when load current is higher than the preset limit. IR3506 can accept an active low signal on its PSI input and force the drivers into tri-state, effectively forcing the phase IC into off state. A PSI-assert signal activates three features in the Phase IC. 1) It disconnects the IOUT pin from the IOUT bus: From a system perspective, IOUT is used to report current and is used for over-current protection. By disconnecting the disabled phase from the IOUT bus, proper current reporting and over-current protection level is ensured. 2) The DFF is disabled and it appears as a pass-through to the daisy chain loop: By removing the DFF from the daisy chain, the system ensures that proper phase delay is activated among the active phases. 3) The gate drivers are forced to tri-state, effectively, disabling the phase IC: Figure 7 shows the impact of PSI-assert on the gate drivers. After 8 cycle PHSIN delay, at the next CLK falling edge, the PSI_SYNC goes from 0 to 1. This disables the gate drives and shorts the DFF. Page 13 of 21 V3.02 IR3506 PSI 8 PHSIN Delay CLK PSI_SYNC D_PWM LATCH Figure 2: PSI assertion. Debugging Mode If CSIN+ pin is pulled up to VCCL voltage, IR3506 enters into debugging mode. Both drivers are pulled low and IOUT output is disconnected from the current share bus, which isolates this phase IC from other phases. However, the phase timing from PHSIN to PHSOUT does not change. Emulated Bootstrap Diode IR3506 integrates a PFET to emulate the bootstrap diode. If two or more top MOSFETs are to be driven at higher switching frequency, an external bootstrap diode connected from VCCL pin to BOOST pin may be needed. Operation at Higher Output Voltage The proper operation of the phase IC is ensured for maximum output voltage up to VCCL-2.5V if the differential input (CSIN(+) – CSIN(-)) to current sense amplifier remain below 30 mV. Otherwise, the maximum voltage output is calculated with the following equation: Vo _ max = VCCL − 1.5 − Gcs * (VCSIN + − VCSIN − ), where, Gcs is the current sense amplifier gain (typically 32.5). Page 14 of 21 V3.02 IR3506 DESIGN PROCEDURES - IR3506 Inductor Current Sensing Capacitor CCS and Resistor RCS The DC resistance of the inductor is utilized to sense the inductor current. Usually the resistor RCS and capacitor CCS in parallel with the inductor are chosen to match the time constant of the inductor, and therefore the voltage across the capacitor CCS represents the inductor current. If the two time constants are not the same, the AC component of the capacitor voltage is different from that of the real inductor current. The time constant mismatch does not affect the average current sharing among the multiple phases, but does effect the current signal IOUT as well as the output voltage during the load current transient if adaptive voltage positioning is adopted. Measure the inductance L and the inductor DC resistance RL. Pre-select the capacitor CCS and calculate RCS as follows. L RL RCS = (1) C CS Bootstrap Capacitor CBST Depending on the duty cycle and gate drive current of the phase IC, a capacitor in the range of 0.1uF to 1uF is needed for the bootstrap circuit. Decoupling Capacitors for Phase IC A 0.1uF-1uF decoupling capacitor is required at the VCCL pin. CURRENT SHARE LOOP COMPENSATION The internal compensation of current share loop ensures that crossover frequency of the current share loop is at least one decade lower than that of the voltage loop so that the interaction between the two loops is eliminated. The crossover frequency of current share loop is approximately 8 kHz. Page 15 of 21 V3.02 IR3506 LAYOUT GUIDELINES The following layout guidelines are recommended to reduce the parasitic inductance and resistance of the PCB layout, therefore minimizing the noise coupled to the IC. • Dedicate at least one middle layer for a ground plane, which is then split into signal ground plane (LGND) and power ground plane (PGND). • Separate analog bus (EAIN, DACIN, and IOUT) from digital bus (CLKIN, PHSIN, and PHSOUT) to reduce the noise coupling. • Connect PGND to LGND pins to their respective ground planes through vias. • Place current sense resistors and capacitors (RCS and CCS) close to phase IC. Use Kelvin connection for the inductor current sense wires, but separate the two wires by ground polygon. The wire from the inductor terminal to CSIN- should not cross over the fast transition nodes, i.e. switching nodes, gate drive outputs and bootstrap nodes. • Place the decoupling capacitor CVCCL as close as possible to the VCCL pin. • Place the phase IC as close as possible to the MOSFETs to reduce the parasitic resistance and inductance of the gate drive paths. • Place the input ceramic capacitors close to the drain of top MOSFET and the source of bottom MOSFET. Use combination of different packages of ceramic capacitors. • There are two switching power loops. One loop includes the input capacitors, top MOSFET, inductor, output capacitors and the load; another loop consists of bottom MOSFET, inductor, output capacitors and the load. Route the switching power paths using wide and short traces or polygons; use multiple vias for connections between layers. To Gate Drive Voltage To Digital Bus To Analog Bus LGND PLANE To VIN IOUT DACIN ISHARE DACIN LGND LGND PHSIN PHSIN To VIN R EAIN CLKIN CSIN- PGND CSIN+ D BST VCC C C VCC CS R CS VCCL C VCCL SW BOOST SW GATEH VCCL C GATEH VCC BOOST GATEL VCCL CS C C CS PHSOUT To Bottom MOSFET PGND PLANE Page 16 of 21 BST C C D BST BST To Switching Node To Top MOSFET To LGND Plane Ground Polygon To Inductor V3.02 IR3506 PCB Metal and Component Placement • Lead land width should be equal to nominal part lead width. The minimum lead to lead spacing should be ≥ 0.2mm to prevent shorting. • Lead land length should be equal to maximum part lead length + 0.3 mm outboard extension + 0.05mm inboard extension. The outboard extension ensures a large and inspectable toe fillet, and the inboard extension will accommodate any part misalignment and ensure a fillet. • Center pad land length and width should be equal to maximum part pad length and width. However, the minimum metal to metal spacing should be ≥ 0.17mm for 2 oz. Copper (≥ 0.1mm for 1 oz. Copper and ≥ 0.23mm for 3 oz. Copper) • Four 0.3mm diameter vias shall be placed in the pad land spaced at 0.85mm, and connected to ground to minimize the noise effect on the IC, and to transfer heat to the PCB • No pcb traces should be routed nor vias placed under any of the four corners of the IC package. Doing so can cause the IC to raise up form the pcb resulting in poor solder joints to the IC leads. Page 17 of 21 V3.02 IR3506 Solder Resist • The solder resist should be pulled away from the metal lead lands by a minimum of 0.06mm. The solder resist mis-alignment is a maximum of 0.05mm and it is recommended that the lead lands are all Non Solder Mask Defined (NSMD). Therefore pulling the S/R 0.06mm will always ensure NSMD pads. • The minimum solder resist width is 0.13mm. • At the inside corner of the solder resist where the lead land groups meet, it is recommended to provide a fillet so a solder resist width of ≥ 0.17mm remains. • The land pad should be Solder Mask Defined (SMD), with a minimum overlap of the solder resist onto the copper of 0.06mm to accommodate solder resist mis-alignment. In 0.5mm pitch cases it is allowable to have the solder resist opening for the land pad to be smaller than the part pad. • Ensure that the solder resist in-between the lead lands and the pad land is ≥ 0.15mm due to the high aspect ratio of the solder resist strip separating the lead lands from the pad land. • The four vias in the land pad should be tented or plugged from bottom board side with solder resist. Page 18 of 21 V3.02 IR3506 Stencil Design • The stencil apertures for the lead lands should be approximately 80% of the area of the lead lands. Reducing the amount of solder deposited will minimize the occurrence of lead shorts. Since for 0.5mm pitch devices the leads are only 0.25mm wide, the stencil apertures should not be made narrower; openings in stencils < 0.25mm wide are difficult to maintain repeatable solder release. • The stencil lead land apertures should therefore be shortened in length by 80% and centered on the lead land. • The land pad aperture should be approximately 70% area of solder on the center pad. If too much solder is deposited on the center pad the part will float and the lead lands will be open. • The maximum length and width of the land pad stencil aperture should be equal to the solder resist opening minus an annular 0.2mm pull back to decrease the incidence of shorting the center land to the lead lands when the part is pushed into the solder paste. Page 19 of 21 V3.02 IR3506 PACKAGE INFORMATION 16L MLPQ (3 x 3 mm Body) – θJA = 38oC/W, θJC = 3oC/W Page 20 of 21 V3.02 IR3506 Data and specifications subject to change without notice. This product has been designed and qualified for the Consumer market. Qualification Standards can be found on IR’s Web site. IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, California 90245, USA Tel: (310) 252-7105 TAC Fax: (310) 252-7903 Visit us at www.irf.com for sales contact information. www.irf.com Page 21 of 21 V3.02