INTEGRATED CIRCUITS SA57254-XX CMOS switching regulator (PWM controlled) Product data Supersedes data of 2001 Aug 01 2003 Nov 11 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX GENERAL DESCRIPTION The SA57254-XX is a fully integrated DC/DC converter circuit. Efficient, compact power conversion is achieved with a pulse width modulation (PWM) controlled switching regulator circuit designed using CMOS processing. Low ripple and high efficiency of typically 83% are achieved through PWM control. The regulator has a high precision output with ±2.4% accuracy. Few external components are required. The SA57254-XX has a built-in soft-start circuit which reduces inrush current and voltage overshoot during start-up. A low resistance CMOS power FET, which has low leakage current and low parasitic capacitance, is on-chip. FEATURES APPLICATIONS • Operates from 0.7 to 9 VDC • Ultra low operating supply current—typically 17 µA • Built-in power FET • High efficiency—typically 83% • High precision output—typically ±2.4% • Operating temperature range of –40 to +85 °C • Available output voltages: 2.0, 2.5, 2.8, 3.0, 3.3, 3.6, 5.0 VDC • Available in a 5-lead small outline surface mount package • Mobile and portable phones • Instrumentation and industrial products • Other portable, battery-operated equipment (SOP003) SIMPLIFIED SYSTEM DIAGRAM VOUT VBATT SW 5 VDD FB 1 2 SA57254-XX VREF R PWM CONTROL R GND 4 SOFT START SA57254-XX as a boost (step-up) converter. SL01501 Figure 1. Simplified system diagram. 2003 Nov 11 2 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX ORDERING INFORMATION PACKAGE TYPE NUMBER SA57254-XXGW NAME DESCRIPTION VERSION TEMPERATURE RANGE SOT23-5, SOT25, SO5 Plastic small outline package; 5 leads; body width 1.6 mm SOP003 –40 to +85 °C NOTE: The device has seven voltage output options, indicated by the XX on the Type Number. XX VOLTAGE (Typical) 20 2.0 V 25 2.5 V 28 2.8 V 30 Part number marking Each device is marked with a four letter code. The first three letters designate the product. The fourth letter, represented by ‘x’, is a date tracking code. Part number Marking SA57254-20GW AEKx SA57254-25GW AELx SA57254-28GW AEMx 3.0 V SA57254-30GW AENx 33 3.3 V SA57254-33GW AEPx 36 3.6 V SA57254-36GW AERx 50 5.0 V SA57254-50GW AESx PIN CONFIGURATION PIN DESCRIPTION PIN 1 VDD 2 N/C 3 5 SA57254-XX FB 4 SW GND SYMBOL DESCRIPTION 1 FB Feedback from the output voltage to the PWM control. 2 VDD Voltage input to regulator. 3 N/C No connection. 4 GND Ground. 5 SW Switching output to inductor. SL01502 Figure 2. Pin configuration. MAXIMUM RATINGS MIN. MAX. UNIT VIN(max) SYMBOL Power supply voltage –0.3 11 V VFB FB pin voltage –0.3 11 V VSW SW pin voltage –0.3 11 V ISW SW pin current – 300 mA Toper Operating temperature –40 +85 °C Tstg Storage temperature –40 +125 °C PD Power dissipation – 150 mW 2003 Nov 11 PARAMETER 3 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX ELECTRICAL CHARACTERISTICS Tamb = 25 °C, unless otherwise specified. SYMBOL PARAMETER CONDITIONS VIN input voltage VST1 operating start voltage VST2 oscillator start voltage VHLD operation hold voltage IOUT = 1.0 mA ISS1 consumption current 1 VOUT = output voltage × 0.95 ISS2 RDS(ON) ( ) consumption current 2 internal switch-on resistance VOUT = output voltage + 0.5 V VSW = 0.4 V MIN. TYP. MAX. UNIT – – – 9.0 V – – – 0.9 V – – – 0.8 V – 0.7 – – V -20 – 11.6 19.4 µA -25 – 14.3 23.9 µA -28 – 16.1 26.8 µA -30 – 17.2 28.7 µA -33 – 19.1 31.8 µA -36 – 22.4 37.3 µA -50 – 38.5 64.1 µA -20 – 3.1 6.2 µA -25 – 3.2 6.3 µA -28 – 3.2 6.4 µA -30 – 3.2 6.4 µA -33 – 3.3 6.5 µA -36 – 3.3 6.5 µA -50 – 3.5 6.9 µA -20 – 5.6 8.9 Ω -25 – 4.1 6.5 Ω -28 – 4.1 6.5 Ω -30 – 3.2 5.1 Ω -33 – 3.2 5.1 Ω -36 – 3.2 5.1 Ω -50 – 2.2 3.5 Ω VOUT = VSW = 9 V – – – 1.0 µA IOUT = 10 mA ≈ IOUT (following) × 1.25 – – 30 60 mV –40 °C ≤ Tamb ≤ +85 °C – – ±50 – ppm/°C oscillator frequency VOUT = output voltage × 0.95 – 42.5 50 57.5 kHz maximum duty ratio VOUT = output voltage × 0.95 – 75 83 90 % IOUT = 1.0 mA – 3.0 6.0 12 ms -20 – 75 – % -25 – 79 – % -28 – 79 – % -30 – 83 – % -33 – 83 – % -36 – 83 – % -50 – 87 – % ISWO switching transistor leak current ∆VOUT2 load ripple voltage ∆VOUT/∆Tamb output voltage temperature coefficient fOSC MaxDuty tSS soft start time EFFI efficiency 2003 Nov 11 IOUT = 1.0 mA Part # 4 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX TYPICAL PERFORMANCE CURVES 50 5 VOUT = OUTPUT VOLTAGE × 0.95 VOUT = OUTPUT VOLTAGE + 0.5 V 40 4 30 3 ISS2 (µA) ISS1 (µA) 20 2 10 1 0 –40 –20 0 20 40 60 80 0 –40 100 –20 0 20 Tamb 〈°C) 40 60 80 100 Tamb 〈°C) SL01456 SL01457 Figure 3. Supply current 1 versus temperature. Figure 4. Supply current 2 versus temperature. 70 50 VOUT = OUTPUT VOLTAGE × 0.95 Tamb = 25 °C 65 40 60 55 fOSC (kHz) 30 ISS1, 2 (µA) 50 20 45 40 10 35 30 –40 0 –20 0 20 40 60 80 100 0 2 4 Tamb 〈°C) 6 8 10 VOUT (V) SL01458 SL01459 Figure 5. Oscillator frequency versus temperature. Figure 6. Supply current 1, 2 versus VOUT. 70 250 Tamb = 25 °C VCONT = 0.4 V Tamb = 25 °C 200 60 fOSC (kHz) 150 ISW (mA) 50 100 40 50 0 0 1 2 3 4 30 5 VOUT (V) 0 1 2 3 4 SL01462 SL01471 Figure 7. Typical switch current versus VOUT. 2003 Nov 11 5 VOUT (V) Figure 8. Oscillator frequency versus VOUT. 5 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX VIN = 1.8 V VIN = 1.8 V VOUT OUTPUT VOLTAGE (20 mV/div) VOUT OUTPUT VOLTAGE (20 mV/div) VSW SW VOLTAGE (1 V/div) VSW SW VOLTAGE (1 V/div) t (10 µs/div) t (10 µs/div) SL01472 SL01473 Figure 9. Ripple voltage at IOUT = 200 µA. Figure 10. Ripple voltage at IOUT = 10 mA. IOUT = 60 mA VIN = 1.8 V VOUT OUTPUT VOLTAGE (20 mV/div) VIN INPUT VOLTAGE (1 V/div) VSW SW VOLTAGE (1 V/div) VOUT OUTPUT VOLTAGE (1 V/div) t (10 µs/div) t (1 ms/div) SL01474 SL01475 Figure 12. Start-up characteristic VIN: 0 V → 1.8 V. Figure 11. Ripple voltage at IOUT = 60 mA. IOUT: 100 µA → 50 mA; VIN = 1.8 V IOUT LOAD CURRENT (20 mA/div) VOUT OUTPUT VOLTAGE (50 mV/div) t (200 µs/div) SL01477 Figure 13. Output load regulation, increasing current. 2003 Nov 11 6 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX IOUT: 50 mA → 100 µA; VIN = 1.8 V VIN: 1.8 V → 2.4 V; IOUT = 50 mA IOUT LOAD CURRENT (20 mA/div) VIN INPUT VOLTAGE (500 mV/div) VOUT OUTPUT VOLTAGE (50 mV/div) VOUT OUTPUT VOLTAGE (50 mV/div) t (100 µs/div) t (5 ms/div) SL01478 SL01479 Figure 14. Output load regulation, decreasing current. Figure 15. Input line regulation, increasing voltage. 1.0 VIN: 2.4 V → 1.8 V; IOUT = 50 mA 0.9 VIN , INPUT VOLTAGE (V) 0.8 VIN INPUT VOLTAGE (500 mV/div) VOUT OUTPUT VOLTAGE (50 mV/div) 0.7 0.6 0.5 0.4 0.3 0.2 VST1 0.1 VDO 0.0 0 1 2 t (200 µs/div) 3 4 5 6 7 8 SL01480 Figure 17. Output current versus starting voltage. 1.0 500 0.9 450 0.8 400 I IN , INPUT CURRENT ( µA) VIN , INPUT VOLTAGE (V) 10 SL01481 Figure 16. Input line regulation, decreasing voltage. 0.7 0.6 0.5 0.4 0.3 0.2 VST1 0.1 VDO VOUT = 2 V VOUT = 3 V VOUT = 5 V 350 300 250 200 150 100 50 0.0 0 0 1 2 3 4 5 6 7 8 9 10 0.0 IOUT, OUTPUT CURRENT (mA) 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 VIN, INPUT VOLTAGE (V) SL01482 SL01483 Figure 18. Input voltage versus output current. 2003 Nov 11 9 IOUT, OUTPUT CURRENT (mA) Figure 19. Input voltage versus supply current. 7 Philips Semiconductors Product data 100 5.08 95 5.06 90 5.04 85 5.02 5.00 4.98 4.96 VIN = 0.9 V VIN = 1.8 V VIN = 3.0 V VIN = 4.0 V 4.94 4.92 80 75 70 65 VIN = 0.9 V VIN = 1.8 V VIN = 3.0 V VIN = 4.0 V 60 55 4.90 50 0.01 0.1 1 10 100 1000 0.01 IOUT, OUTPUT CURRENT (mA) VIN = 0.9 V VIN = 1.8 V VIN = 2.4 V 70 60 50 40 30 20 10 0 0.01 0.1 1 10 100 1000 IOUT, OUTPUT CURRENT (mA) SL01486 Figure 22. Output current versus ripple voltage (L0 = 100 µH; COUT = 33 µF). 2003 Nov 11 10 100 1000 Figure 21. Output current versus efficiency (100 µH inductor). 100 80 1 SL01485 Figure 20. Output current versus voltage (100 µH inductor). 90 0.1 IOUT, OUTPUT CURRENT (mA) SL01484 VR , RIPPLE VOLTAGE (mV) SA57254-XX 5.10 EFFI , EFFICIENCY (%) VOUT , OUTPUT VOLTAGE (V) CMOS switching regulator (PWM controlled) 8 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX TECHNICAL DISCUSSION The SA57254-XX has an internal common-source power switching MOSFET which provides the PWM signal to the inductor. The voltage error amplifier for maintaining a fixed output voltage, the oscillator and a PWM generator are all in the package. General discussion The SA57254-XX is a highly integrated boost-mode switching power supply integrated circuit. Each device is set to provide a fixed output voltage by having a fully compensated internal voltage feedback loop. The SA57254-XX operates at a fixed frequency of 50 kHz and can operate from a single alkaline cell (0.9 V) or up to 9 V. SW 5 VDD 2 FB 1 SA57254-XX VREF R PWM CONTROL R GND 4 SOFT START SL01503 Figure 23. Functional diagram. 2003 Nov 11 9 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX APPLICATION INFORMATION PASSIVE SNUBBER (OPTIONAL) The SA57254-XX can be used for a simple boost (step-up) converter or the less commonly used flyback converter (isolated boost). The major operating restriction of the simple boost converter is that its output voltage must always be above the highest expected value of the input voltage. The flyback converter circuit requires more parts, but the output voltage is not restricted by the input voltage. L0 VIN VOUT D SW VDD FB CIN COUT Boost converter fundamentals The boost or step-up converter is a non-dielectrically isolated switching power supply topology (arrangement of power parts). That is, the input power source is directly connected to the output load (ground and signals). A typical boost converter, with an optional passive snubber, can be seen in Figure 24. SA57254-XX To understand the boost converter’s operation, examine its three periods of operation. These periods are: the power switch on-time (period 1); the inductor discharge period (period 2); and the inductor empty state (period 3). These periods and their associated currents can be seen in Figure 25. GND SL01504 Figure 24. Boost converter. SPIKE ENERGY BEING TRANSFERRED TO OUTPUT SWITCH VOLTAGE (V) +VOUT ENERGY BEING STORED IN INDUCTOR CORE EMPTY, PARASITIC CIRCULATING ENERGY +VIN PERIOD 2 PERIOD 1 PERIOD 3 ISW × RDS(ON) 0 INDUCTOR CURRENT (A) Ipeak VIN (VIN – VOUT) L0 L0 SL01464 Figure 25. Boost converter waveforms (discontinuous mode). 2003 Nov 11 10 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) Period 3: inductor empty state Period 1: power switch on-time During this period, a simple circuit loop is formed when the power switch is on. The input voltage source is connected directly across the boost inductor (L0). A current ramp is exhibited whose slope is described by: IL (on) + V IN L0 DISCONTINUOUS MODE—This period as displayed in Figure 25 occurs in the discontinuous–mode of operation of a boost converter. It is identified by a period of “ringing” following the output period (period 2). The inductor has been completely emptied of its stored energy and the switched node returns to the level of the input voltage. Ringing is seen at this node because a resonant circuit is formed by the inductance of L0 and any parasitic inductances and capacitances connected to that node. This ringing has very little energy and can easily be eliminated by a small passive snubber. Eqn. (1) Energy is then stored within the core material of the inductor and is described by: E sto + 0.5L 0 I peak 2 CONTINUOUS MODE—If the inductor is not completely emptied of its stored energy before the power switch turns on again, the converter is operating in the continuous mode. A small amount of residual flux (energy) remains in the inductor core and the current waveform jumps to an initial value when the power switch is again turned-on. This mode offers some advantages over the discontinuous-mode, because the peak current seen by the power switch is lower. In low voltage applications, the inductor can store more energy with lower peak currents. Eqn. (2) This current ramp continues until the controller turns off the power switch. Period 2: inductor discharge period The instant the power switch turns off, the current flowing through the inductor forces the voltage at its output node (switched node) to rise quickly above the input voltage (spike). This voltage is then clamped when it exceeds the device’s output voltage and the output rectifier becomes forward biased. The inductor empties its stored energy in the form of a linearly decreasing current ramp whose slope is dictated by: I L(off) [ V IN * V OUT L0 SA57254-XX The continuous mode waveforms can be seen in Figure 26. Eqn. (3) The stored energy is transferred to the output capacitor. This output current continues until the magnetic core is completely emptied of its stored energy or the power switch turns back on. SWITCH VOLTAGE (V) SPIKE +VOUT +VIN ENERGY BEING STORED IN INDUCTOR ENERGY BEING TRANSFERRED TO OUTPUT INDUCTOR CURRENT (A) 0 VIN L0 Ipeak (VIN – VOUT) L0 RESIDUAL FLUX SL01465 Figure 26. Continuous mode waveforms. 2003 Nov 11 11 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) The minimum value of the output capacitor can be estimated by Equation (7). Determining the value of the boost inductor The precise value of the boost inductor is not critical to the operation of the SA57254-XX. The value of the boost inductor should be calculated to provide continuous-mode operation over most of its operating range. The converter may enter the discontinuous-mode when the output load current falls to less than about 20 percent of the full-load current. C OUT u T on C IN u Eqn. (8) Forward voltage drop (Vf)—This is the voltage across the rectifier when a forward current is flowing through the rectifier. A P-N ultra-fast diode exhibits a 0.7 – 1.4 volt drop, and this drop is relatively fixed over the range of forward currents. A Schottky diode exhibits a 0.3 – 0.6 volt drop and appears more resistive during the forward conduction periods. That is, its forward voltage drop increases with increasing currents. You can gain an advantage by purposely over-rating the current rating of a Schottky rectifier to minimize this increasing voltage drop. Determining the minimum value of the capacitors The input and output capacitors experience the current waveforms seen in Figures 25 and 26. The peak currents can be typically between 3 to 6 times the average currents flowing into the input and from the output. This makes the choice of capacitor an issue of how much ripple voltage can be tolerated on the capacitor’s terminals and how much heating the capacitor can tolerate. At the power levels produced by the SA57254-XX heating is not a major issue. Reverse recovery time (Trr)—This is an issue when the boost supply is operating in the continuous-mode. Trr is the amount of time required for the rectifier to assume an open circuit when a forward current is flowing and a reverse voltage is then placed across its terminals. P-N ultra-fast rectifiers typically have a 25–40 ns reverse recovery time. Schottky rectifiers have a very short or no reverse recovery time. The Equivalent Series Resistance (ESR) of the capacitor, the resistance that appears between its terminals, and the actual capacitance causes heat to be generated within the case whenever there is current entering or exiting the capacitor. ESR also adds to the apparent voltage drop across the capacitor. The heat that is generated can be approximated by Equation (5). Eqn. (5) Forward recovery time (Tfr)—This is the amount of time before a rectifier begins conducting forward current after a forward voltage is placed across its terminals. This parameter is not always well specified by the rectifier manufacturers. It causes a spike to appear when the power switch turns off. This particular point in its operation causes the most radiated noise. Several rectifiers may have to be evaluated for the prototype. After the final output rectifier selection is made, if the spike is still causing a problem a small passive snubber can be placed across the rectifier. ESR’s effect on the capacitor voltage is given by Equation (6). (expressed as Vp–p) V drop Selecting the output rectifier The output rectifier (D) is critical to the efficiency and low-noise operation of the boost converter. The majority of the loss within the supply will be caused by the output rectifier. Three parameters are important in the rectifier’s operation within a boost-mode supply. These are defined below. This is an estimated inductor value and you can select an inductance value slightly higher or lower with little effect on the converter’s operation. If the design falls out of regulation within the desired operating range, reduce the inductance value, but by no more than 30 percent. DV C ^ I peak(R ESR) (I peak) (T on) These calculations should produce a good estimate of the needed values of the input and output capacitors to yield the desired ripple voltages. Where: VIN(min) is the lowest expected input operating voltage (V). Ton is about 10 µs or one-half the switching period (s). Ipeak is the maximum peak current for the SA57254-XX (0.3 A). P D(in watts) ^ (1.8I av) 2 (R ESR) Eqn. (7) Where: Ipeak is the expected maximum peak current of the switch (A). Ton is the on-time of the switch (sec) [≅10 µs]. Vdrop is the desired amount of voltage drop across the capacitor (Vp–p). Eqn. (4) I peak V ripple(p*p) Finding the value of the input capacitor is done by Equation (8). To determine the nominal value of the inductance, use Equation (4). V IN(min) (I OUT(max)) (T off) Where: IOUT is the average value of the output load current (A). Toff is the nominal off–time of the power switch (sec) [≅10 µs]. Vripple is the desired amount of ripple voltage (Vp–p). At low input voltages, the time required to store the needed energy lengthens, but the time needed to empty the inductor’s core of its energy shrinks. Conversely, at high input voltages, the time needed to store the energy shrinks while the time needed to empty the core increases. See Equations (1) and (3). At the extremes of these conditions, the converter will fall out of regulation, that is the output voltage will begin to fall, because the time needed for either storing or emptying the stored inductor energy is too short to support the output load current. L0 ^ SA57254-XX Eqn. (6) A ceramic capacitor would typically be used in this application if the required value is less than 1 – 10 µF, or a tantalum capacitor for required values of 10 µF and above. Lower cost aluminum electrolytic capacitors can be used, but you should confirm that the higher ESRs typically exhibited by these capacitors does not cause a problem. For this boost application, the best choice of output rectifier is a low forward drop, 0.5 – 1 ampere, 20 volt Schottky rectifier such as the Philips part number BAT120A. 2003 Nov 11 12 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX Flyback converter The SA57254-XX can also be used to create a flyback converter, also known as an isolated boost converter. The advantage of a flyback converter is that the input voltage can go higher or lower than the output voltage without affecting the operation of the converter. The only restrictions are the peak current flowing into the switch pin (SW) and the breakdown voltage of the SW and feedback (VFB) pins. COILTRONICS CTX100-1P VBATT SW FB SA57254-33 The output voltage of the flyback can be changed by using a SA57254-XX with the desired output voltage, with no other changes to the circuit. GND Selecting the components It is best to operate the transformer in the continuous-mode where the highest expected peak primary current is below the maximum current rating of the SA57254-XX switch. SL01505 Figure 27. Flyback converter circuit. Because the SA57254-XX is a peak current-limitied IC, begin with a peak current equal to or less than the maximum current rating of the part (0.3 A). A reasonable value of the primary inductance can be found in Equation (9). VIN + VOUT (1:1 TRANSFORMER) SWITCH VOLTAGE (V) T on I peak Eqn. (9) Where: Ipeak is 0.3 A or less. Ton is the maximum expected on-time of the switch (≈10 µs). VIN(min) is the lowest expected input voltage (V). VIN +VOUT SECONDARY VOLTAGE (V) Then select an off-the-shelf transformer such as the Coiltronics CTX100–1P, a 1:1 turns ratio transformer that has a primary inductance of 100 µH. It does not reach saturation until the primary current reaches 440 mA, which is above the expected peak current of the flyback converter. The 1:1 turns ratio should work for output voltages from 0.8 to 2 times the highest input voltage, and produce the output voltage set by the SA57254-XX. The only other restriction is that the input voltage plus the output voltage must be less than the breakdown voltage of the SA57254-XX (9 V). Use Equation (8) to determine the minimum value for the input capacitor. A 0.1 V drop is desired across this capacitor. (0.3A) (10ms) + 30mF 0.1V GROUND (0 V) –VIN SWITCH CURRENT (A) C IN u 3.3 V @ 0.1 A 47 µF @ 10 V 47 µF @ 10 V One transformer can accommodate a variety of output voltages in different applications, because the circuit will change the on and off–times to provide the desired output voltage. L pri t 5V IN(min) VOUT A 47 µF at 6 V tantalum capacitor would be suitable. For the design example, the output voltage will be +3.3 V with a maximum output current of 50 mA. The input voltage can vary between +1.8 V and 4.0 V. The design can be seen in Figure 27, and the expected waveforms can be seen in Figure 28. ISW(peak) DIODE CURRENT (A) VIN Idiode(peak) Ipeak = Idiode (1:1 TRANSFORMER) SL01468 Figure 28. Flyback converter waveforms. 2003 Nov 11 13 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) Designing a passive snubber If the switching power supply is generating too much radio frequency interference (RFI) a passive snubber can be added. A passive snubber is a series resistor and capacitor placed across any component that exhibits a resonant “ringing”. This series R-L-C loop creates a lossy or damped tank circuit that dissipates the ringing energy. The design is critical, because it introduces another loss within the converter. SA57254-XX PASSIVE SNUBBER VOUT VBATT R Designing a snubber is an empirical process, mainly because it involves undefined parasitic capacitances and inductances contributed by the PCB layout, leakage inductance, and device capacitances. The snubber should be placed across the major source of the spike or ringing which is the output rectifier for a boost converter (see Figure 24) and the primary winding of the transformer for a flyback transformer. SW FB VDD SA57254-XX GND The usual design process is: SL01506 1. Measure the period of the undesired ringing (T0). Figure 29. Flyback converter with passive snubber. 2. Place a very small ceramic capacitor (about 10 pF) across the output rectifier or primary winding. 3. Re-measure the period of the undesired ringing. The new period should be about 3 times that of T0. If it is less than this, place a slightly larger value of capacitor across the output rectifier or primary winding. 4. Once the desired increase in the ringing period is achieved with a capacitance (C0), place a resistor in series with the capacitor whose value is approximately: R snubber T0 2pC 0 Eqn. (10) This should produce a snubber that does not load the circuit and introduces a very small loss. 2003 Nov 11 14 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) • On a 2-sided board, do not run sensitive signals traces under the Laying out the printed circuit board The design of the printed circuit board (PCB) is critical to the proper operation of all switching power supplies. Its design affects the supply stability, radio frequency interference behavior and the reliability of the converter. AC voltage node. • The IC (control) ground is terminated at the output capacitor’s negative terminal. Never use the autoroute feature of any PCB design program because this will always produce traces that are too long and too thin. Designing the PCB for effective heat dissipation The maximum junction temperature is +125 °C, which should not be exceeded under any operating conditions. Designing a PCB that includes a heatsink system under the device is the key to cooler operation of the circuit, and the long–term reliable operation of the converter. The input and output capacitors are the only source or sink of the high frequency currents found in a switching power supply. All connections to the switching power supply from the outside circuits should be made to the input or output capacitor terminals (+ and –). Internally, the layout should adhere to a “one-point” grounding system, as shown in Figure 30. L0 VIN The major sources of heat within the converter are the power switch inside the SA57254-XX, the resistive losses within the inductor, and losses associated with the output rectifier. These losses can be estimated by the following equations: Power switch: VOUT SW VDD CIN P D(sw) ^ T ON FB GND P D(L0) ^ I pk 2 R DS(ON) 2 R winding Output rectifier: P D(rect) ^ I OUT(Vfwd) f SW Eqn. (11) Eqn. (12) Eqn. (13) The thermal resistance (Rth(j-a)) of the SA57254-XX is approximately 220 °C/W, assuming the device is soldered to a 2 oz. copper FR4 fiberglass circuit board, and that the minimum footprint was used (copper just under the leads). A rule of thumb in PCB design is that the thermal resistance can be reduced by 30% for each doubling of the copper area close to the device. This effect diminishes for areas greater than five times the minimum PCB footprint. If you take advantage of this rule, thermal resistance can be reduced by using wide copper lands when connecting to the leads of the major power-producing parts. These PCB traces should almost fill the areas surrounding the converter parts to conduct heat away from the device. For demanding applications, additional heat dissipation area can be created by placing a copper island on the opposite side of the PCB from each wide trace and connecting it to the trace with vias (plated thru holes). OUTPUT GROUND TO ONE POINT SL01507 Figure 30. Grounding trace for converter. The traces between the input and output capacitors and the inductor, power switch and rectifier(s) should be as short and wide as possible. This reduces the series resistance and inductance that can be introduced by traces. The guidelines for a PCB layout can be summarized as: • The traces between the input and output capacitor to the inductor, The junction temperature can be estimated by Equation (14). power switch and the rectifier should be made as short and as wide as possible. T j ^ (P D • Strictly adhere to the one-point wiring practices shown in R th(j-a)Ȁ) ) T amb(max) Eqn. (14) Where: PD is the power dissipation (W). Rth(j-a)′ is the effective thermal resistance with the additional copper (°C/W). Tamb is the highest local expected ambient temperature (°C). Figure 30. 2003 Nov 11 I PK Inductor: COUT SA57254-XX INPUT GROUND TO ONE POINT SA57254-XX 15 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX PACKING METHOD The SA57254-XX is packed in reels, as shown in Figure 31. GUARD BAND TAPE REEL ASSEMBLY TAPE DETAIL COVER TAPE CARRIER TAPE BARCODE LABEL BOX SL01305 Figure 31. Tape and reel packing method. 2003 Nov 11 16 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) Plastic small outline package; 5 leads; body width 1.6 mm 2003 Nov 11 17 SA57254-XX SOP003 Philips Semiconductors Product data CMOS switching regulator (PWM controlled) SA57254-XX REVISION HISTORY Rev Date Description _2 20031111 Product data (9397 750 12317). ECN 853-2272 30331 of 09 September 2003. Supersedes data of 2001 Aug 01 (9397 750 08875). Modifications: • Change package outline version to SOP003 in Ordering information table and Package outline sections. _1 20010801 Product data (9397 750 08875). ECN 853-2272 26807 of 01 August 2001. Data sheet status Level Data sheet status [1] Product status [2] [3] Definitions I Objective data Development This data sheet contains data from the objective specification for product development. Philips Semiconductors reserves the right to change the specification in any manner without notice. II Preliminary data Qualification This data sheet contains data from the preliminary specification. Supplementary data will be published at a later date. Philips Semiconductors reserves the right to change the specification without notice, in order to improve the design and supply the best possible product. III Product data Production This data sheet contains data from the product specification. Philips Semiconductors reserves the right to make changes at any time in order to improve the design, manufacturing and supply. Relevant changes will be communicated via a Customer Product/Process Change Notification (CPCN). [1] Please consult the most recently issued data sheet before initiating or completing a design. [2] The product status of the device(s) described in this data sheet may have changed since this data sheet was published. The latest information is available on the Internet at URL http://www.semiconductors.philips.com. [3] For data sheets describing multiple type numbers, the highest-level product status determines the data sheet status. Definitions Short-form specification — The data in a short-form specification is extracted from a full data sheet with the same type number and title. For detailed information see the relevant data sheet or data handbook. Limiting values definition — Limiting values given are in accordance with the Absolute Maximum Rating System (IEC 60134). Stress above one or more of the limiting values may cause permanent damage to the device. These are stress ratings only and operation of the device at these or at any other conditions above those given in the Characteristics sections of the specification is not implied. Exposure to limiting values for extended periods may affect device reliability. Application information — Applications that are described herein for any of these products are for illustrative purposes only. Philips Semiconductors make no representation or warranty that such applications will be suitable for the specified use without further testing or modification. Disclaimers Life support — These products are not designed for use in life support appliances, devices, or systems where malfunction of these products can reasonably be expected to result in personal injury. Philips Semiconductors customers using or selling these products for use in such applications do so at their own risk and agree to fully indemnify Philips Semiconductors for any damages resulting from such application. Right to make changes — Philips Semiconductors reserves the right to make changes in the products—including circuits, standard cells, and/or software—described or contained herein in order to improve design and/or performance. When the product is in full production (status ‘Production’), relevant changes will be communicated via a Customer Product/Process Change Notification (CPCN). Philips Semiconductors assumes no responsibility or liability for the use of any of these products, conveys no license or title under any patent, copyright, or mask work right to these products, and makes no representations or warranties that these products are free from patent, copyright, or mask work right infringement, unless otherwise specified. Koninklijke Philips Electronics N.V. 2003 All rights reserved. Printed in U.S.A. Contact information For additional information please visit http://www.semiconductors.philips.com. Fax: +31 40 27 24825 Date of release: 11-03 For sales offices addresses send e-mail to: [email protected]. Document order number: 2003 Nov 11 18 9397 750 12317