TI bq24741RHDT Li-ion or li-polymer battery charger with low iq and accurate trickle charge Datasheet

bq24741
www.ti.com ...................................................................................................................................................... SLUS875A – MARCH 2009 – REVISED MARCH 2009
Li-Ion or Li-Polymer Battery Charger with Low Iq and Accurate Trickle Charge
FEATURES
APPLICATIONS
•
•
•
•
•
•
•
•
SW
REGN
LODR
PGND
27
26
25
24
23
22
CE
1
21
DPMDET
ACN
2
20
CELLS
ACP
3
19
CSP
LPMOD
4
18
CSN
ACDET
5
17
BAT
ACSET
6
16
ISET
LPREF
7
15
IADAPT
bq24741
QFN-28
8
9
10
11
12
13
14
FSET
TOP VIEW
EXTPWR
•
•
28
VADJ
•
•
•
•
Text for space
VDAC
•
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HIDRV
•
The bq24741 charges two, three, or four series Li+
cells, supporting up to 10 A of charge current, and is
available in a 28-pin, 5x5-mm2 thin QFN package.
VREF
•
The bq24741 is a high-efficiency, synchronous
battery charger with integrated compensation,
offering low component count for space-constrained
Li-ion or Li-polymer battery charging applications.
Ratiometric charge current and voltage programming
allows high regulation accuracies, and can be either
hardwired with resistors or programmed by the
system power-management microcontroller using a
DAC or GPIOs.
BTST
•
DESCRIPTION
AGND
•
Notebook and Ultra-Mobile Computers
Portable Data Capture Terminals
Portable Printers
Medical Diagnostics Equipment
Battery Bay Chargers
Battery Back-up Systems
PVCC
•
•
NMOS-NMOS Synchronous Buck Converter
Resistor-Programmable Switching Frequency
between 300 kHz and 800 kHz
8 V-24 V Input Voltage Operation Range
Cells Pin Support Two to Four Li-Ion Cells up
to 18 V Battery Voltage
Analog Inputs with Ratiometric Programming
via Resistors or DAC/GPIO
– Charge Voltage (4-4.512 V/cell)
– Charge Current (up to 10 A)
– Adapter Current Limit for DPM
High-Accuracy Voltage and Current Regulation
– ±0.5% Charge Voltage Accuracy
– ±3% Charge Current Accuracy
– ±3% Adapter Current Accuracy
– ±2% Input Current Sense Amp Accuracy
150 mA Trickle-charge Current with ±33%
Accuracy Down to Zero Battery Voltage
Safety Protection
– Input Overvoltage Protection
– Battery Overvoltage Protection
– Charger Overcurrent Protection
– Thermal Shutdown Protection
Status and Monitoring Outputs
– Adapter Present Indicator
– Programmable Input Power Detect with
Adjustable Threshold
– Dynamic Power Management (DPM) with
Status Indicator
– Current Drawn from Input Source
Charge Enable Pin
Internal Soft-Start
Internal Loop Compensation
25 ns Minimum Driver Dead-Time and 99.5%
Maximum Effective Duty Cycle
28-pin, 5x5-mm2 QFN package
Energy Star Low Iq
– < 10 µA Off-State Battery Discharge Current
– < 1.5 mA Off-State Input Quiescent Current
TRICKL
1
1
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas
Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2009, Texas Instruments Incorporated
bq24741
SLUS875A – MARCH 2009 – REVISED MARCH 2009 ...................................................................................................................................................... www.ti.com
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
DESCRIPTION (CONTINUED)
The bq24741 features resistor-programmable PWM switching frequency and accurate 150mA trickle charge (with
20 mΩ sensing resistor), which can be enabled via the TRICKLE pin. The bq24741 also features Dynamic Power
Management (DPM) and input power limiting. These features reduce battery charge current when the input
power limit is reached to avoid overloading the AC adapter when supplying the load and the battery charger
simultaneously. A high-accuracy current sense amplifier enables accurate measurement of input current from the
AC adapter, allowing monitoring the overall system power. If the adapter current is above the programmed
low-power threshold, a signal is sent to host so that the system optimizes its power performance according to
what is available from the adapter.
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Text for space
SYSTEM
ADAPTER +
ADAPTER -
R11
2Ω
C1
2.2 µF
P
P
Q1 (ACFET) Q2 (ACFET)
SI4435
SI4435
Controlled by
HOST
C2
R1
0.1 µF
432 kΩ
1%
RAC
0.010 Ω
C6
10 µF
C7
10 µF
D2
BAT54
C3
ACN
0.1 µF
PVCC
Q3(BATFET)
SI4435
Controlled by
HOST
C8
0.1 µF
ACP
ACDET
AGND
R3
10 kΩ
Q4_A
FDS8978
HIDRV
VREF
SW
EXTPWR
EXTPWR
N
R2
66.5 kΩ
1%
BTST
VREF
R5
10 kΩ
C4
1 µF
D1
bq24741
REGN
DPMDET
GPIO
HOST
PGND
120 kΩ
ISET_PWM
(D = 0.72, Vpeak = VDAC)
R14
CE
SRP
VDAC
SRN
ISET
BAT
Q4_B
FDS8978
VREF
C13
100 nF
ADC
LPREF
IADAPT
C5
100 pF
C13
0.1 µF
CELLS
10 kΩ
VREF
R12
102 kΩ
1%
ACSET
VADJ
PowerPad
FSET
R13
64.9 kΩ
1%
C12
10 µF
C11
10 µF
LODRV
LPMOD
R15
PACK+
C10
1 µF
TRICKLE
RSR
0.020 Ω
10µH
BAT54 0.1 µF
N
R4
10 kΩ
L1
C9
P
VREF
R9
60.4 kΩ
1%
R7
73.2 kΩ
1%
PACK-
C14
0.1 µF
C15
0.1 µF
R8
26.7 kΩ
1%
R10
R6
40.2 kΩ
97.6 kΩ
1%
Text for space
FS = 400 kHz, 90 W Adapter, VADAPTER = 19 V, VBAT = 3-cell Li-Ion (4.2V/cell), Icharge = 3.6 A, Iadapter_limit = 4.0 A
Figure 1. Typical System Schematic, Voltage, and Current Programmed by Resistor
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2
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bq24741
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SYSTEM
ADAPTER +
C1
2.2 µF
RAC
0.010 Ω
P
P
Q1 (ACFET) Q2 (ACFET)
SI4435
SI4435
Controlled by
HOST
C2
R1
0.1 µF
432 kΩ
1%
D2
BAT54
C7
10 µF
C3
ACN
0.1 µF
PVCC
Q3 (BATFET)
SI4435
Controlled by
HOST
C8
0.1 µF
ACP
ACDET
R2
66.5 kΩ
1%
C6
10 µF
bq24741
AGND
EXTPWR
SW
EXTPWR
R5
10 kΩ
C4
1 µF
D1
REGN
DPMDET
LPMOD
HOST
10 kΩ
120 kΩ
ISET_PWM
R14
(D = 0.72, Vpeak = VDAC)
C13
0.1 µF
LODRV
PGND
CE
SRP
VDAC
SRN
ISET
BAT
LPREF
ACSET
IADAPT PowerPad
PACK-
C14
0.1 µF
C15
0.1 µF
R8
26.7 kΩ
1%
VADJ
ADC
Q4_B
FDS8978
VREF
R7
73.2 kΩ
1%
C13
100 nF
DAC
PACK+
C12
10 µF
C11
10 µF
CELLS
R15
RSR
0.020 Ω
4.7 µH
C10
1 µF
TRICKLE
GPIO
BAT54 0.1 µF
N
R4
10 kΩ
L1
C9
BTST
VREF
P
Q4_A
FDS8978
HIDRV
VREF
R3
10 kΩ
N
ADAPTER -
R11
2Ω
FSET
R6
56.2 kΩ
C5
100 pF
Text for space
(1) Pull-up rail could be either VREF or other system rail.
(2) SRSET/ACSET could come from either DAC or resistor dividers.
FS = 650 kHz, 90 W Adapter, VADAPTER = 19 V, VBAT = 3-cell Li-Ion (4.2V/cell), Icharge = 3.6 A, Iadapter_limit = 4.0 A
Figure 2. Typical System Schematic, Voltage and Current Programmed by DAC
ORDERING INFORMATION
Part number
Package
bq24741
28-PIN 5 x 5 mm2 QFN
Ordering Number
(Tape and Reel)
Quantity
bq24741RHDR
3000
bq24741RHDT
250
PACKAGE THERMAL DATA
PACKAGE
QFN – RHD
(1)
(2)
(1) (2)
θJA
TA = 70°C POWER RATING
DERATING FACTOR ABOVE TA = 25°C
39°C/W
2.36 W
0.028 W/°C
For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI
Web site at www.ti.com.
This data is based on using the JEDEC High-K board and the exposed die pad is connected to a Cu pad on the board. This is
connected to the ground plane by a 2x3 via matrix.
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bq24741
SLUS875A – MARCH 2009 – REVISED MARCH 2009 ...................................................................................................................................................... www.ti.com
Table 1. TERMINAL FUNCTIONS – 28-PIN QFN
TERMINAL
NAME
NO.
DESCRIPTION
CE
1
Charge-enable active-HIGH logic input. HI enables charge. LO disables charge. It has an internal 1 MΩ pull-down
resistor. A 10 KΩ external resistor is required to connect the CE pin to the external pull-up rail other than VREF.
ACN
2
Adapter current sense resistor, negative input. A 0.1 µF ceramic capacitor is placed from ACN to ACP to provide
differential-mode filtering. An optional 0.1 µF ceramic capacitor is placed from ACN pin to AGND for common-mode
filtering.
ACP
3
Adapter current sense resistor, positive input. A 0.1 µF ceramic capacitor is placed from ACN to ACP to provide
differential-mode filtering. A 0.1 µF ceramic capacitor is placed from ACP pin to AGND for common-mode filtering.
LPMOD
4
Low-power-mode-detect active-LOW open-drain logic output. Place a 10kohm pull-up resistor from LPMOD pin to the
pull-up voltage rail. The output is HI when IADAPT pin voltage is lower than LPREF pin voltage. The output is LOW
when IADAPT pin voltage is higher than LPREF pin voltage. Internal 6% hysteresis.
ACDET
5
Adapter detected voltage set input. Program the adapter detect threshold by connecting a resistor divider from adapter
input to ACDET pin to AGND pin. Adapter voltage is detected if ACDET pin voltage is greater than 2.4 V. IADAPT
current sense amplifier is active when ACDET pin voltage is greater than 0.6V and PVCC > VUVLO. ACOV is input
over-voltage protection; it disables charge when ACDET > 3.1 V. ACOV does not latch, and normal operation resumes
when ACDET < 3.1 V.
ACSET
6
Adapter current set input. The voltage ratio of ACSET voltage versus VDAC voltage programs the input current
regulation set-point during Dynamic Power Management (DPM). Program by connecting a resistor divider from VDAC
to ACSET to AGND; or by connecting the output of an external DAC to the ACSET pin and connect the DAC supply to
the VDAC pin.
LPREF
7
Low power voltage set input. Connect a resistor divider from VREF to LPREF, and AGND to program the reference for
the LOPWR comparator. The LPREF pin voltage is compared to the IADAPT pin voltage and the logic output is given
on the LPMOD open-drain pin. Connect LPREF to ACSET through a resistor divider to track the adapter power.
TRICKLE
8
Trickle current enable logic input. When CE is HIGH, a HIGH level on this pin enables accurate 150 mA trickle charge
with 20 mΩ sense resistor. A LOW level on this pin enables the ISET pin to program the charge current. It has an
internal 1 MΩ pull-down resistor.
AGND
9
Analog ground. Ground connection for low-current sensitive analog and digital signals. On PCB layout, connect to the
analog ground plane, and only connect to PGND through the PowerPad underneath the IC.
VREF
10
3.3 V regulated voltage output. Place a 1 µF ceramic capacitor from VREF to AGND pin close to the IC. This voltage
could be used for ratio-metric programming of voltage and current regulation and for programming the LPREF
threshold. VREF is also the voltage source for the interal circuit.
VDAC
11
Charge voltage set reference input. Connect the VREF or external DAC voltage source to VDAC pin. Battery voltage,
charge current, and input current are programmed as a ratio of the VDAC pin voltage versus the voltage on VADJ, and
ACSET pin voltages, respectively. Place resistor dividers from VDAC to VADJ, ISET, and ACSET pins to AGND for
programming. A DAC could be used by connecting the DAC supply to VDAC and connecting the output to VADJ,
ISET, or ACSET.
VADJ
12
Charge voltage set input. The voltage ratio of VADJ voltage versus VDAC voltage programs the battery voltage
regulation set-point. Program by connecting a resistor divider from VDAC to VADJ, to AGND; or, by connecting the
output of an external DAC to VADJ pin and connect the DAC supply to VDAC pin.
EXTPWR
13
Valid adapter active-low detect logic open-drain output. Pulled LO when Input voltage is above ACDET programmed
threshold OR input current is greater than 1.25 A with 10 mΩ sense resistor. Connect a 10 kΩ pull-up resistor from
EXTPWR pin to pull-up supply rail.
FSET
14
PWM switching frequency (Fs) program pin. Program the switching frequency by placing a resistor to AGND on this
pin.
IADAPT
15
Adapter current sense amplifier output. IADAPT voltage is 20 times the differential voltage across ACP-ACN. Place a
100 pF (max) or less ceramic decoupling capacitor from IADAPT to AGND.
ISET
16
Charge current set input. The voltage ratio of ISET voltage versus VDAC voltage programs the charge current
regulation set-point. Program by connecting a resistor divider from VDAC to ISET, to AGND; or, by connecting the
output of an external DAC to ISET pin and connect the DAC supply to VDAC pin.
BAT
17
Battery voltage remote sense. Directly connect a kelvin sense trace from the battery pack positive terminal to the BAT
pin to accurately sense the battery pack voltage. Place a 0.1 µF capacitor from BAT to AGND close to the IC to filter
high frequency noise.
CSN
18
Charge current sense resistor, negative input. A 0.1 µF ceramic capacitor is placed from CSN to CSP to provide
differential-mode filtering. An optional 0.1 µF ceramic capacitor is placed from CSN pin to AGND for common-mode
filtering.
CSP
19
Charge current sense resistor, positive input. A 0.1 µF ceramic capacitor is placed from CSN to CSP to provide
differential-mode filtering. A 0.1 µF ceramic capacitor is placed from CSP pin to AGND for common-mode filtering.
CELLS
20
2, 3 or 4 cells selection logic input. Logic Lo programs 3–cell. Logic HI programs 4-cell. Floating programs 2–cell.
4
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Table 1. TERMINAL FUNCTIONS – 28-PIN QFN (continued)
TERMINAL
NAME
NO.
DESCRIPTION
DPMDET
21
Dynamic power management (DPM) input current loop active, open-drain output status. Logic low (LO) indicates input
current is being limited by reducing the charge current. Connect 10-kohm pull-up resistor from DPMDET pin to VREF
or a different pull-up supply rail.
PGND
22
Power ground. Ground connection for high-current power converter node. On PCB layout, connect directly to source of
low-side power MOSFET, to ground connection of in put and output capacitors of the charger. Only connect to AGND
through the PowerPad underneath the IC.
LODRV
23
PWM low side driver output. Connect to the gate of the low–side power MOSFET with a short trace.
REGN
24
PWM low side driver positive 6 supply output. Connect a 1 µF ceramic capacitor from REGN to PGND pin, close to the
IC. Use for low side driver and high-side driver bootstrap voltage by connecting a small signal Schottky diode from
REGN to BTST. REGN is disabled when CE is LOW.
SW
25
PWM high side driver negative supply. Connect to the Phase switching node (junction of the low-side power MOSFET
drain, high-side power MOSFET source, and output inductor). Connect the 0.1 µF bootstrap capacitor from SW to
BTST.
HIDRV
26
PWM high side driver output. Connect to the gate of the high-side power MOSFET with a short trace.
BTST
27
PWM high side driver positive supply. Connect a 0.1 µF bootstrap ceramic capacitor from BTST to SW. Connect a
bootstrap Schottky diode from REGN to BTST. A optional 2.0Ω - 5.1Ω bootstrap resistor can be inserted between the
BTST pin and the common point of the bootstrap capactor and bootstrap diode, thus dampening the SW node voltage
ring and spike.
PVCC
28
IC power positive supply. Connect to the adapter input through a schottky diode. Place a 0.1 uF ceramic capacitor
from PVCC to PGND pin close to the IC.
PowerPad
Exposed pad beneath the IC. AGND and PGND star-connected only at the PowerPad plane. Always solder PowerPad
to the board, and have vias on the PowerPad plane connecting to AGND and PGND planes. It also serves as a
thermal pad to dissipate the heat.
ABSOLUTE MAXIMUM RATINGS
over operating free-air temperature range (unless otherwise noted) (1)
(2)
VALUE
PVCC, ACP, ACN, SRP, SRN, BAT
Voltage range
SW
–1 to 30
REGN, LODRV, VADJ, ACSET, ISET, ACDET, FSET, IADAPT, /LPMOD,
LPREF, CE, CELLS, EXTPWR, DPMDET, TRICKLE
–0.3 to 7
VDAC, VREF
–0.3 to 3.6
BTST, HIDRV with respect to AGND and PGND
–0.3 to 36
AGND, PGND
Maximum difference voltage
UNIT
–0.3 to 30
V
–1 to 1
ACP–ACN, CSP–CSN
-0.5 to 0.5
Junction temperature range
–40 to 155
°C
Storage temperature range
–55 to 155
°C
(1)
(2)
Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings
only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating
conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
All voltages are with respect to GND if not specified. Currents are positive into, negative out of the specified terminal. Consult Packaging
Section of the data book for thermal limitations and considerations of packages.
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RECOMMENDED OPERATING CONDITIONS
over operating free-air temperature range (unless otherwise noted)
MIN
SW
Voltage range
NOM
MAX
UNIT
–0.8
24
V
PVCC, ACP, ACN, CSP, CSN, BAT
0
24
V
REGN, LODRV
0
6.5
V
VREF
3.3
V
VDAC
3.6
V
VADJ, ACSET, ISET, ACDET, FSET, IADAPT, LPMOD, LPREF, CE, CELLS,
EXTPWR, DPMDET, TRICKLE
BTST, HIDRV with respect to AGND and PGND
0
5.5
V
0
30
V
AGND, PGND
–0.3
0.3
V
Maximum difference voltage: ACP–ACN, CSP–CSN
–0.3
0.3
V
Junction temperature range
–40
125
°C
Storage temperature range
–55
150
°C
ELECTRICAL CHARACTERISTICS
8.0 V ≤ VPVCC ≤ 24 V, 0°C < TJ < +125°C, Fs=600 kHz, typical values are at TA = 25°C, with respect to AGND (unless
otherwise noted) (1) (2) (3)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
OPERATING CONDITIONS
VPVCC_OP
PVCC input voltage operating range
8
24
V
0
PVCC
V
CHARGE VOLTAGE REGULATION
VBAT_OP
BAT input voltage operating range
VBAT_REG_RNG
BAT voltage regulation range
VVDAC_OP
VDAC reference voltage range
VVADJ_OP
VADJ voltage range
4-4.512 V per cell, times 2,3,4 cells
Charge voltage regulation accuracy
8
18.048
V
2.6
3.6
V
0
VDAC
V
8 V, 8.4 V, 9.024 V
–0.5
0.5
12 V, 12.6 V, 13.536 V
–0.5
0.5
16 V, 16.8 V, 18.048 V
–0.5
0.5
0
100
0
VDAC
–3%
3%
%
CHARGE CURRENT REGULATION (ENABLE CE & DISABLE TRICKLE)
VIREG_CHG
Charge current regulation differential
voltage range
VISET_OP
SRSET voltage range
VIREG_CHG = VCSP – VCSN
VIREG_CHG = 40 mV
Charge current regulation accuracy
Off-set Voltage of Amplifier
(1)
(2)
(3)
6
VIREG_CHG = 20 mV
–5%
5%
VIREG_CHG = 5 mV
–25%
25%
VIREG_CHG = 3 mV (VBAT ≥ 4 V)
–33%
33%
VIREG_CHG = 3 mV (VBAT < 4 V)
–50%
50%
VBAT ≥ 4 V
–1.0
1.0
VBAT < 4 V
–1.5
1.5
mV
V
mV
Verified by design
Deglitch time and delay are propotional to the period of oscillator, unless specified.
When CE=HIGH, the oscillator frequency is equal to external setting Fs; when CE=LOW, the oscillator frequency is fixed internal setting
700 kHz.
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ELECTRICAL CHARACTERISTICS (continued)
8.0 V ≤ VPVCC ≤ 24 V, 0°C < TJ < +125°C, Fs=600 kHz, typical values are at TA = 25°C, with respect to AGND (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
TRICKLE CHARGE CURRENT REGULATION (ENABLE CE & TRICKLE)
Charge Current Regulation Accuracy
VIREG_CHG = 3 mV
Off-set Voltage of Amplifier
–33%
33%
–1.0
1.0
mV
0
100
mV
INPUT CURRENT REGULATION
VIREG_DPM
Adapter current regulation differential
voltage range
VACSET_OP
ACSET voltage range
Input current regulation accuracy
VIREG_DPM = VACP – VACN
0
VDAC
VIREG_DPM = 40 mV
–3%
3%
VIREG_DPM = 20 mV
–5%
5%
VIREG_DPM = 5 mV
–25%
25%
VIREG_DPM = 1.5 mV
–33%
33%
-500
500
Off-set Voltage of Amplifier
V
µV
VREF REGULATOR
VVREF_REG
VREF regulator voltage
VACDET > 0.6 V, 0-30 mA
IVREF_LIM
VREF current limit
VVREF = 0 V, VACDET > 0.6 V
3.267
35
3.3
3.333
V
80
mA
6.2
V
REGN REGULATOR
VREGN_REG
REGN regulator voltage
VACDET > 0.6 V, 0-75 mA, PVCC > 10 V
5.6
IREGN_LIM
REGN current limit
VREGN = 0 V, VACDET > 0.6 V
90
5.9
145
0
24
0
2
ADAPTER CURRENT SENSE AMPLIFIER
VACP/N_OP
Input common mode range
VIADAPT
IADAPT output voltage range
Voltage on ACP/ACN
IIADAPT
IADAPT output current
AIADAPT
Current sense amplifier voltage gain
0
Adapter current sense accuracy
AIADAPT = VIADAPT / VIREG_DPM
1
20
–2%
VIREG_DPM = 20 mV
–4%
4%
VIREG_DPM = 5 mV
–25%
25%
VIREG_DPM = 1.5 mV
–33%
33%
Output current limit
VIADAPT = 0 V
CIADAPT_MAX
Maximum output load capacitance
For stability with 0 mA to 1 mA load
mA
V/V
VIREG_DPM = 40 mV
IIADAPT_LIM
V
2%
1
mA
100
pF
2.424
V
ACDET COMPARATOR (INPUT UNDER_VOLTAGE, ACVGOOD)
VACDET_CHG
ACDET adapter-detect rising
threshold
Min voltage to enable charging, VACDET
rising
VACDET_CHG_HYS
ACDET falling hysteresis
VACDET falling, PVCC>8V
40
mV
ACDET rising deglitch to turn on
EXTPWR FET (4)
VACDET rising, PVCC>8V
1.2
ms
ACDET rising deglitch to enable
charge (4)
VACDET rising, PVCC>8V, CE=HIGH
333
ms
ACDET falling deglitch to turn off
EXTPWR FET (4)
VACDET falling, PVCC>8V
80
µs
ACDET falling deglitch to disable
charge (4)
VACDET falling, PVCC>8V
80
µs
Power-up delay from VACDET>2.4V to
EXTPWR FET turn-on (4)
First time power up, Fs=300 kHz - 800
kHz
TACDET_EXTPWR
(4)
2.376
2.40
2
ms
Verified by design
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ELECTRICAL CHARACTERISTICS (continued)
8.0 V ≤ VPVCC ≤ 24 V, 0°C < TJ < +125°C, Fs=600 kHz, typical values are at TA = 25°C, with respect to AGND (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
200
250
300
mV
AC CURRENT DETECT COMPARATOR (INPUT UNDER_CURRENT, ACIGOOD)
VACIDET
Adapter current detect falling
threshold
VACI = 20 X IAC x RAC, falling edge
VACIDE_HYS
Adapter current detect hysteresis
Rising edge
50
mV
IADAPT rising
10
µs
IADAPT falling
10
µs
Adapter current detect deglitch
PVCC / BAT COMPARATOR
VPVCC_BAT_OP
Differential Voltage from PVCC to
BAT
VPVCC-BAT_FALL
PVCC to BAT falling threshold
VPVCC-BAT__HYS
PVCC to BAT hysteresis
-20
VPVCC – VBAT to disable charge
24
V
850
900
950
mV
200
225
250
mV
PVCC to BAT rising deglitch
VPVCC – VBAT > VPVCC-BAT_RISE
4.5
ms
PVCC to BAT ralling deglitch
VPVCC – VBAT < VPVCC-BAT_FALL
10
µs
BAT OVERVOLTAGE COMPARATOR
VOV_RISE
Overvoltage rising threshold (5)
As percentage of VBAT_REG
104
%
VOV_FALL
Overvoltage falling threshold (5)
As percentage of VBAT_REG
102
%
BATSHORT COMPARATOR
VBATSHORT_RISE
Battery rising voltage for BATSHORT
exit
2
V/Cell
VBATSHORT_FALL
Battery falling voltage for BATSHORT
entry
1.7
V/Cell
CHARGE OVERCURRENT COMPARATOR
VOC_peak
Peak charge over-current threshold
V(CSP-CSN), when VISET / VDAC < 0.8
V(CSP-CSN), when VISET / VDAC ≥ 0.8
90
110
130
mV
100
125
150
mV
CHARGE UNDERCURRENT PROTECTION COMPARATOR (UCP)
VUCP
Charge under-current threshold,
falling edge
V(CSP-CSN) from synchronous to
non-synchronous operation
25
30
35
mV
Charge under-current threshold,
rising edge
V(CSP-CSN) from non-synchronous to
synchronous operation
35
40
45
mV
Charge under-current rising deglitch
10
µs
Charge under-current falling deglitch
320
µs
INPUT OVERVOLTAGE COMPARATOR (ACOV)
VACOV
AC over-voltage rising threshold on
ACDET
Measure on ACDET pin
3.007
3.1
3.193
V
VACOV_HYS
AC over-voltage deglitch (rising edge)
650
µs
AC over-voltage deglitch (falling
edge)
650
µs
INPUT UNDERVOLTAGE LOCK-OUT COMPARATOR (UVLO)
VUVLO
AC under-voltage rising threshold
Measured on PVCC pin
VUVLO_HYS
AC under-voltage hysteresis, falling
7
8
9
260
V
mV
INPUT LOW POWER MODE COMPARATOR (LPMOD)
VACLP_HYS
AC low power mode comparator
internal hysteresis
VACLP_OFFSET
AC low power mode comparator
offset voltage
(5)
8
5%
7%
1
mV
Verified by design
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ELECTRICAL CHARACTERISTICS (continued)
8.0 V ≤ VPVCC ≤ 24 V, 0°C < TJ < +125°C, Fs=600 kHz, typical values are at TA = 25°C, with respect to AGND (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
THERMAL SHUTDOWN COMPARATOR
TSHUT
Thermal shutdown rising temperature
TSHUT_HYS
Thermal shutdown hysteresis, falling
Temperature Increasing
155
°C
20
°C
PWM HIGH SIDE DRIVER (HIDRV)
RDS_HI_ON
High side driver turn-on resistance
VBTST – VPH = 5.5 V, tested at 100 mA
6
Ω
RDS_HI_OFF
High side driver turn-off resistance
VBTST – VPH = 5.5 V, tested at 100 mA
1.4
Ω
VBTST_REFRESH
Bootstrap refresh comparator
threshold voltage
VBTST – VPH when low side refresh pulse
is requested
IBTST_LEAK
BTST leakage current
High side is on; charge enabled
4
V
200
µA
PWM LOW SIDE DRIVER (LODRV)
RDS_LO_ON
Low side driver turn-on resistance
REGN = 6 V, tested at 100 mA
6
Ω
RDS_LO_OFF
Low side driver turn-off resistance
REGN = 6 V, tested at 100 mA
1.2
Ω
PWM DRIVERS TIMING
Driver Dead Time between HIDRV
and LODRV
25
ns
PWM OSCILLATOR
FS
Programmable PWM switching
frequency range
RFSET=130 kΩ - 45 kΩ
300
PWM switching frequency accuracy
800
-20%
kHz
20%
RAMP amplitude
1.33
V
DC offset of RAMP
300
mV
QUIESCENT CURRENT
Total off-state quiescent current into
pins: CSP, CSN, BAT, VCC, BTST,
SW, PVCC, ACP, ACN
VBAT = 16.8 V, VACDET < 0.6 V,
VPVCC > 8 V, TJ = 0 to 125°C
Total off-state battery current from
ACP, ACN
VBAT = 16.8 V, VACDET < 0.6 V,
VPVCC > 8 V, TJ = 0 to 125°C
Battery on-state quiescent current
VBAT = 16.8 , 0.6 V < VACDET < 2.4 V,
VPVCC > 8V
IBATQ_CD
Total quiescent current into CSP,
CSN, BAT, VCC, BTST, SW
Adapter present, VACDET > 2.4 V, charge
disabled
IAC
Adapter quiescent current
VPVCC = 20 V, charge disabled
IOFF_STATE
IBAT_ON
7
11
µA
1
µA
1
mA
100
200
µA
1
1.5
mA
INTERNAL SOFT START (8 steps to regulation current)
Soft start steps
8
Soft start time of each step (512
PWM cycles)
step
µs
853
LOGIC INPUT PIN CHARACTERISTICS (CE, TRICKLE)
VIN_LO
Input low threshold voltage
VIN_HI
Input high threshold voltage
0.8
RPULLDOWN
PIN pull down resistance inside IC
V = 0 to VREGN
1
MΩ
TCE_ENCHARGE
Delay from CE=HIGH to charge
enable
Fs=300 kHz - 800 kHz
2
ms
2.1
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ELECTRICAL CHARACTERISTICS (continued)
8.0 V ≤ VPVCC ≤ 24 V, 0°C < TJ < +125°C, Fs=600 kHz, typical values are at TA = 25°C, with respect to AGND (unless
otherwise noted)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNIT
LOGIC INPUT PIN CHARACTERISTICS (CELLS)
VIN_LO
Input low threshold voltage, 3 cells
CELLS voltage falling edge
Input mid threshold voltage, 2 cells
CELLS voltage rising for MIN,
CELLS voltage falling for MAX
0.8
VIN_HI
Input high threshold voltage, 4 cells
CELLS voltage rising
2.5
IBIAS_FLOAT
Input bias float current for 2 cell
selection
VCE = 0 to VREGN
–1
VIN_MID
0.5
1.8
V
1
µA
OPEN-DRAIN LOGIC OUTPUT PIN CHARACTERISTICS ( EXTPWR, DPMDET, LPMOD)
VOUT_LO
Output low saturation voltage
Sink Current = 5 mA
Leakage current
Pull up to 3.3 v
DPMDET delay, both edge
10
5
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V
1
µA
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TYPICAL CHARACTERISTICS
Table of Graphs (1) Fs=400 kHz, Ta = 25 °C
Y
X
Figure
VREF Load and Line Regulation
vs Load Current
Figure 3
REGN Load and Line Regulation
vs Load Current
Figure 4
BAT Voltage
vs VADJ/VDAC Ratio
Figure 5
BAT Voltage Regulation Accuracy
vs Setpoint
Figure 6
Charge Current
vs ISET/VDAC Ratio
Figure 7
Charge Current Regulation Accuracy
vs V(CSP-CSN) Setpoint
Figure 8
Input Current
vs ACSET/VDAC Radio
Figure 9
DPM Accuracy
vs V(ACP-ACN) Setpoint
Figure 10
BAT Voltage Regulation Accuracy
vs Charge Current
Figure 11
V_IADAPT Accuracy
vs V(ACP-ACN) Voltage
Figure 12
Trickle Charge Current
vs BAT Voltage
Figure 13
DPM and Charge Current
vs System Current
Figure 14
REF, REGN, and EXTPWR Startup (CE=HIGH)
Figure 15
Transient System Load (DPM) Response Transistion
Figure 16
Transient Response of IADAPT and LPMOD
Figure 17
Battery Overcurrent Protection (OCP)
Figure 18
Battery to Ground Short Transistion
Figure 19
Battery to Ground Short Protection
Figure 20
Charge Enable and Current Soft-Start
Figure 21
Charge Disable
Figure 22
Trickle Disable and Current Soft-Start
Figure 23
Synchronous to Non-synchronous Transistion
Figure 24
Non-synchronous to Synchronous Transistion
Figure 25
Continuous Conduction Mode Switching Waveforms
Figure 26
Efficiency
vs Battery Charge Current
Figure 27
Switch Frequency
vs Setting Resistor
Figure 28
(1)
Test results based on Figure 2 application schematic. VIN = 20 V, VBAT = 3-cell LiIon, ICHG = 3 A, IADAPTER_LIMIT = 4 A, TA = 25°C, unless
otherwise specified.
VREF LOAD AND LINE REGULATION
vs
Load Current
REGN LOAD AND LINE REGULATION
vs
LOAD CURRENT
0
0.50
-0.50
Regulation Error - %
Regulation Error - %
0.40
0.30
PVCC = 10 V
0.20
0.10
0
-1
-1.50
PVCC = 10 V
-2
PVCC = 20 V
-2.50
-0.10
PVCC = 20 V
-0.20
-3
0
10
20
30
VREF - Load Current - mA
40
50
0
Figure 3.
10
20
30
40
50
60
REGN - Load Current - mA
70
80
Figure 4.
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BAT VOLTAGE
vs
VADJ/VDAC RATIO
BAT VOLTAGE REGULATION ACCURACY
vs
SETPOINT
13.6
0.06
3-Cell
0.04
13.2
Regulation Error (%)
Voltage Regulation - V
13.4
0.05
13
12.8
12.6
12.4
0.02
0.01
0
-0.01
12.2
12
0.03
0
0.1 0.2 0.3
0.4 0.5
0.6
0.7 0.8 0.9
-0.02
12
1
12.2
12.4
12.6
13
12.8
13.2
13.4
13.6
VBAT_reg Setpoint (V)
VADJ/VDAC Ratio
Figure 5.
Figure 6.
CHARGE CURRENT
vs
ISET/VDAC
CHARGE CURRENT REGULATION ACCURACY
vs
V(CSP-CSN) SETPOINT
5
25
4.5
20
3.5
Regulation Error - %
Input Current Regulation - A
3-Cell
4
3
2.5
2
1.5
1
15
10
5
0.5
0
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
0
1
0
10
20
30
ACSET/VDAC Ratio
40
50
60
70
80
90
100
80
90
100
ICHG_reg Setpoint (mV)
Figure 7.
Figure 8.
INPUT CURRENT
vs
ACSET/VDAC RATIO
DPM ACCURACY
vs
V(ACP-ACN) SETPOINT
2
10
9
0
Regulation Error - %
Input Current Regulation - A
3-Cell
8
7
6
5
4
3
2
-2
-4
-6
-8
1
-10
0
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
0
10
VACSET/VDAC
Figure 9.
12
20
30
40
50
60
70
IIN_reg Setpoint (mV)
Figure 10.
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BAT VOLTAGE REGULATION ACCURACY
vs
CHARGE CURRENT
V_IADAPT ACCURACY
vs
V(ACP-ACN) VOLTAGE
0.030
0.00%
0.025
-1.00%
V_IADAPT Error
Regulation Voltage Accuracy (%)
-0.50%
0.020
0.015
0.010
-1.50%
-2.00%
-2.50%
-3.00%
-3.50%
-4.00%
0.005
-4.50%
-5.00%
0.000
0
0.5
1.5
1
3
2.5
2
3.5
0
4
30
20
40
50
60
70
80
V(ACP-ACN) (mV)
Figure 11.
Figure 12.
TRICKLE CHARGE CURRENT
vs
BAT VOLTAGE
DPM and CHARGE CURRENT
vs
SYSTEM CURRENT
90
100
4.5
0.165
4
VBAT 0 V to 12.6 V
3.5
0.16
Ichrg & Iin (A)
Trickle Charge Current (A)
10
Charge Current (A)
0.155
Input Current 3 A
3
2.5
2
Charge Current 3 A
1.5
0.15
1
VBAT 12.6 V to 0 V
0.5
0.145
0
0
2
4
6
8
10
12
14
0
BAT Voltage (V)
0.5
1
1.5
2
2.5
3
Figure 13.
Figure 14.
REF, REGN, and EXTPWR
STARTUP (CE=HIGH)
TRANSIENT SYSTEM LOAD
(DPM) RESPONSE TRANSISTION
2 A/div
20 V/div
3.5
4
System Current (A)
Isys
5 v/div 2 V/div
2 A/div
2 V/div
2 A/div
EXTPWR
IIN
Ibat
Time = 200 μs/div
Time =400 μs/div
Figure 15.
Figure 16.
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5 V/div
BATTERY OVERCURRENT PROTECTION
(OCP)
2 A/div
IL
20 V/div
LPMOD
SW
5 v/div
Iadapt
LODRV
Time = 20 μs/div
Figure 17.
Figure 18.
BATTERY TO GROUND
SHORT TRANSISTION
BATTERY TO GROUND
SHORT PROTECTION
2 A/div
Time = 100 μs/div
IL
Vbat
IL
VBAT
SW
20 V/div
5 v/div 20 V/div
SW
5 V/div
VBAT
ISYS
10 V/div
10 V/div
2 A/div
2 V/div
0.2 V/div
2 A/div
TRANSIENT RESPONSE
of
IADAPT and LPMOD
LODRV
LODRV
Time = 10 μs/div
Time = 2 ms/div
CE
IL
2 A/div
20 V/div
5 V/div 20 V/div
2 V/div
CHARGE DISABLE
2 A/div
CHARGE ENABLE
and
CURRENT SOFT-START
20 V/div
Figure 20.
5 V/div
Figure 19.
SW
LODRV
Time = 4 μs/div
Time = 2 ms/div
Figure 21.
14
Figure 22.
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5 V/div
TRICKLE DISABLE
and
CURRENT SOFT-START
SYNCHRONOUS
to
NON-SYNCHRONOUS TRANSIsTION
TRICKLE
5 V/div
LODRV
LODRV
1 A/div
1 A/div
20 V/div
SW
5 V/div
10 V/div
SW
IL
IL
Time = 2 ms/div
Time = 2 μs/div
Figure 23.
Figure 24.
NON-SYNCHRONOUS
to
SYNCHRONOUS TRANSISTION
CONTINUOUS CONDUCTION MODE
SWITCHING WAVEFORMS
HIDRV
2 A/div 20 V/div 5 V/div
5 V/div
10 V/div
10 V/div
SW
1 A/div
LODRV
IL
LODRV
SW
IL
Time = 200 ns/div
Time = 2 μs/div
Figure 25.
Figure 26.
EFFICIENCY
vs
BATTERY CHARGE CURRENT
SWITCH FREQUENCY
vs
SETTING RESISTOR
1200
100
4-Cell 16.8 V
Measurment
Calculation
1000
800
Fsw (kHz)
Efficiency (%)
95
3-Cell 12.6 V
90
600
400
85
200
0
80
0
0.5
1
1.5
2.5
2
Charge Current (A)
3
3.5
4
0
Figure 27.
50
100
150
R_FET (kOhm)
200
250
300
Figure 28.
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FUNCTIONAL BLOCK DIAGRAM
ENA_BIAS_CMP
-
0.6V
EXTPWR
ACGOOD
+
VREFGOOD
3.3V
LDO
ENA_BIAS
VAC20X
PVCC
250mV +-
+
-
UVLO
AC_IGOOD
PVCC-BAT
ACP
FBO
+
20X
-
BAT
900mV
IIN_ER
IIN_REG
-
EAO
VAC20X
PVCC
+
EAI
+
-
ACDET
VREF
AC_VGOOD
-
2.4V
+
CE
COMP
ERROR
AMPLIFIER
+
ACN
1 MΩ
BTST
CE
BAT_ER
-
BAT
VBAT_REG
1V
+
LEVEL
SHIFTER
+
HIDRV
CSP
20 µA
3.5 mA
+
20X
-
VSR20X
ICH_ER
-
BAT_SHORT
+
CSN
DC-DC
CONVERTER
PWM LOGIC
SW
PVCC-BAT
3.5 mA
SYNCH
20 µA
PVCC
AC_VGOOD
CHRG_ON
REGN
6V LDO
CLK
IBAT_ REG
60mV
VREFGOOD
CE
TRICKLE
-
BTST
+
4V _
OSC
CLK
SW
ACSET
PGND
IC Tj
+
155 °C
-
TSHUT
+
20x
-
VAC20X
SRSET
VBATSET
IBATSET
IINSET
VADJ
LODRV
+
1 MΩ
FSET
REFRESH
CBTST
RATIO
PROGRAM
VBAT_REG
IBAT_REG
104% X VBAT_REG
-
BAT
+
2.08 V / 2.5 V
-
VSR20X
+
ACDET
+
V(IADAPT)
IADAPT
BAT_OVP
IIN_REG
DPMDET
DPM_LOOP_ON
CHG_OCP
VDAC
CELLS
+
ACOV
-
VSR20X
+
30mV
-
1.7 V
+
BAT
-
SYNCH
3.1V VAC20X
LPREF
+
PVCC
LPMOD
-
UVLO
BAT_SHORT
AGND
+
8V +-
PGND
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DETAILED DESCRIPTION
Converter Operation
The synchronous buck PWM converter uses a programmable-frequency (300 kHz to 800 kHz) voltage mode
control scheme. A type III compensation network allows using ceramic capacitors at the output of the converter.
The compensation input stage is connected internally between the feedback output (FBO) and the error amplifier
input (EAI). The feedback compensation stage is connected between the error amplifier input (EAI) and error
amplifier output (EAO). The LC output filter should be selected to give a nominal resonant frequency within 8 kHz
to 12.5 kHz to have good loop compensation.
Where resonant frequency, fo, is give by:
1
fo =
2p Lo Co
(1)
An internal saw-tooth ramp is compared to the internal EAO error control signal to vary the duty-cycle of the
converter. The ramp height is fixed 1.33 V. The ramp is offset by 300 mV in order to allow zero percent
duty-cycle, when the EAO signal is below the ramp. The EAO signal is also allowed to exceed the saw-tooth
ramp signal in order to get a 100% duty-cycle PWM request. Internal gate drive logic allows achieving 99.98%
duty-cycle while ensuring the N-channel upper device always has enough voltage to stay fully on. If the BTST pin
to SW pin voltage falls below 4 V, then the high-side n-channel power MOSFET is turned off and the low-side
n-channel power MOSFET is turned on to pull the SW node down and recharge the BTST capacitor. Then the
high-side driver returns to 100% duty-cycle operation until the (BTST-SW) voltage is detected to fall low again
due to leakage current discharging the BTST capacitor below the 4 V, and the reset pulse is reissued.
The oscillator keeps tight control of the switching frequency under all conditions of input voltage, battery voltage,
charge current, and temperature, simplifying output filter design and keeping it out of the audible noise region.
The charge current sense resistor RSR should be placed with at least half or more of the total output capacitance
placed before the sense resistor contacting both sense resistor and the output inductor; and the other half or
remaining capacitance placed after the sense resistor. The output capacitance should be divided and placed
onto both sides of the charge current sense resistor. A ratio of 50:50 percent gives the best performance; but the
node in which the output inductor and sense resistor connect should have a minimum of 50% of the total
capacitance. This capacitance provides sufficient filtering to remove the switching noise and give better current
sense accuracy. The type III compensation provides Phase boost near the cross-over frequency, giving sufficient
Phase margin.
Synchronous and Non-Synchronous Operation
The charger operates in non-synchronous mode when the sensed charge current is below the charge
under-current comparator threshold (30 mV). Otherwise, the charger operates in synchronous mode. This part is
designed for 20 mΩ charge current sense resistor and the SYNC/NON-SYNC threshold is 1.5 A. If 10 mΩ is
used, the SYNC/NON-SYNC threshold will be 3 A.
During synchronous mode, the low-side n-channel power MOSFET is on, when the high-side n-channel power
MOSFET is off. The internal gate drive logic ensures there is break-before-make switching to prevent
shoot-through currents. During the 25 ns dead time where both FETs are off, the back-diode of the low-side
power MOSFET conducts the inductor current. Having the low-side FET turn-on keeps the power dissipation low,
and allows safely charging at high currents. During synchronous mode the inductor current is always flowing and
operates in Continuous Conduction Mode (CCM), creating a fixed two-pole system.
During non-synchronous operation, the low side MOSFET will stay off during the off-time unless the voltage on
the bootstrap capacitor drops below 4 V. If this occurs, the high side FET will be turned off and the 80ns low-side
MOSFET recharge pulse will be initiated. The 80 ns pulse pulls the SW node (connection between high and
low-side MOSFET) down, allowing the bootstrap capacitor to recharge up to the REGN LDO value. After the 80
ns, the low-side MOSFET is kept off to prevent negative inductor current from occurring. The inductor current is
blocked by the off low-side MOSFET, and the inductor current will become discontinuous. This mode is called
Discontinuous Conduction Mode (DCM).
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During the DCM mode the loop response automatically changes and has a single pole system at which the pole
is proportional to the load current, because the converter does not sink current, and only the load provides a
current sink. This means at very low currents the loop response is slower, as there is less sinking current
available to discharge the output voltage. At very low currents during non-synchronous operation, there may be a
small amount of negative inductor current during the 80ns recharge pulse. The charge should be low enough to
be absorbed by the input capacitance.
Whenever the converter goes into zero percent duty-cycle, the high-side MOSFET does not turn on, and the
low-side MOSFET does not turn on (no 80ns recharge pulse) either, and there is no discharge from the battery.
Battery Voltage Regulation
The bq24741 uses a high-accuracy voltage regulator for charging voltage. The regulation voltage is ratio-metric
with respect to VDAC. The ratio of VADJ and VDAC provides extra 12.5% adjust range on VBATT regulation
voltage. By limiting the adjust range to 12.5% of the regulation voltage, the external resistor mismatch error is
reduced from ±1% to ±0.1%. Therefore, an overall voltage accuracy as good as 0.5% is maintained, while using
1% mis-match resistors. Ratio-metric conversion also allows compatibility with D/As or microcontrollers (µC). The
battery voltage is programmed through VADJ and VDAC using the following equation:
é
æ
öù
V
VBATT = cell count ´ ê 4 V + ç 0.512 ´ VADJ ÷ ú
V
VDAC ø ú
è
ëê
û
(2)
The input voltage range of VDAC is between 2.6V and 3.6V. VADJ is set between 0 and VDAC.
CELLS pin is the logic input for selecting cell count. Connect CELLS to charge 2, 3, or 4 Li+ cells. When
charging other cell chemistries, use CELLS to select an output voltage range for the charger.
Table 2. Cell-Count Selection
CELLS
CELL COUNT
Float
2
AGND
3
VREF
4
The per-cell battery termination voltage is function of the battery chemistry. Consult the battery manufacturer to
determine this voltage.
The BAT pin is used to sense the battery voltage for voltage regulation and should be connected as close to the
battery as possible, or directly on the output capacitor. A 0.1µF ceramic capacitor from BAT to AGND is
recommended to be as close to the BAT pin as possible to decouple high frequency noise.
Battery Current Regulation
The ISET input sets the maximum charging current. Battery current is sensed by resistor RSR connected
between CSP and CSN. The full-scale differential voltage between CSP and CSN is 100 mV. Thus, for a 0.020 Ω
sense resistor, the maximum charging current is 5 A. ISET is ratio-metric with respect to VDAC using the
following equation:
ICHARGE =
VISET
0.10
´
VVDAC R SR
The input voltage range of ISET is between 0 and VDAC, up to 3.6 V.
The CSP and CSN pins are used to sense across RSR with default value of 20 mΩ. However, resistors of other
values can also be used. For a larger the sense resistor, you get a larger sense voltage, and a higher regulation
accuracy; but, at the expense of higher conduction loss.
18
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Trickle Charge Current Regulation
The TRICKLE pin is provided to allow accurate current regulation at very low charge current. When CE is set to
HIGH, a logic HIGH level is applied to the TRICKLE pin, the charger will regulate 3 mV from CSP to CSN (150
mA with a 20 mΩ sense resistor), regardless of the voltage applied to the ISET pin. When TRICKLE is LOW,
ISET is used to program the charge current.
Input Adapter Current Regulation
The total input current from an AC adapter or other DC sources is a function of the system supply current and
the battery charging current. System current normally fluctuates as portions of the systems are powered up or
down. Without Dynamic Power Management (DPM), the source must be able to supply the maximum system
current and the maximum charger input current simultaneously. By using DPM, the input current regulator
reduces the charging current when the input current exceeds the input current limit set by ACSET. The current
capacity of the AC adapter can be lowered, reducing system cost.
Similar to setting battery-regulation current, adapter current is sensed by resistor RAC connected between ACP
and ACN. Its maximum value is set by ACSET, which is ratiometric with respect to VDAC, using Equation 3.
IADAPTER =
VACSET 0.10
´
VVDAC R AC
(3)
The input voltage range of ACSET is between 0 and VDAC, up to 3.6 V.
The ACP and ACN pins are used to sense RAC with a default value of 10 mΩ. However, resistors of other values
can also be used. A larger sense-resistor value yields a larger sense voltage, and a higher regulation accuracy.
However, this is at the expense of a higher conduction loss.
Adapter Detect and Power Up
An external resistor voltage divider attenuates the adapter voltage before it goes to ACDET. The adapter detect
threshold should typically be programmed to a value greater than the maximum battery voltage and lower than
the minimum allowed adapter voltage. The ACDET divider should be placed before the ACFET in order to sense
the true adapter input voltage whether the ACFET is on or off.
If ACDET is below 0.6 V but PVCC is above 8 V, part of the bias is enabled, including a crude bandgap
reference, IADAPT is disabled and pulled down to GND. The total quiescent current is less than 10 µA.
Once ACDET rises above 0.6 V and PVCC is above 8 V, all the bias circuits are enabled and VREF goes to 3.3
V; and REGN output goes to 6 V if CE is HIGH. IADAPT becomes valid to proportionally reflect the adapter
current.
When ACDET keeps rising and passes 2.4 V, a valid AC adapter is present. 8 ms later, charge is allowed to turn
on.
Programming the PWM Switching Frequency
To program the PWM switching frequency, place a resistor from the FSET pin to ground, according to the
following formula:
R FSET =
41 ´ 103
- 6.25 (k W )
Fs
Where RFSET (kΩ) is the resistor from the FSET pin to ground, and Fs (kHz) is the desired switching frequency.
The switching frequency should be programmed between 300 kHz and 800 kHz.
Enable and Disable Charging
The following conditions must be valid before the charge function is enabled:
• CE is HIGH
• Adapter is detected
• Adapter voltage is higher than PVCC-BAT threshold
• Adapter is not over voltage
• The VREF and REGEN regulators are above 90% of the final values
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•
•
Thermal Shut (TSHUT) is not active
The PWM frequency is programmed inside the allowable range
There’s a 8ms charge enable delay from when adapter is detected to when the charger is allowed to turn on.
One of the following conditions will stop on-going charging:
• CE is LOW
• Adapter is removed
• Adapter Voltage is lower than PVCC-BAT threshold
• Adapter is over voltage
• Adapter is over current
• TSHUT IC temperature threshold is reached (155 °C on rising-eduge with 20 °C hysteresis).
Automatic Internal Soft-Start Charger Current
The charger automatically soft-starts the charger regulation current every time the charger is enabled to ensure
there is no overshoot or stress on the output capacitors or the power converter. The soft-start consists of
stepping-up the charge regulation current into 8 evenly divided steps up to the programmed charge current. Each
step lasts around 1ms, for a typical rise time of 8ms. No external components are needed for this function.
High Accuracy IADAPT Using Current Sense Amplifier (CSA)
An industry standard, high accuracy current sense amplifier (CSA) is used to monitor the input current by the
host or some discrete logic through the analog voltage output of the IADAPT pin. The CSA amplifies the input
sensed voltage of ACP–ACN by 20x through the IADAPT pin. The IADAPT output is a voltage source 20 times
the input differential voltage. Once PVCC is above 8 V and ACDET is above 0.6 V, IADAPT no longer stays at
ground, but becomes active. If the user wants to lower the voltage, they could use a resistor divider from IOUT to
AGND, and still achieve accuracy over temperature as the resistors can be matched their thermal coefficients.
Input Overvoltage Protection (ACOV)
ACOV provides protection to prevent system damage due to high input voltage. Once the adapter voltage is 30%
above adapter detect voltage, (ACDET pin voltage is 30% above 2.4 V (2.4 V X 130% = 3.1 V), charge is
disabled. ACOV does not latch, and normal operation resumes when ACDET < 3.1 V.
Input Undervoltage Lock Out (UVLO)
The system must have a minimum 8 V PVCC voltage to allow proper operation. This PVCC voltage could come
from either input adapter or battery, using a diode-OR input. When the PVCC voltage is below 8 V the bias
circuits REGN and VREF stay inactive, even with ACDET above 0.6 V.
Battery Overvoltage Protection
The converter will not allow the high-side FET to turn-on until the BAT voltage goes below 102% of the regulation
voltage. This allows one-cycle response to an over-voltage condition – such as occurs when the load is removed
or the battery is disconnected.
Charge Overcurrent Protection
The charger has a secondary over-current protection. It monitors the charge current, and prevents the current
from exceeding 6.25A peak value with a 20 mΩ sensing resistor. The high-side gate drive turns off when the
over-current is detected, and automatically resumes at the next switching cycle that occurs after the current falls
below the OCP threshold. When the BAT-GND short is detected, the charger will be automatically shut down
immediately and then restarts again 100 µs later..
Thermal Shutdown Protection
The QFN package has low thermal impedance, which provides good thermal conduction from the silicon to the
ambient, to keep junctions temperatures low. As added level of protection, the charger converter turns off and
self-protects whenever the junction temperature exceeds the TSHUT threshold of 155 °C. The charger stays off
until the junction temperature falls below 135 °C.
20
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Input Low Power Detection
In order to optimize the system performance, the HOST keeps an eye on the adapter current. Once the adapter
current is above a threshold set via LPREF, the LPMOD pin sends a signal to the HOST. The signal alarms the
host that input power has exceeded the programmed limit. The LPMOD pin is an open-drain output. Connect a
pull-up resistor to LPMOD. The LPMOD output is logic LOW when the 20X current sense voltage (20 x
V(ACP-ACN)) is higher than the LPREF input voltage. The LPREF threshold may be set by an external resistor
divider using VREF, or may be programmed from a resistor divider off of ACSET to maintain an LPREF voltage
proportional to the adapter current. The LPMOD comparator has an internal 6% hysteresis built in.
ACDET
2.4 V
+
-
AC_VGOOD
ACVDET
Comparator
EXTPWR
ACP
1 kΩ
Adaptor
Current Sense
Amplifier
+
-
ACN
ACIDET
Comparator
250 mV
AC_IGOOD
(1.25 A)
+
IADAPT Error
Amplifier
Disable
20 kΩ
20xV(ACP-ACN)
+
IADAPT
IADAPT
Disable
IADAPT
OUTPUT
BUFFER
LPMOD
Comparator
+
LPREF
-
LPMOD
LOPWR_DET
Hysteresis = 6%
Figure 29. EXTPWR, LPREF, and LPMOD Logic
Status Outputs (EXTPWR, LPMOD, DPMDET Pin)
Three status outputs are available, and they all require external pull up resistors to pull the pins to system digital
rail for a high level.
EXTPWR open-drain output goes low under each of the three conditions:
1. ACDET is above 2.4 V
2. Adapter current is above 1.25 A using a 10mohm sense resistor (IADAPT voltage above 250 mV)
Internally, the AC current detect comparator looks between the output of the 20x adapter current amplifier and an
internal 250mV threshold. EXTPWR indicates a good adapter is connected because of valid voltage or current.
LPMOD output goes low when the input current is higher than the programmed threshold via LPREF pin.
Hysteresis is internally set to 6% of the programmed LPMOD threshold.
DPMDET open-drain output goes low when the DPM loop is active to reduce the battery charge current.
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Table 3. Component List for Typical System Circuit of Figure 1
PART DESIGNATOR
QTY
DESCRIPTION
Q1, Q2, Q3
3
P-channel MOSFET, -30V,-6A, SO-8, Vishay-Siliconix, Si4435
Q4
1
N-channel Dual-MOSFET, 30V, 7.5A, SO-8, Fairchild, FDS8978
D1, D2
2
Diode, Dual Schottky, 30V, 200mA, SOT23, Fairchild, BAT54C
RAC
1
Sense Resistor, 10mΩ, 1%, 1W, 2010, Vishay-Dale, WSL2010R0100F
RSR
1
Sense Resistor, 20mΩ, 1%, 1W, 2010, Vishay-Dale, WSL2010R0200F
L1
1
Inductor, 10µH, 24.8mΩ Vishay-Dale, IHLP5050CE-01
C1
1
Capacitor, Ceramic, 2.2µF, 35V, 20%, X5R, 1206, Panasonic, ECJ-3YB1E225M
C6, C7, C11, C12
4
Capacitor, Ceramic, 10µF, 35V, 20%, X5R, 1206, Panasonic, ECJ-3YB1E106M
C4, C10
2
Capacitor, Ceramic, 1µF, 25V, 10%, X7R, 2012, TDK, C2012X7R1E105K
C13
1
Capacitor, Ceramic, 100nF, 25V, 10%, X7R, 0805, Kemet, C0805C104K5RACTU
C2, C3, C8, C9, C13, C14, C15, C16
6
Capacitor, Ceramic, 0.1µF, 50V, 10%, X7R, 0805, Kemet, C0805C104K5RACTU
C5
1
Capacitor, Ceramic, 100pF, 25V, 10%, X7R, 0805, Kemet, C0805C101K5RACTU
R3, R4, R5, R15
4
Resistor, Chip, 10kΩ, 1/16W, 5%, 0402
R1
1
Resistor, Chip, 464kΩ, 1/16W, 1%, 0402
R2
1
Resistor, Chip, 66.5kΩ, 1/16W, 1%, 0402
R6
1
Resistor, Chip, 97.6kΩ, 1/16W, 1%, 0402
R7
1
Resistor, Chip, 73.2kΩ, 1/16W, 1%, 0402
R8
1
Resistor, Chip, 26.7kΩ, 1/16W, 1%, 0402
R14
1
Resistor, Chip, 120kΩ, 1/16W, 1%, 0402
R11
1
Resistor, Chip, 2Ω, 1W, 5%, 2012
R9
1
Resistor, Chip, 60.4KΩ, 1/16W, 1%, 0402
R10
1
Resistor, Chip, 40.2KΩ, 1/16W, 1%, 0402
R12
1
Resistor, Chip, 102KΩ, 1/16W, 1%, 0402
R13
1
Resistor, Chip, 64.9KΩ, 1/16W, 1%, 0402
22
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APPLICATION INFORMATION
PCB Layout Design Guideline
1. It is critical that the exposed power pad on the backside of the IC package be soldered to the PCB ground.
Ensure that there are sufficient thermal vias directly under the IC, connecting to the ground plane on the
other layers.
2. The control stage and the power stage should be routed separately. At each layer, the signal ground and the
power ground are connected only at the power pad.
3. The AC current-sense resistor must be connected to ACP (pin 3) and ACN (pin 2) with a Kelvin contact. The
area of this loop must be minimized. An additional 0.1µF decoupling capacitor for ACN is reqired to further
reduce the noise. The decoupling capacitors for these pins should be placed as close to the IC as possible.
4. The charge-current sense resistor must be connected to SRP (pin 19), SRN (pin 18) with a Kelvin contact.
The area of this loop must be minimized. An additional 0.1µF decoupling capacitor for SRN is required to
further reduce the noise. The decoupling capacitors for these pins should be placed as close to the IC as
possible.
5. Decoupling capacitors for PVCC (pin 28), VREF (pin 10), REGN (pin 24) should be placed underneath the IC
(on the bottom layer) with the interconnections to the IC as short as possible.
6. Decoupling capacitors for BAT (pin 17), IADAPT (pin 15) must be placed close to the corresponding IC pins
with the interconnections to the IC as short as possible.
7. Decoupling capacitor CX for the charger input must be placed very close to the MOSFET's drain and source.
Refer to the EVM design (SLUU284) for the recommended component placement with trace and via locations.
For the QFN information, please refer to SCBA017 and SLUA271.
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23
PACKAGE OPTION ADDENDUM
www.ti.com
3-Apr-2009
PACKAGING INFORMATION
Orderable Device
Status (1)
Package
Type
Package
Drawing
Pins Package Eco Plan (2)
Qty
BQ24741RHDR
ACTIVE
QFN
RHD
28
3000 Green (RoHS &
no Sb/Br)
CU NIPDAU
Level-2-260C-1 YEAR
BQ24741RHDT
ACTIVE
QFN
RHD
28
250
CU NIPDAU
Level-2-260C-1 YEAR
Green (RoHS &
no Sb/Br)
Lead/Ball Finish
MSL Peak Temp (3)
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check
http://www.ti.com/productcontent for the latest availability information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and
package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS
compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame
retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
Addendum-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
1-Apr-2009
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
Diameter Width
(mm) W1 (mm)
A0 (mm)
B0 (mm)
K0 (mm)
P1
(mm)
W
Pin1
(mm) Quadrant
BQ24741RHDR
QFN
RHD
28
3000
330.0
12.4
5.3
5.3
1.5
8.0
12.0
Q2
BQ24741RHDT
QFN
RHD
28
250
180.0
12.4
5.3
5.3
1.5
8.0
12.0
Q2
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
1-Apr-2009
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
BQ24741RHDR
QFN
RHD
28
3000
346.0
346.0
29.0
BQ24741RHDT
QFN
RHD
28
250
190.5
212.7
31.8
Pack Materials-Page 2
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