19-6178; Rev 0; 1/12 EVALUATION KIT AVAILABLE MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators General Description The MAX17595/MAX17596/MAX17597 is a family of peakcurrent-mode controllers which contain all the circuitry required for the design of wide input-voltage flyback and boost regulators. The MAX17595 offers optimized input rising and falling thresholds for universal input AC-DC converters and telecom DC-DC (36V–72V input range) power supplies. The MAX17596 offers input rising and falling thresholds suitable for low-voltage DC-DC applications (4.5V–36V input range). The MAX17597 offers all circuitry needed to implement a boost converter controller. All three controllers contain a built-in gate driver for external n-channel MOSFETs. The MAX17595/MAX17596/MAX17597 house an internal error amplifier with 1% accurate reference, useful in implementations without the need for an external reference. The switching frequency is programmable from 100kHz to 1MHz with an accuracy of 8% using an external resistor, allowing optimization of magnetic and filter components, resulting in compact and cost-effective power conversion solutions. For EMI sensitive applications, the MAX17595/MAX17596/MAX17597 family incorporates a programmable-frequency dithering scheme, enabling low-EMI spread-spectrum operation. An EN/UVLO input allows the user to start the power supply precisely at the desired input voltage, while also functioning as an on/off pin. The OVI pin enables implementation of an input overvoltage protection scheme, ensuring that the converter shuts down when the DC input voltage exceeds a set maximum value. The SS pin allows programmable soft-start time for the power converter, and helps limit inrush current during startup. The MAX17595/MAX17596/MAX17597 family also allows the designer to choose between voltage soft-start and current soft-start modes, useful in optoisolated designs. A programmable slope compensation scheme is provided to enhance the stability of the peak-current-mode control scheme. Hiccup-mode overcurrent protection and thermal shutdown are provided to minimize dissipation in overcurrent and overtemperature fault conditions. The IC is available in a space-saving 16-pin, 3mm x 3mm TQFN package with 0.5mm lead spacing. Benefits and Features S Peak Current Mode Offline (Universal Input AC) and Telecom (36V–72V) Flyback Controller (MAX17595) S Peak-Current-Mode DC-DC Flyback Controller (4.5V–36V Input Range) (MAX17596) S Peak-Current-Mode Nonsynchronous Boost PWM Controller (4.5V–36V Input Range) (MAX17597) S Current Mode Control Provides Excellent Transient Response S Low 20µA Startup Supply Current S 100kHz to 1MHz Programmable Switching Frequency S Programmable Frequency Dithering for Low-EMI Spread-Spectrum Operation S Switching Frequency Synchronization S Adjustable Current Limit with External CurrentSense Resistor S Fast Cycle-By-Cycle Peak Current Limiting S Hiccup-Mode Short-Circuit Protection S Overtemperature Protection S Programmable Soft-Start and Slope Compensation S Programmable Voltage or Current Soft-Start Schemes S Input Overvoltage Protection S Space-Saving, 3mm x 3mm TQFN Package Applications Universal Input Offline AC-DC Power Supplies Wide-Range DC-Input Flyback/Boost Battery Chargers Battery-Powered Applications Industrial, Telecom, and Automotive Applications Ordering Information/Selector Guide appears at end of data sheet. For related parts and recommended products to use with this part, refer to www.maxim-ic.com/MAX17595.related. ����������������������������������������������������������������� Maxim Integrated Products 1 For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com. MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators ABSOLUTE MAXIMUM RATINGS VIN to SGND...........................................................-0.3V to +40V VDRV to SGND...................................-0.3V to +16V (MAX17595) VDRV to SGND...........-0.3V to +6V (MAX17596 and MAX17597) NDRV to SGND..................................... -0.3V to +(VDRV + 0.3)V EN/UVLO to SGND................................... -0.3V to +(VIN + 0.3)V OVI, RT, DITHER, COMP, SS, FB, SLOPE to SGND..................................................... -0.3V to +6V CS to SGND.............................................................-0.8V to +6V PGND to SGND.....................................................-0.3V to +0.3V Maximum Input/Output Current (Continuous) VIN, NDRV.........................................................................100mA NDRV (pulsed, for less than 100ns)..................................... Q1A Continuous Power Dissipation TQFN (single-layer board) (derate 20.8mW/NC above +70NC).............................1666mW Operating Temperature Range......................... -40NC to +125NC Storage Temperature Range............................. -65NC to +150NC Junction Temperature......................................................+150NC Lead Temperature (soldering, 10s).................................+300NC Soldering Temperature (reflow).......................................+260NC Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. PACKAGE THERMAL CHARACTERISTICS (Note 1) Junction-to-Ambient Thermal Resistance (qJA)...............48°C/W Junction-to-Case Thermal Resistance (qJC)......................7°C/W Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial. ELECTRICAL CHARACTERISTICS (VIN = 12V (for the MAX17595, bring VIN up to 21V for startup), VCS = VSLOPE = VDITHER = VFB = VOVI = VSGND = 0V, VEN/UVLO = +2V; NDRV, SS, COMP are unconnected, RRT = 25kI, CVIN = 1FF, CVDRV = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = TJ = +25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS INPUT SUPPLY (VIN) VIN Voltage Range VIN MAX17595 MAX17596/MAX17597 8 29 4.5 36 V MAX17595 18.5 20 21.5 MAX17596/MAX17597 3.5 4 4.4 MAX17595 6.5 7 7.7 MAX17596/MAX17597 3.3 3.9 4.25 VIN < UVLO 20 32 FA 32 FA VIN Bootstrap UVLO Wakeup VIN-UVR VIN rising # VIN Bootstrap UVLO Shutdown Level VIN-UVF VIN falling $ VIN Supply Start-Up Current (Under UVLO) IVINSTARTUP VIN Supply Shutdown Current IIN-SH VEN = 0V 20 VIN Supply Current IIN-SW Switching, fSW = 400kHz 2 V V mA VIN CLAMP (INC) (MAX17595 ONLY) VIN Clamp Voltage VINC MAX17595, IVIN = 2mA sinking, VEN = 0V (Note 3) VENR VENF 30 33 36 VEN rising # 1.16 1.21 1.26 VEN falling $ 1.1 1.15 1.2 V ENABLE (EN) EN Undervoltage Threshold EN Input Leakage Current IEN VEN = 1.5V, TA = +25NC -100 +100 V nA ����������������������������������������������������������������� Maxim Integrated Products 2 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators ELECTRICAL CHARACTERISTICS (continued) (VIN = 12V (for the MAX17595, bring VIN up to 21V for startup), VCS = VSLOPE = VDITHER = VFB = VOVI = VSGND = 0V, VEN/UVLO = +2V; NDRV, SS, COMP are unconnected, RRT = 25kI, CVIN = 1FF, CVDRV = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = TJ = +25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX 8V < VIN < 15V and 0mA < IVDRV < 50mA (MAX17595) 7.1 7.4 7.7 6V < VIN < 12V and 0mA < IVDRV < 50mA (MAX17596/MAX17597) 4.7 4.9 70 100 UNITS INTERNAL LDO (VDRV) VDRV Output Voltage Range VDRV Current Limit VDRV Dropout VDRV IVDRV-MAX V 5.1 mA VIN = 4.5V, IVDRV = 20mA (MAX17596/ MAX17597) 4.2 VOVIR VOVI rising # 1.16 1.21 1.26 VOVIF VOVI falling $ 1.1 1.15 1.2 VVDRV-DO V OVERVOLTAGE PROTECTION (OVI) OVI Overvoltage Threshold OVI Masking Delay OVI Input Leakage Current tOVI-MD IOVI 2 VOVI = 1V, TA = +25NC V Fs -100 +100 nA 100 1000 kHz -8 +8 % OSCILLATOR (RT) NDRV Switching Frequency Range fSW NDRV Switching Frequency Accuracy Maximum Duty Cycle DMAX (MAX17595/MAX17596) 46 48 50 (MAX17597) 90 92.5 95 % SYNCHRONIZATION (DITHER) Synchronization Logic-High Input VHI-SYNC 3 Synchronization Pulse Width Synchronization Frequency Range V 50 fSYNCIN (MAX17595/MAX17596) 1.1 x fSW ns 1.8 x fSW Hz DITHERING RAMP GENERATOR (DITHER) Charging Current VDITHER = 0V 45 50 55 FA Discharging Current VDITHER = 2.2V 43 50 57 FA Ramp-High Trip Point 2 V Ramp-Low Trip Point 0.4 V ����������������������������������������������������������������� Maxim Integrated Products 3 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators ELECTRICAL CHARACTERISTICS (continued) (VIN = 12V (for the MAX17595, bring VIN up to 21V for startup), VCS = VSLOPE = VDITHER = VFB = VOVI = VSGND = 0V, VEN/UVLO = +2V; NDRV, SS, COMP are unconnected, RRT = 25kI, CVIN = 1FF, CVDRV = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = TJ = +25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS 9 4.4 10 11 FA 5 5.6 FA 1.21 1.23 V SOFT-START/SOFT-STOP (SS) Soft-Start Charging Current Soft-Stop Discharging Current ISSCH ISSDISCH SS Bias Voltage For soft-stop enabled parts VSS SS Discharge Threshold 1.19 VSSDISCH Soft-stop completion 0.15 V Pulldown Impedance RNDRV-N INDRV (sinking) = 100mA 1.37 3 I Pullup Impedance RNDRV-P INDRV (sourcing) = 5mA 4.26 8.5 I Peak Sink Current CNDRV = 10nF 1.5 A Peak Source Current CNDRV = 10nF 0.9 A NDRV DRIVER (NDRV) Fall Time tNDRV-F CNDRV = 1nF 10 ns Rise Time tNDRV-R CNDRV = 1nF 20 ns CURRENT-LIMIT COMPARATOR (CS) Cycle-by-Cycle Peak -CurrentLimit Threshold VCS-PEAK 290 305 320 mV Cycle-by-Cycle Runaway Current-Limit Threshold VCS-RUN 340 360 380 mV Cycle-by-Cycle ReverseCurrent Limit Threshold VCS-REV -122 -102 -82 mV Current-Sense Leading-Edge Blanking Time tCS-BLANK Propagation Delay from Comparator Input to NDRV tPDCS From NDRV rising # edge 70 ns From CS rising (10mV overdrive) to NDRV falling (excluding leading edge blanking) 40 ns Number of Consecutive PeakCurrent-Limit Events to Hiccup NHICCUP-P 8 event Number of Runaway-CurrentLimit Events to Hiccup NHICCUP-R 1 event Overcurrent Hiccup Timeout Minimum On-Time 32768 tON-MIN cycle 90 130 170 ns 9 10 11 FA SLOPE COMPENSATION (SLOPE) Slope Bias Current ISLOPE Slope Resistor Range 25 200 kI Slope Voltage Range to Enable Current Soft-Start and Minimum Slope Compensation 0 200 mV ����������������������������������������������������������������� Maxim Integrated Products 4 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators ELECTRICAL CHARACTERISTICS (continued) (VIN = 12V (for the MAX17595, bring VIN up to 21V for startup), VCS = VSLOPE = VDITHER = VFB = VOVI = VSGND = 0V, VEN/UVLO = +2V; NDRV, SS, COMP are unconnected, RRT = 25kI, CVIN = 1FF, CVDRV = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted. Typical values are at TA = TJ = +25NC.) (Note 2) PARAMETER SYMBOL CONDITIONS Slope Voltage Range to Enable Voltage Soft-Start and Minimum Slope Compensation MIN TYP MAX 4 Slope Voltage Range to Enable Voltage Soft-Start and Programmable Slope Compensation V 0.2 Slope Compensation Ramp RSLOPE = 100kW Default Slope Compensation Ramp VSLOPE < 0.2V or 4V < VSLOPE 140 UNITS 165 4 V 190 mV/Fs 50 mV/Fs PWM COMPARATOR Comparator Offset Voltage VPWM-OS Current-Sense Gain ACS-PWM VCOMP - VCS DCOMP/DCS (TA = +25NC) CS Peak Slope Ramp Current ICSSLOPE Comparator Propagation Delay 1.65 1.81 2 V 1.75 1.97 2.15 V/V Ramp current peak (TA = +25NC) 13 20 FA tPWM Change in VCS = 10mV (including internal lead-edge blanking) 110 VREF VFB, when ICOMP = 0 and VCOMP = 1.8V 1.19 VFB = 1.5V, TA = +25NC -100 ns ERROR AMPLIFIER FB Reference Voltage FB Input Bias Current Voltage Gain IFB AEAMP Transconductance Gm Transconductance Bandwidth BW 1.21 1.23 +100 80 1.5 Open-loop (gain = 1), -3dB frequency 1.8 V nA dB 2.1 10 mS MHz Source Current VCOMP = 1.8V, VFB = 1V 80 120 210 FA Sink Current VCOMP = 1.8V, VFB = 1.75V 80 120 210 FA THERMAL SHUTDOWN Thermal Shutdown Threshold Thermal Shutdown Hysteresis Temperature rising +160 NC 20 NC Note 2: All devices 100% production tested at TA = +25°C. Limits over temperature are guaranteed by design. Note 3: The MAX17595 is intended for use in universal input power supplies. The internal clamp circuit at VIN is used to prevent the bootstrap capacitor from changing to a voltage beyond the absolute maximum rating of the device when EN is low (shutdown mode). Externally limit the maximum current to VIN (hence to clamp) to 2mA (max) when EN is low. ����������������������������������������������������������������� Maxim Integrated Products 5 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Typical Operating Characteristics (VIN = 15V, VEN/UVLO = +2V, COMP = open, CVIN = 1FF, CVCC = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted.) BOOTSTRAP UVLO WAKE-UP LEVEL vs. TEMPERATURE (MAX17595) MAX17595/6/7 toc01 MAX17595/6/7 toc02 4.15 20.03 VIN WAKE-UP LEVEL (V) 4.10 20.02 20.01 20.00 4.05 4.00 3.95 19.99 19.98 VIN FALLING THRESHOLD vs. TEMPERATURE (MAX17595) VIN BOOTSTRAP UVLO SHUTDOWN LEVEL (V) 20.04 3.90 -40 -20 0 20 40 60 80 100 120 0 20 40 60 80 7.020 7.015 7.010 7.005 7.000 100 120 -40 -20 0 3.90 3.85 3.80 60 80 100 120 MAX17595/6/7 toc05 1.209 EN/UVLO RISING THRESHOLD (V) 3.95 40 EN/ UVLO RISING THRESHOLD vs. TEMPERATURE MAX17595/6/7 toc04 4.00 20 TEMPERATURE (°C) TEMPERATURE (°C) VIN FALLING THRESHOLD vs. TEMPERATURE (MAX17596/MAX17597) VIN UVLO SHUTDOWN THRESHOLD (V) MAX17595/6/7 toc03 7.025 6.995 -40 -20 TEMPERATURE (°C) 1.208 1.207 1.206 1.205 1.204 1.203 1.202 3.75 -40 -20 0 20 40 60 80 -40 -20 100 120 0 EN/UVLO FALLING THRESHOLD vs. TEMPERATURE 40 60 80 100 120 OVI RISING THRESHOLD vs. TEMPERATURE MAX17595/6/7 toc06 1.148 1.147 1.146 1.145 MAX17595/6/7 toc07 1.211 OVI RISING THRESHOLD (V) 1.149 20 TEMPERATURE (°C) TEMPERATURE (°C) EN / UVLO FALLING THRESHOLD (V) BOOTSTRAP UVLO WAKE-UP LEVEL (V) VIN WAKE-UP LEVEL vs. TEMPERATURE (MAX17596/MAX17597) 1.210 1.209 1.208 1.207 -40 -20 0 20 40 60 TEMPERATURE (°C) 80 100 120 -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) ����������������������������������������������������������������� Maxim Integrated Products 6 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Typical Operating Characteristics (continued) (VIN = 15V, VEN/UVLO = +2V, COMP = open, CVIN = 1FF, CVCC = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted.) OVI FALLING THRESHOLD vs. TEMPERATURE MAX17595/6/7 toc08 1.1495 1.1490 1.1485 1.1480 1.1475 2.4 24.5 23.5 22.5 21.5 0 20 40 60 80 0 20 40 60 80 2.0 1.9 1.8 100 120 -40 -20 0 TEMPERATURE (°C) NDRV SWITCHING FREQUENCY (kHz) 700 600 500 400 300 200 80 100 120 RRT = 10kI 850 750 650 550 450 350 250 RRT = 100kI 150 100 60 MAX17595/6/7 toc12 950 800 40 NDRV SWITCHING FREQUENCY vs. TEMPERATURE MAX17595/6/7 toc11 900 20 TEMPERATURE (°C) NDRV SWITCHING FREQUENCY vs. RESISTOR NDRV SWITCHING FREQUENCY (kHz) 2.1 1.5 -40 -20 TEMPERATURE (°C) 50 0 5 15 25 35 45 55 65 75 85 95 FREQUENCY SELECTION RESISTOR (kI) FREQUENCY DITHERING vs. RDITHER -40 -20 0 20 40 60 80 100 120 TEMPERATURE (°C) SWITCHING WAVEFORMS (MAX17595) MAX17595/6/7 toc13 14 FREQUENCY DITHERING (%) 2.2 1.6 100 120 1000 2.3 1.7 20.5 19.5 -40 -20 MAX17595/6/7 toc10 2.5 SWITCHING CURRENT (mA) 1.1500 SWITCHING CURRENT vs. TEMPERATURE MAX17595/6/7 toc09 25.5 VIN SUPPLY CURRENT UNDER UVLO (µA) 1.1505 OVI FALLING THRESHOLD (V) VIN SUPPLY CURRENT UNDER UVLO vs. TEMPERATURE MAX17595/6/7 toc14 12 10 VDRAIN 100V/div 8 6 IPRI 1A /div 4 2 200 300 400 500 600 700 800 900 1000 4µs/div RDITHER (kI) ����������������������������������������������������������������� Maxim Integrated Products 7 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Typical Operating Characteristics (continued) (VIN = 15V, VEN/UVLO = +2V, COMP = open, CVIN = 1FF, CVCC = 1FF, TA = TJ = -40NC to +125NC, unless otherwise noted.) ENABLE STARTUP ENABLE SHUTDOWN MAX17595/6/7 toc15 HICCUP OPERATION MAX17595/6/7 toc16 EN/UVLO 5V/div MAX17595/6/7 toc17 EN/UVLO 5V/div VOUT 10V/div VOUT 10V/div VOUT 10V/div COMP 1V/div VDRAIN 100V/div COMP 1V/div 2ms /div IPRI 2A/div 1ms/div 400µs/div LOAD TRANSIENT RESPONSE (15V OUTPUT) SWITCHING CURRENT vs. SWITCHING FREQUENCY MAX17595/6/7 toc18 MAX17595/6/7 toc19 VOUT (AC) 0.5V/div 2.3 2.1 ILOAD 0.5A/div 1.9 1.7 1.5 100 200 300 400 500 600 700 800 900 1000 20ms/div SWITCHING FREQUENCY (Hz) EFFICIENCY GRAPH (15V OUTPUT) BODE PLOT (15V OUTPUT) MAX17595/6/7 toc20 MAX17595/6/7 toc21 100 90 80 PHASE 36°/div EFFICIENCY (%) SWITCHING CURRENT (mA) 2.5 VDC = 120V 70 60 50 40 30 BANDWIDTH = 8.8kHz PHASE MARGIN = 64° 6 81 2 4 6 81 GAIN 10dB/div 2 4 6 8 20 10 0 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 LOAD CURRENT (A) ����������������������������������������������������������������� Maxim Integrated Products 8 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators NDRV PGND CS TOP VIEW N.C. Pin Configuration 12 11 10 9 VDRV 13 MAX17595 MAX17596 MAX17597 VIN 14 EN / UVLO 15 EP 1 2 3 4 SLOPE RT DITHER / SYNC + N.C. OVI 16 TQFN 8 SGND 7 SS 6 FB 5 COMP Pin Description PIN NAME 1, 12 N.C. FUNCTION No Connection 2 SLOPE Slope Compensation Input. A resistor, RSLOPE, connected from SLOPE to SGND programs the amount of slope compensation with reference-voltage soft-start mode. Connecting this pin to SGND enables duty-cycle soft-start with minimum slope compensation of 50mV/Fs. Setting VSLOPE > 4V enables reference voltage soft-start with minimum slope compensation of 50mV/Fs. 3 RT Switching Frequency Programming Resistor Connection. Connect resistor RRT from RT to SGND to set the PWM switching frequency. 4 DITHER/SYNC Frequency Dithering Programming or Synchronization Connection. For spread-spectrum frequency operation, connect a capacitor from DITHER to SGND, and a resistor from DITHER to RT. To synchronize the internal oscillator to the externally applied frequency, connect DITHER/SYNC to the synchronization pulse. 5 COMP 6 FB Transconductance Amplifier Inverting Input 7 SS Soft-Start Capacitor Pin for Flyback Regulator. Connect a capacitor CSS from SS to SGND to set the soft-start time interval. 8 SGND 9 CS 10 PGND Power Ground. Connect PGND to the power ground plane. 11 NDRV External Switching nMOS Gate-Driver Output Transconductance Amplifier Output. Connect the frequency compensation network between COMP and SGND. Signal Ground. Connect SGND to the signal ground plane. Current-Sense Input. Peak-current-limit trip voltage is 300mV. ����������������������������������������������������������������� Maxim Integrated Products 9 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Pin Description (continued) PIN NAME FUNCTION 13 VDRV Linear Regulator Output and Driver Input. Connect input bypass capacitor from VDRV to SGND as close as possible to the IC. 14 VIN Internal VDRV Regulator Input. Connect VIN to the input voltage source. Bypass VIN to PGND with a 1FF minimum ceramic capacitor. 15 EN/UVLO 16 OVI Overvoltage Comparator Input. Connect a resistive divider between the input supply, OVI, and SGND to set the input overvoltage threshold. — EP Exposed Pad Enable/Undervoltage Lockout. To externally program the UVLO threshold of the input supply, connect a resistive divider between input supply, EN, and SGND. Detailed Description The MAX17595 offers a bootstrap UVLO wakeup level of 20V with a wide hysteresis of 15V minimum, and is optimized for implementing isolated and non-isolated universal (85V to 265V AC) offline single-switch flyback converter or telecom (36V to 72V) power supplies. The MAX17596/MAX17597 offer a UVLO wakeup level of 4.4V and are well-suited for low-voltage DC-DC flyback/ boost power supplies. An internal 1% reference (1.21V) can be used to regulate the output down to 1.21V in nonisolated flyback and boost applications. Additional semi-regulated outputs, if needed, can be generated by using additional secondary windings on the flyback converter transformer. The MAX17595/MAX17596/MAX17597 family utilizes peak-current-mode control and external compensation for optimizing closed-loop performance. The devices include cycle-by-cycle peak current limit, and eight consecutive occurrences of current-limit-event trigger hiccup mode, which protects external components by halting switching for a period of 32,768 cycles. The devices also include voltage soft-start for nonisolated designs, and current soft-start for isolated designs to allow monotonic and smooth rise of the outpu voltage during startup. The voltage and current soft-start modes can be selected using the SLOPE pin. See Figure 1 for more information. Input Voltage Range (VIN) The MAX17595 has different rising and falling undervoltage lockout (UVLO) thresholds on the VIN pin than the thresholds of the MAX17596/MAX17597. The thresholds for the MAX17595 are optimized for implementing power supply startup schemes, typically used for offline AC-DC power supplies. The MAX17595 is well-suited for operation from rectified DC bus in AC-DC power-supply applications, which are typical of front-end industrial power-supply applications. As such, the MAX17595 has no limitation on maximum input voltage, as long as the external components are rated suitably and the maximum operating voltages of the MAX17595 are respected. The MAX17595 can be successfully used in universal input (85V to 265V AC) rectified bus applications, in rectified 3-phase DC bus applications, and in telecom (36V to 72V DC) applications. The MAX17596/MAX17597 are intended to implement flyback (isolated and nonisolated) and boost converters. The VIN pin of the MAX17596/MAX17597 has a maximum operating voltage of 36V. The MAX17596/ MAX17597 implement rising and falling thresholds on the VIN pin that assume power-supply startup schemes typical of low-voltage DC-DC applications, down to an input voltage of 4.5V DC. Therefore, flyback /boost converters with a 4.5V to 36V supply voltage range can be implemented with the MAX17596/MAX17597. ���������������������������������������������������������������� Maxim Integrated Products 10 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators THERMAL SENSOR VDRV 7.5V (MAX17595) OR 5V (MAX17596/ MAX17597) LDO ±50µA MAX17595 MAX17596 MAX17597 5V LDO DITHER CONTROL AND DRIVER LOGIC AV POK 2V/0.4V HICCUP VDRV VIN NDRV DRIVER UVLO 8 PEAK EVENTS OR 1 RUNAWAY CHIPEN OSC EN / UVLO SSDONE PGND OSC PEAKLIM COMP PGND 305mV DITHER (SYNC) 1.21V RUNAWAY COMP OVI 360mV 1.21V BLANKING CS 70ns PWM COMP RT FIXED OR VAR 10µA 10µA SS CHIPPEN SLOPE SLOPE DECODE SS SSDONE 5µA SS MODE 1.23V OSC R COMP 1X SGND CHIPPEN/ HICCUP (FACTORY OPTION) VOLTAGE SOFT-START 1.21V R SS MODE SS FB SS SS MODE CURRENT SOFT-START Figure 1. MAX17595/MAX17596/MAX17597 Block Diagram ���������������������������������������������������������������� Maxim Integrated Products 11 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Internal Linear Regulator (VDRV) The internal functions and driver circuits are designed to operate from 7.5V (MAX17595) or 5V (MAX17596/ MAX17597) power supply voltages. The MAX17595/ MAX17596/MAX17597 family has an internal linear regulator that is powered from the VIN pin. The output of the linear regulator is connected to the VDRV pin, and should be decoupled with a 1FF capacitor to ground for stable operation. The VDRV regulator output supplies all the operating current of the MAX17595/MAX17596/MAX17597. The maximum operating voltage on the VIN pin is 29V for the MAX17595, and 36V for the MAX17596/MAX17597. n-Channel MOSFET Gate Driver (NDRV) The MAX17595/MAX17596/MAX17597 family offers a built-in gate driver for driving an external n-channel MOSFET. The NDRV pin can source/sink currents in excess of 650mA/1000mA. Maximum Duty Cycle The MAX17595/MAX17596 operate at a maximum duty cycle of 49%. The MAX17597 offers a maximum duty cycle of 94% to implement flyback and boost converters involving large input-to-output voltage ratios in DC-DC applications. Slope compensation is necessary for stable operation of peak-current-mode controlled converters such as the MAX17595/MAX17596/MAX17597 at duty cycles greater than 50%, in addition to the loop compensation required for small signal stability. The MAX17595/ MAX17596/MAX17597 implement a SLOPE pin for this purpose. See the Slope Compensation section for more details. Soft-Start (SS) The MAX17595/MAX17596/MAX17597 devices implement soft-start operation for the flyback/boost regulator. A capacitor connected to the SS pin programs the softstart period. The soft-start feature reduces input inrush current during startup. The devices allow the end user to select between voltage soft-start, usually preferred in nonisolated applications, and current soft-start, which is useful in isolated applications to get a monotonic and smooth rise in output voltage. See the Input Voltage Range (VIN) section. Soft-Stop A soft-stop feature can be requested from the factory. This feature ramps down the duty cycle of operation of the converter to zero in a controlled fashion, and enables controlled ramp down of output voltage. The soft-stop duration is twice that of the programmed soft-start period. This is particularly useful in implementing controlled shutdown of output voltage in isolated power converters. Switching Frequency Selection (RT) The ICs’ switching frequency is programmable between 100kHz and 1MHz with a resistor RRT connected between RT and SGND. Use the following formula to determine the appropriate value of RRT needed to generate the desired output-switching frequency (fSW): R RT = 10 10 fSW where fSW is the desired switching frequency. Frequency Dithering for Spread-Spectrum Applications (Low EMI) The switching frequency of the converter can be dithered in a range of Q10% by connecting a capacitor from DITHER/SYNC to SGND, and a resistor from DITHER to RT, as shown in the Typical Operating Circuits. Spread-spectrum modulation technique spreads the energy of switching frequency and its harmonics over a wider band while reducing their peaks, helping to meet stringent EMI goals. Applications Information Startup Voltage and Input Overvoltage Protection Setting (EN/UVLO, OVI) The devices’ EN/UVLO pin serves as an enable/disable input, as well as an accurate programmable input UVLO pin. The devices do not commence startup operation unless the EN/UVLO pin voltage exceeds 1.21V (typ). The devices turn off if the EN/UVLO pin voltage falls below 1.15V (typ). A resistor-divider from the input DC bus to ground can be used to divide down and apply a fraction of the input DC voltage (VDC) to the EN/UVLO pin. The values of the resistor-divider can be selected so that the EN/UVLO pin voltage exceeds the 1.23V (typ) turn-on threshold at the desired input DC bus voltage. The same resistor-divider can be modified with an additional resistor (ROVI) to implement input overvoltage protection in addition to the EN/UVLO functionality as shown in Figure 2. When voltage at the OVI pin exceeds 1.21V (typ), the devices stop switching and resume switching operations only if voltage at the OVI pin falls below 1.15V (typ). For given values of startup DC input ���������������������������������������������������������������� Maxim Integrated Products 12 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators voltage (VSTART) and input overvoltage-protection voltage (VOVI), the resistor values for the divider can be calculated as fol lows, assuming a 24.9kI resistor for ROVI: VOVI R EN = R OVI × − 1 k I VSTART where ROVI is in kI, while VSTART and VOVI are in volts. VSTART R SUM = R OVI + R EN × − 1 k I 1.21 where REN RDC1 RSUM RDC2 RDC3 EN/UVLO REN , ROVI is in kI, while VSTART is in volts. In universal AC input applications, RSUM might need to be implemented as equal resistors in series (RDC1, RDC2, and RDC) so that voltage across each resistor is limited to its maximum operation voltage. R = DC1 R= DC2 R= DC3 R SUM 3 kI For low-voltage DC-DC applications based on the MAX17596/MAX17597, a single resistor can be used in the place of RSUM, as the voltage across it is approximately 40V. Startup Operation The MAX17595 is optimized for implementing an offline single-switch flyback converter and has a 20V VIN UVLO wake-up level with hysteresis of 15V (min). In offline applications, a simple cost-effective RC startup circuit is used. When the input DC voltage is applied, the startup resistor (RSTART) charges the startup capacitor (CSTART), causing the voltage at the VIN pin to increase towards the wake-up VIN UVLO threshold (20V typ). During this time, the MAX17595 draws a low startup current of 20FA (typ) through RSTART. When the voltage at VIN reaches the wake-up VIN UVLO threshold, the MAX17595 commences switching and control operations. In this condition, the MAX17595 draws 2mA (typ) current from CSTART, when operated at 1MHz switching frequency, for its internal operation. In addition, the average value of gate drive current is also drawn from CSTART, which is a function of the gate charge of the external MOSFET used. Since this total current cannot be supported by the current through RSTART, the voltage on CSTART starts to drop. When suitably configured, as shown in Figure 9, the external MOSFET is switched by the NDRV pin and the flyback converter generates an output voltage OVI MAX17595 MAX17596 MAX17597 ROVI Figure 2. Programming EN /UVLO and OVI (VOUT), and a bias voltage (VBIAS) that is bootstrapped to the VIN pin through the diode (D2). If VBIAS exceeds the sum of 7V, and the drop across D2 before the voltage on CSTART falls below 7V, then the VIN voltage is sustained by VBIAS, allowing the MAX17595 to continue operating with energy from VBIAS. The large hysteresis (13V typ) of the MAX17595 allows for a small startup capacitor (CSTART). The low startup current (20FA typ) allows the use of a large startup resistor (RSTART), thus reducing power dissipation at higher DC bus voltages. Figure 3 shows the typical RC startup scheme for the MAX17595, when the output voltage VOUT is used as the bias voltage to sustain switching operation. RSTART might need to be implemented as equal, multiple resistors in series (RIN1, RIN2, and RIN3) to share the applied high DC voltage in offline applications so that the voltage across each resistor is limited to its maximum continuous operating voltage rating. RSTART and CSTART can be calculated as: Q GATE × fSW t SS C START = FF IIN + × 10 6 10 where IIN is the supply current drawn at the VIN pin in mA, QGATE is the gate charge of the external MOSFET used in nC, fSW is the switching frequency of the converter in Hz, and tSS is the soft-start time programmed for the flyback converter in ms. See the Programming Soft-Start of Flyback/Boost Converter (SS) section. ���������������������������������������������������������������� Maxim Integrated Products 13 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators R START = (VSTART − 10) × 50 The startup capacitor (CSTART) can be calculated as: kI 1 + C START Q GATE × fSW t SS C START = FF IIN + × 10 6 10 where CSTART is the startup capacitor in FF. For designs that cannot accept power dissipation in the startup resistors at high DC input voltages in offline appli cations, the startup circuit can be set up with a current source instead of a startup resistor as shown in Figure 4. where IIN is the supply current drawn at the VIN pin in mA, QGATE is the gate charge of the external MOSFET used in nC, fSW is the switching frequency of the converter in kHz, and tSS is the soft-start time programmed for the flyback converter in ms. VDC VOUT RIN1 RSTART RIN2 VDC VOUT CF MAX17595 RIN3 NDRV VIN CSTART LDO DRV CS VDRV CVDRV Figure 3. MAX17595 RC-Based Startup Circuit VDC RIN1 RSUM RIN2 VDC VOUT D1 RIN3 COUT VOUT MAX17595 RISRC NDRV VIN CSTART LDO DRV CS VDRV CVDRV RS Figure 4. MAX17595 Current-Source-Based Startup Circuit ���������������������������������������������������������������� Maxim Integrated Products 14 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Resistors RSUM and RISRC can be calculated as: = R SUM = RISRC VSTART 10 VBEQ1 70 incurred to supply the operating current of the MAX17596/ MAX17597 can be tolerated, the VIN pin is directly connected to the DC input, as shown in Figure 5. In the case of higher DC input voltages (e.g., 16V to 32V DC), a startup circuit, such as that shown in Figure 6, can be used to minimize power dissipation in the startup circuit. In this startup scheme, the transistor (Q1) supplies the switching current until a bias winding NB comes up. The resistor (RZ) can be calculated as: MW MW The VIN UVLO wakeup threshold of the MAX17596/ MAX17597 is set to 4.1V (typ) with a 200mV hysteresis, optimized for low-voltage DC-DC applications down to 4.5V. For applications where the input DC voltage is low enough (e.g., 4.5V to 5.5V DC) that the power loss RZ = 9 × (VINMIN − 6.3) kW VDC VOUT D1 VIN VIN VDRV LDO COUT CDRV Np Ns NDRV CS MAX17596 MAX17597 RS Figure 5. MAX17596/MAX17597 Typical Startup Circuit with VIN Connected Directly to DC Input VDC D1 RZ Q1 VIN LDO VDRV ZD1 6.3V NB COUT CDRV Np Ns VIN CIN NDRV CS MAX17596 MAX17597 RS Figure 6. MAX17596/MAX17597 Typical Startup Circuit with Bias Winding to Turn Off Q1 and Reduce Power Dissipation ���������������������������������������������������������������� Maxim Integrated Products 15 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Programming Soft-Start of Flyback /Boost Converter (SS) The soft-start period in the voltage soft-start scheme of the devices can be programmed by selecting the value of the capacitor connected from the SS pin to SGND. The capacitor CSS can be calculated as: = C SS 8.2645 × t SS nF where tSS is expressed in ms. The soft-start period in the current soft-start scheme depends on the load at the output and the soft-start capacitor. Programming Output Voltage The devices incorporate an error amplifier with a 1% precision voltage reference that enables negative feedback control of the output voltage. The output voltage of the switching converter can be programmed by selecting the values for the resistor-divider connected from VOUT, and the flyback /boost output to ground, with the midpoint of the divider connected to the FB pin (Figure 7). With RB selected in the 20kI to 50kI range, RU can be calculated as: VOUT R U =× RB − 1 kI, where R B is in kI. 1.21 Peak-Current-Limit Setting (CS) The devices include a robust overcurrent protection scheme that protects the device under overload and short-circuit conditions. A current-sense resistor (RCS in the Typical Operating Circuits), connected between the source of the MOSFET and PGND, sets the peak current limit. The current-limit comparator has a voltage trip level (VCS-PEAK) of 300mV. Use the following equation to calculate the value of RCS: R CS = 300mV IMOSFET I where IMOSFET is the peak current flowing through the MOSFET. When the voltage produced by this current (through the current-sense resistor) exceeds the currentlimit comparator threshold, the MOSFET driver (NDRV) terminates the current on-cycle within 30ns (typ). The devices implement 65ns of leading-edge blanking to ignore leading-edge current spikes. These spikes are caused by reflected secondary currents, capacitance discharging current at the MOSFET’s drain, and gate charging current. Use a small RC network for additional filtering of the leading edge spike on the sense waveform VOUT RU FB RB MAX17595 MAX17596 MAX17597 Figure 7. Programming Output Voltage when needed. Set the corner frequency between 10MHz and 20MHz. After the leading-edge blanking time, the device monitors VCS. The duty cycle is terminated immediately when VCS exceeds 300mV. The devices offer a runaway current limit scheme that protects the devices under high-input-voltage shortcircuit conditions when there is insufficient output voltage available to restore inductor current built up during the on period of the flyback/boost converter. Either eight consecutive occurrences of the peak-current-limit event or one occurrence of the runaway current limit trigger a hiccup mode that protects the converter by immediately suspending switching for a period of time (tRSTART). This allows the overload current to decay due to power loss in the converter resistances, load, and the output diode of the flyback/boost converter before soft-start is attempted again. The runaway current limit is set at a VCS-PEAK of 360mV (typ). The peak-current-limittriggered hiccup operation is disabled until the end of the soft-start period, while the runaway current-limittriggered hiccup operation is always enabled. Programming Slope Compensation (SLOPE) The MAX17595/MAX17596 operate at a maximum duty cycle of 49%. In theory, they do not require slope compensation to prevent subharmonic instability that occurs naturally in continuous-conduction mode (CCM) peak-current-mode-controlled converters operating at duty cycles greater than 50%. In practice, the MAX17595/ MAX17596 require a minimum amount of slope compensation to provide stable operation. The devices allow the user to program this default value of slope compensation simply by leaving the SLOPE pin unconnected. It is recommended that discontinuous-mode designs also use this minimum amount of slope compensation to provide better noise immunity and jitter-free operation. ���������������������������������������������������������������� Maxim Integrated Products 16 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators The MAX17597 flyback/boost converter can be designed to operate in either discontinuous-conduction mode (DCM) or to enter into the continuous-conduction mode at a specific load condition for a given DC input voltage. In continuous-conduction mode, the flyback/ boost converter needs slope compensation to avoid subharmonic instability that occurs naturally over all specified load and line conditions in peak-current-mode controlled converters operating at duty cycles greater than 50%. A minimum amount of slope signal is added to the sensed current signal even for converters operating below 50% duty to provide stable, jitter-free operation. The SLOPE pin allows the user to program the necessary slope compensation by setting the value of the resistor (RSLOPE) connected from the SLOPE pin to ground. R SLOPE = SE − 8 1.55 kI where the slope (SE) is expressed in mV/Fs. Frequency Dithering for Spread-Spectrum Applications (Low EMI) The switching frequency of the converter can be dithered in a range of Q10% by connecting a capacitor from DITHER/SYNC to SGND, and a resistor from DITHER to RT as shown in the Typical Operating Circuits. This results in lower EMI. A current source at DITHER/SYNC charges the capacitor CDITHER to 2V at 50FA. Upon reaching this trip point, it discharges CDITHER to 0.4V at 50FA. The charging and discharging of the capacitor generates a triangular waveform on DITHER/SYNC with peak levels at 0.4V and 2V and a frequency that is equal to: fTRI = Error Amplifier, Loop Compensation, and Power Stage Design of Flyback/Boost Converter The flyback /boost converter requires proper loop compensation to be applied to the error-amplifier output to achieve stable operation. The goal of the compensator design is to achieve desired closed-loop bandwidth, and sufficient phase margin at the crossover frequency of the open-loop gain-transfer function of the converter. The error amplifier provided in the devices is a transconductance amplifier. The compensation network used to apply the necessary loop compensation is shown in Figure 8. The flyback/boost converter can be used to implement the following converters and operating modes: • Nonisolated flyback converter in discontinuous-conduction mode (DCM flyback) • Nonisolated flyback converter in continuous-conduction mode (CCM flyback) • Boost converter in discontinuous-conduction mode (DCM boost) • Boost converter in continuous-conduction mode (CCM boost) Calculations for loop-compensation values (RZ, CZ, and CP) for these converter types and design procedures for power-stage components are detailed in the following sections. COMP 50 FA C DITHER × 3.2V typically, fTRI should be set close to 1kHz. The resistor RDITHER connected from DITHER/SYNC to RT determines the amount of dither as follows: %DITHER = RZ CZ CP MAX17595 MAX17596 MAX17597 R RT R DITHER Figure 8. Error-Amplifier Compensation Network where %DITHER is the amount of dither expressed as a percentage of the switching frequency. Setting RDITHER to 10 x RRT generates Q10% dither. ���������������������������������������������������������������� Maxim Integrated Products 17 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators DCM Flyback Primary Inductance Selection In a DCM flyback converter, the energy stored in the primary inductance of the flyback transformer is delivered entirely to the output. The maximum primary inductance value for which the converter remains in DCM at all operating conditions can be calculated as: L PRIMAX ≤ (VINMIN × D MAX ) 2 × 0.4 (VOUT + VD ) × IOUT × fSW where DMAX is chosen as 0.35 for the MAX17595/ MAX17596 and 0.7 for the MAX17597; VD is the voltage drop of the output rectifier diode on the secondary winding, and fSW is the switching frequency of the power converter. Choose the primary inductance value to be less than LPRIMAX. Duty Cycle Calculation The accurate value of the duty cycle (DNEW) for the selected primary inductance (LPRI) can be calculated using the following equation: D NEW = 2.5 × L PRI × (VOUT + VD ) × IOUT × fSW VINMIN Turns Ratio Calculation (Ns/Np) Transformer turns ratio (K = Ns/Np) can be calculated as: K= (VOUT + VD ) × (1 − D MAX ) VINMIN × D MAX Peak/RMS Current Calculation The transformer manufacturer needs RMS current values in the primary and secondary to design the wire diameter for the different windings. Peak current calculations are useful in setting the current limit. Use the following equations to calculate the primary and secondary peak and RMS currents. Maximum primary peak current: IPRIPEAK = VINMIN × D NEW L PRI × fSW I SECPEAK = IPRIPEAK K Maximum primary peak current: I SECRMS = IPRIPEAK I SECPEAK x L PRI x fSW 3 x (VOUT + VD ) For the purpose of current-limit setting, ILIM can be calculated as follows: = ILIM IPRIPEAK × 1.2 Primary Snubber Selection Ideally, the external MOSFET experiences a drain-source voltage stress equal to the sum of the input voltage and reflected voltage across the primary winding during the off period of the MOSFET. In practice, parasitic inductors and capacitors in the circuit, such as leakage inductance of the flyback transformer, cause voltage overshoot and ringing, in addition to the ideally expected voltage stress. Snubber circuits are used to limit the voltage overshoots to safe levels within the voltage rating of the external MOSFET. The snubber capacitor can be calculated using the following equation: C SNUB = 2 × L LK × IPRIPEAK 2 × K 2 VOUT 2 where LLK is the leakage inductance that can be obtained from the transformer specifications (usually 1.5%–2% of the primary inductance). The power to be dissipated in the snubber resistor is calculated using the following formula: PSNUB = 0.833 × L LK × IPRIPEAK 2 × fSW The snubber resistor is calculated based on the equation below: R SNUB = 6.25 × VOUT 2 PSNUB × K 2 The voltage rating of the snubber diode is: Maximum primary RMS current: I= PRIRMS IPRIPEAK × Maximum secondary peak current: D NEW VOUT VDSNUB = VINMAX + 2.5 × K 3 ���������������������������������������������������������������� Maxim Integrated Products 18 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Output Capacitor Selection X7R ceramic output capacitors are preferred in industrial applications due to their stability over temperature. The output capacitor is usually sized to support a step load of 50% of the maximum output current in the application, so that the output-voltage deviation is contained to 3% of the output-voltage change. The output capacitance can be calculated as follows: C OUT = I STEP × t RESPONSE ∆VOUT 1 0.33 t RESPONSE ≅ + fSW fC where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable output voltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be one-tenth of the switching frequency, fSW. For the flyback converter, the output capacitor supplies the load current when the main switch is on; therefore, the output voltage ripple is a function of load current and duty cycle. Use the following equation to calculate the output capacitor ripple: D NEW × IPRIPEAK − K × IOUT ∆VCOUT = 2 × IPRIPEAK × fSW × C OUT 2 where IOUT is load current and DNEW is the duty cycle at minimum input voltage. Input Capacitor Selection The MAX17595 is optimized to implement offline AC-DC converters. In such applications, the input capacitor must be selected based on either the ripple due to the rectified line voltage, or based on holdup-time requirements. Holdup time can be defined as the time period over which the power supply should regulate its output voltage from the instant the AC power fails. The MAX17596/MAX17597 are useful in implementing low-voltage DC-DC applications where the switchingfrequency ripple must be used to calculate the input capacitor. In both cases, the capacitor must be sized to meet RMS current requirements for reliable operation. A) Capacitor selection based on switching ripple (MAX17596/MAX17597) For DC-DC applications, X7R ceramic capacitors are recommended due to their stability over the operating temperature range. The ESR and ESL of a ceramic capacitor are relatively low, so the ripple voltage is dominated by the capacitive component. For the flyback converter, the input capacitor supplies the current when the main switch is on. Use the following equation to calculate the input capacitor for a specified peak-to-peak input switching ripple (VIN_RIP): CIN = D NEW × IPRIPEAK 1 − (0.5 × D NEW ) 2 2 × fSW × VIN_RIP B) Capacitor selection based on rectified line voltage ripple (MAX17595) For the flyback converter, the input capacitor supplies the input current when the diode rectifier is off. The voltage discharge (VIN_RIP), due to the input average current, should be within the limits specified: CIN = 0.5 × IPRIPEAK × D NEW fRIPPLE × VIN_RIP where fRIPPLE, the input AC ripple frequency equal to the supply frequency for half-wave rectification, is two times the AC supply frequency for full-wave rectification. C) Capacitor selection based on holdup time requirements (MAX17595) For a given output power (PHOLDUP) that needs to be delivered during holdup time (tHOLDUP), DC bus voltage at which the AC supply fails (VINFAIL), and the minimum DC bus voltage at which the converter can regulate the output voltages (VINMIN), the input capacitor (CIN) is estimated as: CIN = 3 × PHOLDUP × t HOLDUP (VINFAIL 2 − VINMIN 2 ) the input capacitor RMS current can be calculated as: IINCRMS = 0.6 × VINMIN × (D MAX ) 2 fSW × L PRI ���������������������������������������������������������������� Maxim Integrated Products 19 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators External MOSFET Selection A MOSFET selection criterion includes maximum drain voltage, peak/RMS current in the primary, and the maximum allowable power dissipation of the package without exceeding the junction temperature limits. The voltage seen by the MOSFET drain is the sum of the input voltage, the reflected secondary voltage on the transformer primary, and the leakage inductance spike. The MOSFET’s absolute maximum VDS rating must be higher than the worst-case drain voltage: VOUT + VDIODE VDSMAX = VINMAX + × 2.5 K The drain current rating of the external MOSFET is selected to be greater than the worst-case peak-currentlimit setting. Secondary Diode Selection Secondary-diode selection criteria includes the maxi mum reverse voltage, average current in the secondaryreverse recovery time, junction capacitance, and the maximum allowable power dissipation of the package. The voltage stress on the diode is the sum of the output voltage and the reflected primary voltage. The maximum operating reverse-voltage rating must be higher than the worst-case reverse voltage: VSECDIODE= 1.25 × (K × VINMAX + VOUT ) The current rating of the secondary diode should be selected so that the power loss in the diode (given as the product of forward-voltage drop and the average diode current) should be low enough to ensure that the junction temperature is within limits. This necessitates that the diode current rating be in the order of 2 x IOUT to 3 x IOUT. Select fast-recovery diodes with a recovery time less than 50ns, or Schottky diodes with low junction capacitance. Error Amplifier Compensation Design The loop compensation values are calculated as: = R Z 450 × 0.1× f 2 SW 1 + × VOUT × IOUT fP 2 × L PRI × fSW where: fP = IOUT π × VOUT × C OUT CZ = CP = 1 π × R Z × fP 1 π × R Z × fSW fSW is the switching frequency of the devices. CCM Flyback Transformer Turns Ratio Calculation (K = Ns / Np) The transformer turns ratio can be calculated using the following formula: K= (VOUT + VD ) × (1 − D MAX ) VINMIN × D MAX where DMAX is the duty cycle assumed at minimum input (0.35 for the MAX17595/MAX17596 and 0.7 for the MAX17597). Primary Inductance Calculation Calculate the primary inductance based on the ripple: (VOUT + VD ) × (1 − D NOM) × K L PRI = 2 × IOUT × β × fSW where DNOM, the nominal duty cycle at nominal operating DC input voltage VINNOM, is given as: D NOM = (VOUT + VD ) × K VINNOM + (VOUT + VD ) × K The output current, down to which the flyback converter should operate in CCM, is determined by selection of the fraction A in the above primary inductance formula. For example, A should be selected as 0.15 so that the converter operates in CCM down to 15% of the maximum output load current. Since the ripple in the primary current waveform is a function of duty cycle and is maximum at maximum DC input voltage, the maximum (worst-case) load current down to which the converter operates in CCM occurs at maximum operating DC input voltage. VD is the forward drop of the selected output diode at maximum output current. ���������������������������������������������������������������� Maxim Integrated Products 20 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Peak and RMS Current Calculation RMS current values in the primary and secondary are needed by the transformer manufacturer to design the wire diameter for the different windings. Peak current calculations are useful in setting the current limit. Use the following equations to calculate the primary and secondary peak and RMS currents. Maximum primary peak current: IOUT × K VINMIN × D MAX IPRIPEAK = + 1 − D MAX 2 × L PRI × fSW Maximum primary RMS current: IPRIRMS = IPRIPEAK 2 + ∆IPRI 2 − (IPRIPEAK × ∆IPRI) 3 × D MAX where DIPRI is the ripple current in the primary current waveform and is given by: VINMIN × D MAX ∆IPRI = L PRI × fSW Maximum secondary peak current: IPRIPEAK I SECPEAK = K Maximum secondary RMS current: I SECRMS = of 50% of the maximum output current in the application so that the output-voltage deviation is contained to 3% of the output-voltage change. The output capacitance can be calculated as: C OUT = × 1 − D MAX where DISEC is the ripple current in the secondary current waveform and is given by: VINMIN × D MAX ∆I SEC = L PRI × fSW × K For the purpose of current-limit setting, the peak current can be calculated as follows: = ILIM IPRIPEAK × 1.2 Primary RCD Snubber Selection The design procedure for primary RCD snubber selection is identical to that outlined in the DCM Flyback section. Output Capacitor Selection X7R ceramic output capacitors are preferred in industrial applications due to their stability over temperature. The output capacitor is usually sized to support a step load ∆VOUT t RESPONSE ≅ ( 0.33 fC + 1 fSW ) where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable output voltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be less than one-fifth of the worst-case (lowest) RHP zero frequency fRHP. The right half-plane zero frequency is calculated as follows: fZRHP = (1 − D MAX ) 2 × VOUT 2 × π × D MAX × L PRI × IOUT × K 2 For the CCM flyback converter, the output capacitor supplies the load current when the main switch is on; therefore, the output voltage ripple is a function of load current and duty cycle. Use the following equation to estimate the output voltage ripple: IOUT × D MAX ∆VCOUT = fSW × C OUT I SECPEAK 2 + ∆I SEC 2 + (I SECPEAK × ∆I SEC ) 3 I STEP × t RESPONSE Input Capacitor Selection The design procedure for input capacitor selection is identical to that outlined in the DCM Flyback section. External MOSFET Selection The design procedure for external MOSFET selection is identical to that outlined in the DCM Flyback section. Secondary-Diode Selection The design procedure for secondary-diode selection is identical to that outlined in the DCM Flyback section. Error Amplifier Compensation Design In the CCM flyback converter, the primary inductance and the equivalent load resistance introduces a right half-plane zero at the following frequency: fZRHP = (1 − D MAX ) 2 × VOUT 2 × π × D MAX × L PRI × IOUT × K 2 ���������������������������������������������������������������� Maxim Integrated Products 21 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators The loop compensation values are calculated as: = RZ 225 × IOUT fRHP × 1+ (1 − D MAX ) 5 × fP 2 where fP, the pole due to output capacitor and load is given by: fP = (1 + D MAX ) × IOUT 2 × π × C OUT × VOUT where is IPK given by: 2 × (VOUT − VIN_MIN) × IOUT IPK = L INMIN × fSW LINMIN is the minimum value of the input inductor taking into account tolerance and saturation effects. Output Capacitor Selection The output capacitance can be calculated as follows: The above selection of RZ sets the loop-gain crossover frequency (fC, where the loop gain equals 1) equal to 1/5th the right half-plane zero frequency. fZRHP fC ≤ 5 With the control loop zero placed at the load pole frequency: 1 CZ = 2π × R Z × fP With the high-frequency pole placed at half the switching frequency: CP = 1 π × R Z × fSW C OUT = Inductance Selection The design procedure starts with calculating the boost converter’s input inductor, such that it operates in DCM at all operating line and load conditions. The critical inductance required to maintain DCM operation is calculated as: L IN ≤ where VINMIN is the minimum input voltage. Peak/RMS Currents Calculation For the purposes of setting the current limit, the peak current in the inductor can be calculated as: ILIM = IPK × 1.2 0.33 fC + 1 fSW ) where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable output voltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be one-tenth of the switching frequency fSW. For the boost converter, the output capacitor supplies the load current when the main switch is on; therefore, the output voltage ripple is a function of duty cycle and load current. Use the following equation to calculate the output capacitor ripple: IOUT × L IN × IPK ∆VCOUT = VINMIN × C OUT Input Capacitor Selection The input ceramic capacitor value required can be calculated based on the ripple allowed on the input DC bus. The input capacitor should be sized based on the RMS value of the AC current handled by it. The calculations are: 3.75 × IOUT CIN = V f (1 D ) × × − MAX INMIN SWMIN The capacitor RMS can be calculated as: (V − VIN_MIN ) × VIN_MIN 2 × 0.4 OUT IOUT × VOUT 2 × fSW ∆VOUT t RESPONSE ≅ ( DCM Boost In a DCM boost converter, the inductor current returns to zero in every switching cycle. Energy stored during the on-time of the main switch Q1 is delivered entirely to the load in each switching cycle. I STEP × t RESPONSE I CIN_RMS = IPK 2× 3 Error Amplifier Compensation Design The loop compensation values for the error amplifier can now be calculated as: = CZ G DC × G M × 10 = 2 × π × fSW (GDC × 10) nF where GDC, the DC gain of the power stage, is given as: ���������������������������������������������������������������� Maxim Integrated Products 22 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators G DC = 8 × (VOUT − VINMIN) × fSW × VOUT 2 × L IN RZ = (2VOUT − VINMIN) 2 × IOUT VOUT × C OUT × (VOUT − VINMIN) IOUT × C Z × (2VOUT − VINMIN) where VINMIN is the minimum operating input voltage, and IOUT is the maximum load current. CP = C OUT × ESR RZ Slope Compensation In theory, the DCM boost converter does not require slope compensation for stable operation. In practice, the converter needs a minimum amount of slope for good noise immunity at very light loads. The minimum slope is set for the MAX17596/MAX17597 by leaving the SLOPE pin unconnected. Output Diode Selection The voltage rating of the output diode for the boost converter ideally equals the output voltage of the boost converter. In practice, parasitic inductances and capacitances in the circuit interact to produce voltage overshoot during the turn-off transition of the diode that occurs when the main switch Q1 turns on. The diode rating should therefore be selected with the necessary margin to accommodate this extra voltage stress. A voltage rating of 1.3 x VOUT provides the necessary design margin in most cases. The RMS current in the MOSFET is useful in estimating the conduction loss, and is given as: IMOSFETRMS = IPK 3 × L INS × fSW 3 × VINMIN where IPK is the peak current calculated at the lowest operating input voltage, VINMIN. CCM Boost In a CCM boost converter, the inductor current does not return to zero during a switching cycle. Since the MAX17597 implements a nonsynchronous boost converter, the inductor current will enter DCM operation at load currents below a critical value equal to half of the peak-peak ripple in the inductor current. Inductor Selection The design procedure starts with calculating the boost converter’s input inductor at nominal input voltage for a ripple in the inductor current equal to 30% of the maximum input current. L IN = VIN × D × (1 − D) 0.3 × IOUT × fSW where D is the duty cycle calculated as: D= VOUT + VD − VIN VOUT + VD − (R DS × IOUT ) VD is the voltage drop across the output diode of the boost converter at maximum output current, and RDS is the resistance of the MOSFET in the on state. The current rating of the output diode should be selected so that the power loss in the diode (given as the product of forward-voltage drop and the average diode Peak/RMS Current Calculation current) should be low enough to ensure that the junction For the purposes of setting the current limit, the peak temperature is within limits. This necessitates the diode current in the inductor and MOSFET can be calculated current rating to be in the order of 2 x IOUT to 3 x IOUT. as follows: Select fast-recovery diodes with a recovery time less than VOUT × D MAX × (1 − D MAX ) IOUT 50ns or Schottky diodes with low junction capacitance. IPK = + L INMIN × fSW (1 − D) MOSFET RMS Current Calculation The voltage stress on the MOSFET ideally equals the sum of the output voltage and the forward drop of the output diode. In practice, voltage overshoot and ringing occur due to action of circuit parasitic elements during the turn-off transition. The MOSFET voltage rating should be selected with the necessary margin to accommodate this extra voltage stress. A voltage rating of 1.3 x VOUT provides the necessary design margin in most cases. ×1.2 for D MAX < 0.5 IOUT 0.25 × VOUT And, IPK = + L INMIN × fSW (1 − D) ×1.2 for D MAX ≥ 0.5 ���������������������������������������������������������������� Maxim Integrated Products 23 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators DMAX, the maximum duty cycle, is obtained by substituting the minimum input operating voltage VINMIN in the equation above for duty cycle. LINMIN is the minimum value of the input inductor taking into account tolerance and saturation effects. Output Capacitor Selection The output capacitance can be calculated as follows: C OUT = I STEP × t RESPONSE ∆VOUT t RESPONSE ≅ ( 0.33 fC + 1 fSW RZ = 250 × VOUT 2 × C OUT × (1 − D MIN) IOUTMIN × L IN where DMIN is the duty cycle at the highest operating input voltage, and IOUTMIN is the minimum load current. CZ = VOUT × C OUT ) where ISTEP is the load step, tRESPONSE is the response time of the controller, DVOUT is the allowable output voltage deviation, and fC is the target closed-loop crossover frequency. fC is chosen to be one-tenth of the switching frequency fSW. For the boost converter, the output capacitor supplies the load current when the main switch is on; therefore, the output voltage ripple is a function of duty cycle and load current. Use the following equation to calculate the output capacitor ripple: IOUT × D MAX ∆VCOUT = C OUT × fSW Input Capacitor Selection The input ceramic capacitor value required can be calculated based on the ripple allowed on the input DC bus. The input capacitor should be sized based on the RMS value of the AC current handled by it. The calculations are: 3.75 × IOUT CIN = V f (1 D ) × × − MAX INMIN SW The input capacitor RMS current can be calculated as: I CIN_RMS = Error Amplifier Compensation Design The loop compensation values for the error amplifier can now be calculated as: ∆ILIN 2× 3 where: CP = 2 × IOUT × R Z 1 π × fSW × R Z Slope Compensation Ramp The slope required to stabilize the converter at duty cycles greater than 50% can be calculated as follows: SE = 0.5 × (0.82 × VOUT − VINMIN) L IN where LIN is in µH. Output Diodes Selection The design procedure for output-diode selection is identical to that outlined in the DCM Boost section. MOSFET RMS Current Calculation The voltage stress on the MOSFET ideally equals the sum of the output voltage and the forward drop of the output diode. In practice, voltage overshoot and ringing occur due to action of circuit parasitic elements during the turn-off transition. The MOSFET voltage rating should be selected with the necessary margin to accommodate this extra voltage stress. A voltage rating of 1.3 x VOUT provides the necessary design margin in most cases. The RMS current in the MOSFET is useful in estimating the conduction loss, and is given as: IMOSFETRMS = VOUT × D MAX × (1 − D MAX ) ∆ILIN = L INMIN × fSW for D MAX < 0.5, V/Fs, IOUT × D MAX (1 − D MAX ) where DMAX is the duty cycle at the lowest operating input voltage, and IOUT is the maximum load current. 0.25 × VOUT ∆ILIN = L INMIN × fSW for D MAX ≥ 0.5 ���������������������������������������������������������������� Maxim Integrated Products 24 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Typical Operating Circuits R15 402kI VIN R16 402kI VOUT D2 D6 T1 8 C9 2.2µF 50V R14 402kI 5 VOUT D4 C12 1µF 7 C19 OPEN R1 0I 6 VIN PGND C13 22µF 1 AC1 AC2 C15 22µF C16 22µF 1 R1 10 C1 0.1µF/ 275V AC C14 22µF 4 D1 L1 6.6mH R8 1.5MI C5 100µF 450V 3 R17 100kI R18 100kI C10 3300pF R7 1.5MI 11 D3 2 2 GND0 12 VIN PGND GND0 C6 0.47µF C7 47nF R10 0I DITHER/ SYNC PGND SLOPE VOUT NDRV R9 10kI DITHER/ SYNC DITHER / SYNC VIN R2 2.67MI R3 2.67MI R4 2.67MI RT C2 SHORT R13 10kI C3 SHORT DITHER / SYNC FB C11 330pF VFB R21 R22 0.1I 1.2kI N.C. 2 4 R27 20kI VDRV PGND C18 15000pF C17 47pF SGND C8 2.2µF EN/UVLO R28 562kI R25 OPEN VDRV VDRV 1 R20 100I CS C4 OPEN 5 VDRV PGND COMP R26 8.06kI 6 PGND MAX17595 R23 OPEN SGND N1 SGND R12 12.1kI VFB R19 0I SGND R11 OPEN SGND EN /UVLO VIN SS 2 R24 OPEN 3 1 U3 R29 49.9kI N.C. OVI SGND R5 75kI OVI R6 24.9kI SGND SGND SGND Figure 9. MAX17595 Typical Application Circuit (Universal Offline Isolated Power Supply) ���������������������������������������������������������������� Maxim Integrated Products 25 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Typical Operating Circuits (continued) VOUT VIN T1 VIN C1 18V TO 36V 47µF INPUT 50V C2 4.7µF 50V R1 7.5kI VOUT C9 22µF 16V C4 33nF, 50V D1 C3 0.22µF 50V PGND D2 C10 22µF 16V C11 22µF 16V 5V, 1.5A OUTPUT GND VIN SS EP C5 47nF NDRV SLOPE R2 SHORT R8 100I CS R3 10kI C6 300pF FB VFB R4 15kI R5 348kI EN /UVLO VCC COMP VDRV RT C7 2.2µF 16V R16 20kI VFB R13 511I R15 30.3kI C14 33pF U2 U3 DITHER C13 4.7nF R10 17.4kI R11 OPEN OVI R14 1kI C12 OPEN EN /UVLO R6 20kI OVI R12 OPEN VDRV SGND VIN VOUT R9 0.5I MAX17596 PGND PGND N1 2 3 1 R17 10kI C8 SHORT R7 10kI Figure 10. MAX17596 Typical Application Circuit (Power Supply for DC-DC Applications) ���������������������������������������������������������������� Maxim Integrated Products 26 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Typical Operating Circuits (continued) VIN VIN VIN C1 47µF 10.8V TO 13.2V DC C2 0.1µF EP PGND SS C3 47µF VDRV C4 2.2µF VDRV R1 120kI R2 9.92kI VIN SLOPE L1 220µH MAX17597 FB D1 R3 184kI SS26-TP VOUT NDRV R4 5kI C5 47nF COMP CS C6 120pF VIN C7 4.7µF/35V VOUT 24V, 0.3A N1 R8 100I C8 300pF R9 0.5I PGND RT R5 481kI R10 17.4kI EN /UVLO R11 OPEN R6 25kI DITHER OVI R7 49.9kI C9 SHORT SGND SGND PGND Figure 11. MAX17597 Typical Application Circuit (Nonsynchronous Boost Converter) ���������������������������������������������������������������� Maxim Integrated Products 27 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Layout, Grounding and Bypassing All connections carrying pulsed currents must be very short and as wide as possible. The inductance of these connections must be kept to an absolute minimum due to the high di/dt of the currents in high-frequency-switching power converters. This implies that the loop areas for forward and return pulsed currents in various parts of the circuit should be minimized. Additionally, small current loop areas reduce radiated EMI. Similarly, the heatsink of the MOSFET presents a dV/dt source; therefore, the surface area of the MOSFET heatsink should be minimized as much as possible. Ground planes must be kept as intact as possible. The ground plane for the power section of the converter should be kept separate from the analog ground plane, except for a connection at the least noisy section of the power ground plane, typically the return of the input filter capacitor. The negative terminal of the filter capacitor, the ground return of the power switch and current sensing resistor, must be close together. PCB layout also affects the thermal performance of the design. A number of thermal vias that connect to a large ground plane should be provided under the exposed pad of the part for efficient heat dissipation. For a sample layout that ensures first-pass success, refer to the MAX17595 evaluation kit layout available at www.maxim-ic.com. For universal AC input designs, follow all applicable safety regulations. Offline power supplies can require UL, VDE, and other similar agency approvals. Ordering Information/Selector Guide PART TEMP RANGE PIN PACKAGE MAX17595ATE+ -40NC to +125NC 16 TQFN-EP* Offline Flyback Controller MAX17596ATE+ -40NC to +125NC 16 TQFN-EP* Low-Voltage DC-DC Flyback Controller Boost Controller MAX17597ATE+ -40NC to +125NC 16 TQFN-EP* +Denotes a lead(Pb)-free/RoHS-compliant package. *Exposed pad. FUNCTIONALITY UVLO, VIN CLAMP DMAX 20V, Yes 46% 4V, No 46% 4V, No 93% Package Information For the latest package outline information and land patterns (footprints), go to www.maxim-ic.com/packages. Note that a “+”, “#”, or “-” in the package code indicates RoHS status only. Package drawings may show a different suffix character, but the drawing pertains to the package regardless of RoHS status. PACKAGE TYPE PACKAGE CODE OUTLINE NO. LAND PATTERN NO. 16 TQFN T1633+4 21-0136 90-0032 ���������������������������������������������������������������� Maxim Integrated Products 28 MAX17595/MAX17596/MAX17597 Peak-Current-Mode Controllers for Flyback and Boost Regulators Revision History REVISION NUMBER REVISION DATE 0 1/12 DESCRIPTION Initial release PAGES CHANGED — Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance. Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 © 2012 Maxim Integrated Products 29 Maxim is a registered trademark of Maxim Integrated Products, Inc.