AD ADP3031 2 mhz pwm boost switching regulator Datasheet

PRELIMINARY TECHNICAL DATA
2 MHz PWM Boost
Switching Regulator
ADP3031
a
Preliminary Technical Data
FEATURES
Up to 2 MHz PWM Frequency
Fully Integrated 1.5 A Power Switch
3% Output Regulation Accuracy
Adjustable Output Voltage from 3 V to 12 V
Simple Compensation
Small Inductor and MLC Capacitors
90% Efficiency
Under-voltage Lockout
Shutdown
APPLICATIONS
TFT LCD Bias Supplies
Portable Electronic Instruments
Industrial/Instrumentation Equipment
Functional Block Diagram
ERROR
AMP
REF
FB 2
1
ADP3031
gm
BIAS
5 SW
F/F
R Q
RAMP
GEN
S
COMPARATOR
RT 7
DRIVER
OSC
CURRENT
SENSE
AMPLIFIER
SD 3
GENERAL DESCRIPTION
The ADP3031 is a high frequency, step-up DC-DC
switching regulator capable of 12 V boosted output voltage
in a space saving MSOP-8 package. It provides high
efficiency, low-noise operation, and is easy to use. Capable
of operating up to 2 MHz, the high switching frequency and
PWM current mode architecture allow for excellent transient response, ease of noise filtering, and also small, costsaving external inductive and capacitive components. The
current limit and the power switch are integrated completely
on chip.
IN
6
COMP
8
GND
4
PGND
Capable of operating from 2.5 V to 5.5 V input, the
ADP3031 is ideal for Thin-Film Transistor (TFT) Liquid
Crystal Display (LCD) module applications, where local
point-of-use power regulation is required. Featuring an
adjustable output that supports voltages down to 3 V, the
ADP3031 is ideal to generate today’s low voltage rails,
delivering power efficiently, and simply with minimal
printed circuit board area.
The device is specified over the industrial temperature range
of -40°C to +85°C.
REV. PrB
8/2/02
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel:781/329-4700
World Wide Web Stie: http://www.analog.com
Fax:781/326-8703
©ANALOG DEVICES, INC., 2002
PRELIMINARY TECHNICAL DATA
ADP3031–SPECIFICATIONS1(VIN = +3.3 V, T = –40°C to 85°C, unless otherwise noted)
A
Parameter
SUPPLY
Input Voltage
Quiescent Current
Switching State3
Non-switching State
Shutdown
Symbol
IQSW
IQ
ISD
gm
AV
SWITCH
On Resistance
Output Load Current2
RON
ILOAD
Typ
Max
Units
5.5
V
2
300
5
500
1
mA
µA
µA
VIN = 2.5 V to 5.5 V
TBD
–0.1
1.233
TBD
+0.1
V
%/V
nA
%
µA/V
V/V
Line, Load, Temp
–3
100
Efficiency2
Maximum Duty Cycle
f = 600 kHz, light load
VFB
Leakage Current
Efficiency2
OSCILLATOR
Oscillator Frequency
Min
2.5
VIN
ERROR AMPLIFIER
Feedback Voltage
Line Regulation
Bias Current
Overall Regulation
Transconductance
Gain
Peak Current Limit
Output Voltage Range
Conditions
ICLSET
VOUT
3
100
1000
at ISW=1.5 A, VIN = 3.3 V
continuous operation
200
VIN = 3.3 V,VOUT =10 V
VSWITCH = 12 V
ILOAD = 200 mA,
VOUT = 10 V, f = 600 kHz
ILOAD = 100 mA,
VOUT = 10 V, f = 600 kHz
1.5
3
FOSC
RT = Open
RT = GND
DMAX
COMP = open, FB = 1 V
SHUTDOWN
Shutdown Input Voltage Low
Shutdown Input Voltage High
0.5
1.7
200
300
300
mΩ
mA
5
85
µA
%
90
%
1.8
7
12
A
V
0.6
2
80
0.7
2.3
85
MHz
MHz
%
0.8
V
V
2.5
V
mV
2.2
UNDER-VOLTAGE LOCKOUT
UVLO Threshold
UVLO Hysteresis
2.2
2.45
100
NOTES
1
All limits at temperature extremes are guaranteed via correlation and characterization using standard Statistical Quality Control (SQC).
2
See Figure xx.
3
This is the average current while switching.
Specifications subject to change without notice.
–2–
REV. PrB
PRELIMINARY TECHNICAL DATA
ADP3031
ABSOLUTE MAXIMUM RATINGS*
PIN FUNCTION DESCRIPTIONS
Input Voltage .............................................. –0.3 V to +6 V
SW Voltage ................................................................. 14 V
COMP Voltage ........................................ –0.3 V to +2.5 V
FB Voltage ............................................... –0.3 V to +1.3 V
SD Voltage .................................................. –0.3 V to +6 V
RT Voltage ............................................... –0.3 V to +1.2 V
PGND TO GND ................................................ ±200 mV
Operating Ambient Temperature Range ...... –40°C to 85°C
Operating Junction Temperature Range ..... –40°C to 125°C
Storage Temperature Range .................... –65°C to +150°C
θJA Two Layer ..................................................... 206°C/W
θJA Four Layer ..................................................... 142°C/W
Lead Temperature Range (Soldering, 60 sec.) .......... 300°C
Pin
Mnemonic
Function
1
COMP
Compensation Input.
2
FB
Feedback voltage sense input.
3
SD
Shutdown Input.
4
PGND
Power Ground. Ground return for
power transistor.
5
SW
Switching Output.
6
IN
Main Power Supply Input.
7
RT
Frequency Setting Input. A resistor
between this pin and GND sets the
switching frequency of the device.
8
GND
Analog Ground. The control
circuitry is referenced to this ground.
*This is a stress rating only; operation beyond these limits can cause the device
to be permanently damaged. Unless otherwise specified, all voltages are
referenced to GND
ORDERING GUIDE
Model
Voltage Package Branding
Output Option Information
PIN CONFIGURATION
ADP3031ARM
ADJ
COMP 1
MSOP-8 P8A
FB
2
SD 3
8
MSOP8
7
TOP VIEW
(Not to Scale) 6
IN
CC
VOUT
COMP
1
6
CIN
IN
ERROR
AMP
R1
REF
FB
ADP3031
L1
BIAS
gm
2
SW
COMPARATOR
R2
5
F/F
S
DRIVER
RT
7
OSC
RT
SD
3
CURRENT
SENSE
AMPLIFIER
GND
8
4
PGND
Figure 1. Typical Application
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the device features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
REV. PrB
–3–
D1
VOUT
COUT
R Q
RAMP
GEN
RT
IN
5 SW
PGND 4
RC
GND
PRELIMINARY TECHNICAL DATA
ADP3031
THEORY OF OPERATION
The ADP3031 is a boost converter driver which stores
energy from an input voltage in an inductor, and delivers
that energy, augmented by the input, to a load at a higher
output voltage. It includes a voltage reference and error
amplifier to compare some fraction of the load voltage to the
reference, and amplify any difference between them. The
amplified error signal is compared to a dynamic signal
produced by an internal ramp generator incorporating
switch current feedback. The comparator output timing sets
the duty ratio of a switch driving the inductor to maintain
the desired output voltage.
maintain steady state we can say that t1*VIN will equal
t2*(VOUT-VIN), if we neglect the effect of resistance in the
inductor and switch, and the forward voltage drop of the
diode. From this equality we can derive t1/T=1-VIN/VOUT,
where T is the period of a cycle, t1+t2. This result gives us
the switch duty ratio, t1/T in terms of the input and output
voltages.
In practice the duty ratio will need to be slightly higher than
this calculation. Because of series resistance in the inductor
and the switch, the voltage across the actual inductance is
somewhat less than applied VIN, and the actual output
voltage is less than our aproximation by the amount of the
diode forward voltage drop. However, the feedback control
within the ADP3031 will adjust the duty ratio to maintain
the output voltage. Changes in load current and input
voltage are also accomodated by the feedback control.
Referring to Figure 1, a typical application will power both
the IC and the inductor from the same input voltage. The
on chip MOSFET will be driven on, pulling pin SW close to
PGND. The resulting voltage across the inductor will cause
its current to increase aproximately linearly, with respect to
time.
Changes in load current alone require a change in duty
ratio, in order to change the average inductor current. But
once the inductor current adapts to the new load current,
the duty ratio should return to nearly its original value, as
we see from the duty cycle calculation which depends on
input and output voltages, but not on current. Increasing
the switch duty ratio initially reduces the output voltage,
until the average inductor current increases enough to offset
the reduction of the t2 interval. By limiting the duty ratio we
prevent this effect from regeneratively increasing the duty
ratio to 100%, which would cause the output to fall and the
switch current to rise without limit. The duty ratio is limited
to about 80% by the design of the Oscillator and an additional flip-flop reset.
When the MOSFET switch is turned off the inductor
current cannot drop to zero, and so this current drives the
SW node capacitance rapidly positive until the diode
becomes forward biased. The inductor current will now
begin to charge the load capacitor, causing a slight increase
in output voltage. Generally, the load capacitor is made
large enough that this increase is very small during the time
the switch is off. During this time inductor current is also
delivered to the load. In steady state operation, the inductor
current will exceed the load current, and the excess will be
what charges the load capacitor. The inductor current will
fall during this time, though not necessarily to zero.
During the next cycle, initiated by the on-chip oscillator, the
switch will again be turned on so that the inductor current
will be ramped back up. The charge on the load capacitor
will provide load current, during that interval. The remainder of the chip is arranged to control the duty ratio of the
switch, to maintain a chosen output voltage despite changes
in input voltage or load current.
A comparator compares the current sense amplifier output
to a factory set limit which resets the flip-flop, turning off
the switch. This prevents runaway or overload conditions
from damaging the switch and reflecting fault overloads
back to the input. Of course, the load is directly connected
to the input by way of the diode and inductor, so protection
against short circuited loads must be done at the power
input.
The output voltage is scaled down by a resistor voltage
divider and presented to the gm amplifier. This amplifier
operates on the difference between an on-chip reference and
the voltage at the FB pin so as to bring them to balance.
This will be when the output voltage equals the reference
voltage, multiplied by the resistor voltage divider ratio.
The gm amplifier has high voltage gain, to insure the output
voltage accuracy and invariance with load and input voltage.
However, because it is a gm amplifier with a specified
current response to input signal voltages, its high frequency
response can be controlled by the compensation impedance.
This permits the high frequency gain of the gm amplifier to
be optimized for the best compromise between speed of
response and frequency stability.
The gm amplifier drives an internal comparator, which has at
its other input a positive going ramp produced by the
Oscillator and modified by the current sense amplifier. The
MOSFET switch is turned on as the modified ramp voltage
rises. When this voltage exceeds the output of the gm
amplifier the comparator will turn off the switch, by resetting the flip-flop, previously set by the oscillator. The output
of the flip-flop is buffered by a high current driver which
turned on the MOSFET switch at the beginning of the
Oscillator cycle.
The stable closed loop bandwidth of the system can be
extended by the current feedback shown. A signal representing the magnitude of the switch current is added to the
ramp. This dynamically reduces the duty ratio, as the
current in the inductor increases, until the gm amplifier
restores it, improving the closed loop frequency stability.
The ADP3031 is intended to operate over a range of
frequencies, set by the RT pin. If the pin is open, the
oscillator runs at its lowest frequency: if the pin is
In the steady state with constant load and input voltage, the
current in the inductor will cycle around some average
current level. The increasing ramp of current will depend on
input voltage and t1, the switch on-time, while the decreasing ramp will depend on the difference between input and
output voltage and t2, the remainder of the cycle. In order
for the peaks of these two ramps to be equal and opposite to
–4–
REV. PrB
PRELIMINARY TECHNICAL DATA
ADP3031
“grounded” it runs at its highest frequency. A resistor from
RT to ground can be used to set intermediate operating
frequencies.
As a rule, powdered iron cores saturate softly, whereas
Ferrite cores saturate abruptly. Open drum core inductors
tend to saturate gradually, are low cost and are small in size,
making these types of inductors attractive in many applications. However, care must be exercised in their placement
because they have high magnetic fields. In applications that
are sensitive to magnetic fields, shielded geometries are
recommended.
Because of the large currents which flow in the main
MOSFET switch, it is provided with a separate PGND
return to the negative supply terminal, to avoid corrupting
the small signal return, GND, that can be used as a sense
line at the output load point.
In addition, inductor losses must be considered. Both core
and copper losses contribute to loss in converter efficiency.
To minimize core losses, look for inductors rated for
operation at high switching frequencies. To minimize
copper losses, it is best to use low dc resistance inductors.
Typically, it is best to use an inductor with a dc resistance
lower than 20 mΩ per µH.
APPLICATION INFORMATION
Frequency Selection
The ADP3031's frequency can be user selected to operate at either
600 KHz or 2 MHz and programmable by setting the RT pin. Tie
RT to GND for 2 MHz operation. For 600 KHz operation, float the
RT pin.
The nominal resistance at the RT pin to get a switching
frequency, fSW, is given by:
The inductor value can be estimated using the following:
L = (VOUT - VIN) × MSLOPE
Where MSLOPE = scaling factor for proper slope compensation.
RT (Ω) = 320,000 x (2,000,000 - fSW)/(3.6667 x fSW –
2,000,000)
(1)
Output Voltage
MSLOPE =
The ADP3031 features an adjustable output voltage range of
VIN to 12 V. The output voltage is fed back to the ADP3031
via resistor dividers R1 and R2 (Figure 1.). The feedback
voltage is 1.233 V, so the output voltage is set by the formula:
VOUT = 1.233 × ( 1+ R1/R2)
1.456
fSW
Choose the closest standard inductor value as a starting
point.
The corresponding peak inductor current can then be
calculated:
(2)
Since the feedback bias current is 100 nA maximum, R2 may
have a value up to 100 KΩ with minimum error due to the bias
current.

I L ( PEAK ) =  I OUT ×

VOUT
VIN
 1  VIN × (VOUT − VIN ) 
 (3)
+ 
 2  L × VOUT × f S

Inductor Selection
It is recommended to try several different inductor values,
sizes and types to find the best inductor for the application.
In general, large inductor values lead to lower ripple
current, less output noise, and either larger size or higher
DC resistance. Conversely, low inductor values lead to
higher ripple current, more noise, and either smaller size or
lower DC resistance. The final inductor selection should be
based on the best trade-off of size, cost, and performance.
For most of the applications, the inductor used with the
ADP3031 should be in the range of 2 µH to 22 µH. Several
inductor manufacturers are listed in Table 1. When selecting an inductor, it is important to make sure that the
inductor used with the ADP3031 is able to handle the peak
current without saturation and that the peak current is
below the current limit of the ADP3031.
Table 1. Inductor Manufacturers
Part
L(µH)
Max DC
Current
Max DCR
Ω
mΩ
Height
(mm)
CMD4D11-2R2MC
CMD4D11-4R7MC
CDRH4D28-100
CDRH5D18-220
CR43-4R7
CR43-100
2.2
4.7
10
22
4.7
10
0.95
0.75
1.00
0.80
1.15
1.04
116
216
128
290
109
182
1.2
1.2
3.0
2.0
3.5
3.5
Sumida
847-956-0666
www.sumida.com
DS1608-472
DS1608-103
4.7
10
1.40
1.00
60
75
2.9
2.9
Coilcraft
847-639-6400 www.coilcraft.com
D52LC-4R7M
D52LC-100M
4.7
10
1.14
0.76
87
150
2.0
2.0
Toko
847-297-0070 www.tokoam.com
REV. PrB
–5–
Vendor
PRELIMINARY TECHNICAL DATA
ADP3031
Capacitor Selection
Loop Compensation
The ADP3031 requires an input capacitor to reduce the
switching ripple and noise on the IN pin. The value of the
input capacitor will be dependent on the application. For
most applications, a minimum of 10 µF is required. For
applications that are running close to current limit or that
have large transient loads, input capacitors in the range of
22 µF to 47 µF are required. The selection of the output
capacitor is also dependent on the application. Given the
allowable output ripple voltage, ∆VOUT, the criteria for
selecting the output capacitor can be calculated using the
following equations:
Like most current programmed PWM converters, the
ADP3031 needs compensation to maintain stability over the
operating conditions of the particular application. For
operation at duty cycles above 50%, the choice of inductor
is critical in maintaining stability. If the slope of the inductor
current is too small or too large, the circuit will be unstable.
See Inductor Selection for more information on choosing
the proper inductor.

COUT ≥ 8 × I OUT 
(VOUT − VIN )
 f s × VOUT × ∆VOUT
ESRCOUT ≤



∆VOUT
I L ( PEAK )
The ADP3031 provides a pin (COMP) for compensating
the voltage feedback loop. This is done by connecting a
series RC network from the COMP pin to GND. See
Figure 2. For most applications, the compensation resistor,
RC, should be in the range of 30 kΩ < RC < 300 kΩ and the
compensation capacitor, CC, in the range of 100 pF < CC <
4 nF.
(4)
(5)
ERROR AMP
COMP
REF
gm
When selecting an output capacitor, make sure that the
ripple current rating is sufficient to cover the rms switching
current of the ADP3031.
FB
RC
C2
CC
The ripple current in the output capacitor is given by the
following:
I RMS (COUT ) = IOUT
VOUT − VIN
VIN
Figure 2. Compensation Components
(6)
Shutdown
The ADP3031 shuts down to reduce the supply current to
1 µA maximum when the shutdown pin is pulled low. In this
mode, the internal reference, error amplifier, comparator,
biasing circuitry, and the internal MOSFET switch are
turned off. Note that the output is still connected to the
input via the inductor and Schottky diode when in shutdown.
Multi-layer ceramic capacitors are a good choice, as they
have low ESR, high ripple current rating and a very small
package size. Tantalum or OS-CON capacitors can be used,
however they have a larger package size and have higher
ESR. Table 2 shows a list of several capacitor manufacturers. Consult the manufacturer for more information.
Table 2. Capacitor Manufacturers
Layout Procedure
Vendor
Phone #
Web Address
AVX
408-573-4150
www.avxcorp.com
Murata
714-852-2001
www.murata.com
Sanyo
408-749-9714
www.sanyovideo.com
Taiyo-Yuden
408-573-4150
www.t-yuden.com
In order to get high efficiency, good regulation and stability,
a good printed circuit board layout is required. It is strongly
recommended that the evaluation board layout be followed
as closely as possible. Use the following general guidelines
when designing printed circuit boards (refer to Figure 1):
1. Keep CIN close to the IN pin of the ADP3031.
2. Keep the high current path from CIN, through L1, to the
SW pin and PGND pin as short as possible.
Diode Selection
3. Similarly, keep the high current path from CIN, through
L1, D1, and COUT as short as possible.
In specifying a diode, consideration must be given to speed,
the forward current, the forward voltage drop, reverse
leakage current, and the breakdown voltage. The output
diode should be rated to handle the maximum output
current. If the output can be subjected to accidental short
circuits then the diode must be rated to handle currents up
to the current limit of the ADP3031. The breakdown rating
of the diode must exceed the output voltage. A high-speed
diode with low forward drop, and low leakage will help
improve the efficiency of the converter by lowering the
losses of the diode. Schottky diodes are recommended.
4. High current traces should be kept as short and as wide
as possible.
5. Place the compensation components as close to the
COMP pin as possible.
6. Place the feedback resistors as close to the FB pin as
possible to prevent noise pickup.
7. Avoid routing noise sensitive traces near the high current
traces and components.
–6–
REV. PrB
PRELIMINARY TECHNICAL DATA
ADP3031
R5
200Ω
D1
BAV99
VGL
C4
10nF
C7
10µ F
D7
BZX84C5V1
C1
10µ F
R1
200kΩ
C15
100pF
R3
100kΩ
1 COMP
GND 8
RT 7
C13
1µ F
3 SD
IN 6
OPTIONAL
10µ H
D8
1N5818
SW 5
C8
10µ F
* FREQUENCY ADJUST
Figure 3. Typical Application for TFT LCD
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
RM-8
8-Lead Mini/micro SOIC Package [Mini_SO]
0.122 (3.10)
0.114 (2.90)
8
5
0.122 (3.10)
0.114 (2.90)
0.199 (5.05)
0.187 (4.75)
1
4
PIN 1
0.0256 (0.65) BSC
0.120 (3.05)
0.112 (2.84)
0.120 (3.05)
0.112 (2.84)
0.006 (0.15)
0.002 (0.05)
0.043 (1.09)
0.037 (0.94)
0.018 (0.46)
SEATING 0.008 (0.20)
PLANE
REV. PrB
+VGH
D9
BZX84C22
GND
L1
R2*
R4
866kΩ
GND
C14
1µ F
D5
BAV99
2 FB
4 PGND
SD
R6
1kΩ
D4 BAV99
ADP3031
R7
10kΩ
D3
BAV99
D6
C6 BAV99
10nF
D2
BAV99
VIN+
C5
10nF
0.011 (0.28)
0.003 (0.08)
–7–
33ⴗ
27ⴗ
0.028 (0.71)
0.016 (0.41)
OUT+
C9
10µ F
C10
10µ F
RTN-
PRELIMINARY TECHNICAL DATA
PRINTED IN U.S.A.
000000000
ADP3031
–8–
REV. PrB
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