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LM4651 & LM4652 Overture™ Audio Power Amplifier 170W Class D Audio Power
Amplifier Solution
Check for Samples: LM4651, LM4652
FEATURES
DESCRIPTION
•
•
•
•
•
•
•
The IC combination of the LM4651 driver and the
LM4652 power MOSFET provides a high efficiency,
Class D subwoofer amplifier solution.
1
23
•
Conventional Pulse Width Modulation.
Externally Controllable Switching Frequency.
50kHz to 200kHz Switching Frequency Range.
Integrated Error Amp and Feedback Amp.
Turn−on Soft Start and Under Voltage Lockout.
Over Modulation Protection (Soft Clipping).
Externally Controllable Output Current
Limiting and Thermal Shutdown Protection.
Self Checking Protection Diagnostic.
APPLICATIONS
•
•
•
Powered Subwoofers for Home Theater and
PC's
Car Booster Amplifier
Self-powered Speakers
KEY SPECIFICATIONS
•
•
•
•
Output power into 4Ω with < 10% THD. 170W
(Typ)
THD at 10W, 4Ω, 10 − 500Hz. < 0.3% THD (Typ)
Maximum efficiency at 125W 85% (Typ)
Standby attenuation. >100dB (Min)
The LM4651 is a fully integrated conventional pulse
width modulator driver IC. The IC contains short
circuit, under voltage, over modulation, and thermal
shut down protection circuitry. The LM4651also
contains a standby function which shuts down the
pulse width modulation minimizing supply current.
The LM4652 is a fully integrated H-bridge power
MOSFET IC in a TO-220 power package. The
LM4652 has a temperature sensor built in to alert the
LM4651 when the die temperature of the LM4652
exceeds the threshold. Together, these two IC's form
a simple, compact high power audio amplifier solution
complete with protection normally seen only in Class
AB amplifiers. Few external components and minimal
traces between the IC's keep the PCB area small and
aids in EMI control.
The near rail-to-rail switching amplifier substantially
increases the efficiency compared to Class AB
amplifiers. This high efficiency solution significantly
reduces the heat sink size compared to a Class AB
IC of the same power level. This two-chip solution is
optimum for powered subwoofers and self powered
speakers.
1
2
3
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Overture is a trademark of dcl_owner.
All other trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
Copyright © 2000–2013, Texas Instruments Incorporated
OBSOLETE
LM4651, LM4652
SNOS507H – MAY 2000 – REVISED APRIL 2013
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Connection Diagram
Figure 1. LM4651 Plastic Package - Top View
See Package Number N28B
(1)
Figure 2. LM4652 Plastic Package (1)
Isolated TO-220 Package
See Package Number NDB0015B
or
Non-Isolated TO-220 Package
See Package Number NDL0015A
The LM4652TA package NDL0015A is a non-isolated package, setting the tab of the device and the heat sink at −V potential when the
LM4652 is directly mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound,
θCS (case to sink) is increased, but the heat sink will be isolated from −V.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
2
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Absolute Maximum Ratings (1) (2) (3)
Supply Voltage
± 22V
Output Current (LM4652)
10A
Power Dissipation (LM4651) (4)
1.5W
Power Dissipation (LM4652) (4)
32W
(5)
LM4652 (pins 2,6,10,11)
500V 2000V
ESD Susceptibility (LM4651) (6)
LM4652 (pins 2,6,10,11)
100V 200V
ESD Susceptibility (LM4651)
Junction Temperature (7)
150°C
Soldering Information
N, NDL and NDB Package (10 seconds)
(1)
(2)
(3)
(4)
(5)
(6)
(7)
260°C
−40°C to + 150°C
Storage Temperature
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
All voltages are measured with respect to the GND pin unless otherwise specified.
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and
specifications.
For operating at case temperatures above 25°C, the LM4651 must be de−rated based on a 150°C maximum junction temperature and a
thermal resistance of θJA = 62 °C/W (junction to ambient), while the LM4652 must be de−rated based on a 150°C maximum junction
temperature and a thermal resistance of θJC = 2.0 °C/W (junction to case) for the isolated package (NDB) or a thermal resistance of θJC
= 1.0°C/W (junction to case) for the non-isolated package (NDL).
Human body model, 100 pF discharged through a 1.5 kΩ resistor.
Machine Model, 220pF-240pF discharge through all pins.
The operating junction temperature maximum, Tjmax is 150°C.
Operating Ratings (1) (2)
−40°C ≤ TA ≤ +85°C
Temperature Range
Supply Voltage |V+| + |V−|
Thermal Resistance
22V to 44V
LM4651 N Package
LM4652 NDB, TO−220 Package
LM4652 NDL, TO−220 Package
(1)
(2)
θJA
52°C/W
θJC
22°C/W
θJA
43°C/W
θJC
2.0°C/W
θJA
37°C/W
θJC
1.0°C/W
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
All voltages are measured with respect to the GND pin unless otherwise specified.
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System Electrical Characteristics for LM4651 and LM4652 (1) (2)
The following specifications apply for +VCC = +20V, −VEE = −20V, f SW = 125kHz, fIN = 100Hz, RL = 4Ω, unless otherwise
specified. Typicals apply for TA = 25°C. For specific circuit values, refer to Figure 3 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
See (3)
ICQ
Total Quiescent Power Supply
Current
VIN = 0V, IO = 0mA
RDLY = 0Ω
RDLY = 10kΩ
ISTBY
Standby Current
AM
Standby Attenuation
PO
Output Power (Continuous Average)
Units
(Limits)
237
124
mA
mA
VPIN13 = 5V, Stby: On
17
mA
VPIN13 = 5V, Stby: On
>115
dB
RL = 4Ω, 1% THD
125
W
RL = 4Ω, 10% THD
155
W
RL = 8Ω, 1% THD
75
W
RL = 8Ω, 10% THD
90
W
fSW = 75kHz, RL = 4Ω, 1% THD
135
W
fSW = 75kHz, RL = 4Ω, 10% THD
170
W
η
Efficiency at PO = 5W
PO = 5W, RDLY = 5kΩ
55
%
η
Efficiency
(LM4651 & LM4652)
PO = 125W, THD = 1%
85
%
Power Dissipation
(LM4651 + LM4652)
PO = 125W, THD = 1% (max)
22
W
Pd
fSW = 75kHz, PO = 135W,
THD = 1% (max)
22
W
THD+N
Total Harmonic Distortion Plus Noise
10W, 10Hz ≤ fIN ≤ 500Hz,
AV = 18dB
10Hz ≤ BW ≤ 80kHz
0.3
%
εOUT
Output Noise
A Weighted, no signal, RL = 4Ω
550
µV
A-Wtg, Pout = 125W, RL = 4Ω
92
dB
22kHz BW, Pout = 125W, RL = 4Ω
89
dB
0.07
V
37
dB
SNR
Signal-to-Noise Ratio
VOS
Output Offset Voltage
VIN = 0V, IO = 0mA, ROFFSET = 0Ω
PSRR
Power Supply Rejection Ratio
RL = 4Ω, 10Hz ≤ BW ≤ 30kHz
+VCCAC = −VEEAC = 1VRMS,
fAC = 120Hz
(1)
(2)
(3)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
All voltages are measured with respect to the GND pin unless otherwise specified.
Limits are guaranteed to TI's AOQL (Average Outgoing Quality Level).
Electrical Characteristics for LM4651 (1) (2) (3)
The following specifications apply for +VCC = +20V, −VEE = −20V, fSW = 125kHz, unless otherwise specified. Limits apply for
TA = 25°C. For specific circuit values, refer to Figure 3 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
See
ICQ
(1)
(2)
(3)
4
Total Quiescent Current
LM4652 not connected, IO = 0mA,
|VCC+| + |VEE-|, RDLY = 0Ω
36
(3)
15
45
Units
(Limits)
mA (min)
mA (max)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
All voltages are measured with respect to the GND pin unless otherwise specified.
Limits are guaranteed to TI's AOQL (Average Outgoing Quality Level).
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Electrical Characteristics for LM4651(1)(2)(3) (continued)
The following specifications apply for +VCC = +20V, −VEE = −20V, fSW = 125kHz, unless otherwise specified. Limits apply for
TA = 25°C. For specific circuit values, refer to Figure 3 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
See (3)
VIL
Standby Low Input Voltage
Not in Standby Mode
VIH
Standby High Input Voltage
In Standby Mode
2.0
ROSC = 15kΩ
65
ROSC = 0Ω
200
Units
(Limits)
0.8
V (max)
2.5
V (min)
kHz
fSW
Switching Frequency Range
fSWerror
50% Duty Cycle Error
ROSC = 4kΩ, fSW = 125kHz
1
Tdead
Dead Time
RDLY = 0Ω
27
ns
TOverMod
Over Modulation Protection Time
Pulse Width Measured at 50%
310
ns
kHz
3
% (max)
Electrical Characteristics for LM4652 (1) (2) (3)
The following specifications apply for +VCC = +20V, −VEE = −20V, unless otherwise specified. Limits apply for TA = 25°C. For
specific circuit values, refer to Figure 3 (Typical Audio Application Circuit).
LM4651 & LM4652
Symbol
Parameter
Conditions
Typical
Limit
See (3)
Units
(Limits)
V(BR)DSS
Drain−to−Source Breakdown Voltage
VGS = 0
55
V
IDSS
Drain−to−Source Leakage Current
VDS = 44VDC, VGS = 0V
1.0
mA
VGSth
Gate Threshold Voltage
VDS = VGS, ID = 1mADC
0.85
RDS(ON)
Static Drain−to−Source On Resistance VGS = 6VDC, ID = 6ADC
200
tr
Rise Time
VGD = 6VDC, VDS = 40VDC,
RGATE = 0Ω
25
ns
tf
Fall Time
VGD = 6VDC, VDS = 40VDC,
RGATE = 0Ω
26
ns
ID
Maximum Saturation Drain Current
VGS = 6VDC, VDS = 10VDC
10
(1)
(2)
(3)
V
300
8
mΩ (max)
ADC (min)
Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical
specifications under particular test conditions which guarantee specific performance limits. This assumes that the device is within the
Operating Ratings. Specifications are not guaranteed for parameters where no limit is given, however, the typical value is a good
indication of device performance.
All voltages are measured with respect to the GND pin unless otherwise specified.
Limits are guaranteed to TI's AOQL (Average Outgoing Quality Level).
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Typical Application
Figure 3. Typical Application Circuit and Test Circuit
LM4651 PIN DESCRIPTIONS
6
Pin No.
Symbol
1
OUT1
The reference pin of the power MOSFET output to the gate drive circuitry.
Description
2,27
BS1,BS2
The bootstrap pin provides extra bias to drive the upper gates, HG1,HG2.
3
HG1
High−Gate #1 is the gate drive to a top side MOSFET in the H-Bridge.
4
HG2
High−Gate #2 is the gate drive to a top side MOSFET in the H-Bridge.
5,15
GND
The ground pin for all analog circuitry.
6
+6VBYP
7
+VCC
8
−6VBYP
The internally regulated negative voltage output for analog circuitry. This pin is available for internal
regulator bypassing only.
9
FBKVO
The feedback instrumentation amplifier output pin.
10
ERRIN
The error amplifier inverting input pin. The input audio signal and the feedback signal are summed at
this input pin.
11
ERRVO
The error amplifier output pin.
12
TSD
13
STBY
Standby function input pin. This pin is CMOS compatible.
14
FBK1
The feedback instrumentation amplifier pin. This must be connected to the feedback filter from VO1
(pin 15 on the LM4652 ).
16
OSC
The switching frequency oscillation pin. Adjusting the resistor from 15.5kΩ to 0Ω changes the
switching frequency from 75kHz to 225kHz.
17
Delay
The dead time setting pin.
The internally regulated positive voltage output for analog circuitry. This pin is available for internal
regulator bypassing only.
The positive supply input for the IC.
The thermal shut down input pin for the thermal shut down output of the LM4652.
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LM4651 PIN DESCRIPTIONS (continued)
18
SCKT
Short circuit setting pin. Minimum setting is 10A.
19
FBK2
The feedback instrumentation amplifier pin. This must be connected to the feedback filter from VO2
(pin 7 on the LM4652 ).
20,21
−VDDBYP
22,23
−VEE
24
START
25
LG1
Low−Gate #1 is the gate drive to a bottom side MOSFET in the H-Bridge.
26
LG2
Low−Gate #2 is the gate drive to a bottom side MOSFET in the H-Bridge.
28
OUT2
The reference pin of the power MOSFET output to the gate drive circuitry.
The regulator output for digital blocks. This pin is for bypassing only.
The negative voltage supply pin for the IC.
The start up capacitor input pin. This capacitor adjusts the start up time of the diagnostic sequence
for the modulator. Refer to Start Up Sequence and Self-Diagnostic Timing in the Application
Information section.
LM4652 PIN DESCRIPTIONS (1)
Pin No.
Symbol
1
GND
A ground reference for the thermal shut down circuitry.
2
LG1
Low−Gate #1 is the gate input to a bottom side MOSFET in the H-Bridge.
3
−VEE
The negative voltage supply input for the power MOSFET H-Bridge.
4
TSD
The thermal shut down flag pin. This pin transitions to 6V when the die temperature exceeds 150°C.
5
NC
No connection
6
LG2
Low−Gate #2 is the gate input to a bottom side MOSFET in the H-Bridge.
7
VO2
The switching output pin for one side of the H-Bridge.
8
NC
No connection.
9
NC
No connection.
10
HG2
High−Gate #2 is the gate input to a top side MOSFET in the H-Bridge.
11
NC
No connection.
12
NC
No connection.
13
+VCC
The positive voltage supply input for the power MOSFET H-Bridge.
14
HG1
High−Gate #1 is the gate input to a top side MOSFET in the H-Bridge.
15
VO2
The switching output pin for one side of the H-Bridge.
(1)
Description
Note: NC, no connect pins are floating pins. It is best to connect the pins to GND to minimize any noise from being coupled into the
pins.
External Components Description
(Refer to Figure 3)
Components
Functional Description
1.
R1
Works with R2, Rfl1 and Rfl2 to set the gain of the system. Gain = {[R2/(R1 + 100)] x [(Rfl1 + Rfl2)/Rfl2] −
[R2/(R1 + 100)] + 0.5} + [(VCC - 20) * 0.0175] (V/V).
2.
R2
See description above for R1.
3.
Rf
Sets the gain and bandwidth of the system by creating a low pass filter for the Error Amplifier's
feedback with Cf. 3dB pole is at fC = 1/(2πRfCf) (Hz).
4.
Cf
See description above for Rf.
5.
RfI1
Provides a reduction in the feedback with RfI2. RfI1should be 10 X RfI2 minimum to reduce effects on
the pole created by RfI2 and CfI1. See also note for R1, R2 for effect on System Gain.
6.
RfI2
RfI2 and CfI1 creates a low pass filter with a pole at fC = 1/(2πRfI2CfI1) (Hz). See also note for R1, R2
for effect on System Gain.
7.
CfI1
See description above for RfI2.
8.
RfI3
Establish the second pole for the low pass filter in the feedback path at fC = 1/(2πRfI3CfI2) (Hz).
9.
CfI2
See description above for RfI3.
10.
L1
Combined with CBYP creates a 2−pole, low pass output filter that has a −3dB pole at fC =
1/{2π[L1(2CBYP + C1)]1/2} (Hz).
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Filters the commom mode high frequency noise from the amplifier's outputs to GND. Recommended
value is 0.1µF to 1µF.
11.
C1
12.
Cbyp
13.
CB1−CB4
14.
CBT
Provides the bootstrap capacitance for the boot strap pin.
15.
RDLY
Sets the dead time or break before make time to TDLY = (1.7x10−12)(500 + RDLY) (seconds) or RDLY =
[TDLY/(1.7x10−12)] - 500 (Ω).
16.
CSTART
Controls the startup time with TSTART = (8.5x104) CSTART (seconds) or CSTART = TSTART /(8.5x104)
(F).
17.
RSCKT
Sets the output current limit with ISCKT = (1x105)/(10kΩ ‖ RSCKT) (A) or RSCKT = [(1x109)/ISCKT] / [10k (1x105/ISCKT)] (Ω).
18.
ROSC
Controls the switching frequency with fSW = 1x109 / (4000 + ROSC) (Hz) or ROSC = (1x109/fSW) - 4000
(Ω).
See description for L1.
Bypass capacitors for VCC, VEE, analog and digital voltages (VDD, +6V, −6V). See Supply Bypassing
and High Frequency PCB Design in the Application Information section for more information.
19.
D1
20.
CSBY1, CSBY2, CSBY3
Schottky diode to protect the output MOSFETs from fly back voltages.
21.
ROFFSET
Provides a small DC voltage at the input to minimize the output DC offset seen by the load. This also
minimize power on pops and clicks.
22.
CIN
Blocks DC voltages from being coupled into the input and blocks the DC voltage created by ROFFSET
from the source.
23.
Rgate
Supply de-coupling capacitors. See Supply Bypassing in the Application Information section.
Slows the rise and fall time of the gate drive voltages that drive the output FET's.
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Typical Performance Characteristics
Output Power
vs.
Supply Voltage
Output Power
vs.
Supply Voltage
Figure 4.
Figure 5.
THD+N
vs.
Output Power
RL = 4Ω
THD+N
vs.
Output Power
RL = 8Ω
Figure 6.
Figure 7.
THD+N
vs.
Output Power
RL = 4Ω
THD+N
vs.
Output Power
RL = 8Ω
Figure 8.
Figure 9.
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Typical Performance Characteristics (continued)
10
THD+N
vs.
Frequency
vs.
Bandwidth
RL = 4Ω
THD+N
vs.
Frequency
vs.
Bandwidth
RL = 8Ω
Figure 10.
Figure 11.
THD+N
vs.
Frequency
vs.
Bandwidth
RL = 4Ω
THD+N
vs.
Frequency
vs.
Bandwidth
RL = 8Ω
Figure 12.
Figure 13.
Power Dissipation & Efficiency
vs. Output Power
Clipping Power Point & Efficiency
vs. Switching Frequency (fSW)
Figure 14.
Figure 15.
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Typical Performance Characteristics (continued)
Frequency Response
RL = 4Ω
Supply Current
vs.
Switching Frequency
(LM4651 & LM4652)
Figure 16.
Figure 17.
Supply Current
vs.
Supply Voltage
(LM4651 & LM4652)
RDS(ON)
vs.
Temperature
Figure 18.
Figure 19.
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APPLICATION INFORMATION
GENERAL FEATURES
System Functional Information
The LM4651 is a conventional pulse width modulator/driver. As Figure 20 shows the incoming audio signal is
compared with a triangle waveform with a much higher frequency than the audio signal (not drawn to scale). The
comparator creates a variable duty cycle squarewave. The squarewave has a duty cycle proportional to the
audio signal level. The squarewave is then properly conditioned to drive the gates of power MOSFETs in an Hbridge configuration, such as the LM4652. The pulse train of the power MOSFETs are then fed into a low pass
filter (usually a LC) which removes the high frequency and delivers an amplified replica of the audio input signal
to the load.
Figure 20. Conventional Pulse Width Modulation
Standby Function
The standby function of the LM4651 is CMOS compatible, allowing the user to perform a muting of the music by
shutting down the pulse width waveform. Standby has the added advantage of minimizing the quiescent current.
Because standby shuts down the pulse width waveform, the attenuation of the music is complete (>120dB), EMI
is minimized, and any output noise is eliminated since there is no modulation waveform. When in Standby mode,
the outputs of the LM4652 will both be at VCC. By placing a logic "1" or 5V at pin 13, the standby function will be
enabled. A logic "0" or 0V at pin 13 will disable the standby function allowing modulation by the input signal.
Under Voltage Protection
The under voltage protection disables the output driver section of the LM4651 while the supply voltage is below ±
10.5V. This condition can occur as power is first applied or when low line, changes in load resistance or power
supply sag occurs. The under voltage protection ensures that all power MOSFETs are off, eliminating any shootthrough current and minimizing pops or clicks during turn-on and turn-off. The under voltage protection gives the
digital logic time to stabilize into known states providing a popless turn on.
Start Up Sequence and Self-Diagnostic Timing
The LM4651 has an internal soft start feature (see Figure 21) that ensures reliable and consistent start-up while
minimizing turn-on thumps or pops. During the start-up cycle the system is in standby mode. This start-up time is
controlled externally by adjusting the capacitance (CSTART) value connected to the START pin. The start-up time
can be controlled by the capacitor value connected to the START pin given by Equation 1 or Equation 2:
tSTART = (8.4x104)CSTART
CSTART = TSTART/(8.5x104)
(seconds)
(Farads)
(1)
(2)
The value of CSTART sets the time it takes for the IC to go though the start-up sequence and the frequency that
the diagnostic circuitry checks to see if an error condition has been corrected. An Error condition occurs if current
limit, thermal shut down, under voltage detection, or standby are sensed. The self-diagnostic circuit checks to
see if any one of these error flags has been removed at a frequency set by the CSTART capacitor. For example, if
the value of CSTART is 10µF then the diagnostic circuitry will check approximately every second to see if an error
condition has been corrected. If the error condition is no longer present, the LM4651/52 will return to normal
operation.
12
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Figure 21. Startup Timing Diagram
Current Limiting and Short Circuit Protection
The resistor value connected between the SCKT pin and GND determines the maximum output current. Once
the output current is higher than the set limit, the short circuit protection turns all power MOSFETs off. The
current limit is set to a minimum of 10A internally but can be increased by adjusting the value of the RSCKT
resistor. Equation 3 shows how to find RSCKT.
ISCKT = 1X105/(10kΩ‖ RSCKT)
(Amps)
(3)
This feature is designed to protect the MOSFETs by setting the maximum output current limit under short circuit
conditions. It is designed to be a fail-safe protection when the output terminals are shorted or a speaker fails and
causes a short circuit condition.
Thermal Protection
The LM4651 has internal circuitry (pin 12) that is activated by the thermal shutdown output signal from the
LM4652 (pin 4). The LM4652 has thermal shut down circuitry that monitors the temperature of the die. The
voltage on the TSD pin (pin 4 of the LM4652) goes high (6V) once the temperature of the LM4652 die reaches
150°C. This pin should be connected directly to the TSD pin of the LM4651 (pin 12). The LM4651 disables the
pulse width waveform when the LM4652 transmits the thermal shutdown flag. The pulse width waveform remains
disabled until the TSD flag from the LM4652 goes low, signaling the junction temperature has cooled to a safe
level.
Dead Time Setting
The DELAY pin on the LM4651 allows the user to set the amount of dead time or break before make of the
system. This is the amount of time one pair of FETs are off before another pair is switched on. Increased dead
time will reduce the shoot through current but has the disadvantage of increasing THD. The dead time should be
reduced as the desired bandwidth of operation increases. The dead time can be adjusted with the RDLY resistor
by Equation 4:
TDLY = 1.7x10−12 (500 + RDLY)
(Seconds)
(4)
Currently, the recommended value is 5kΩ.
Oscillator Control
The modulation frequency is set by an external resistor, ROSC, connected between pin 16 and GND. The
modulation frequency can be set within the range of 50kHz to 225kHz according to the design requirements. The
values of ROSC and fOSC can be found by Equation 5 and Equation 6:
fOSC = 1x109/ (4000 + ROSC)
ROSC = (1x109/ fOSC) − 4000
(Hz)
(Ω)
(5)
(6)
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Equation 5 and Equation 6 are for RDLY = 0. Using a value of RDLY greater than zero will increase the value
needed for ROSC. For RDLY = 5kΩ, ROSC will need to be increased by about 2kΩ. As the graphs show, increasing
the switching frequency will reduce the THD but also decreases the efficiency and maximum output power level
before clipping. Increasing the switching frequency increases the amount of loss because switching currents
lower the efficiency across the output power range. A higher switching frequency also lowers the maximum
output power before clipping or the 1% THD point occur.
Over-Modulation Protection
The over-modulation protection is an internally generated fixed pulse width signal that prevents any side of the Hbridge power MOSFETs from remaining active for an extended period of time. This condition can result when the
input signal amplitude is higher than the internal triangle waveform. Lack of an over modulation signal can
increase distortion when the amplifier's output is clipping. Figure 22 shows how the over modulation protection
works.
Figure 22. Over Modulation Protection
The over modulation protection also provides a "soft clip" type response on the top of a sine wave. This minimum
pulse time is internally set and cannot be adjusted. As the switching frequency increases this minimum time
becomes a higher percentage of the period (TPERIOD = 1/fSW). Because the over modulation protection time is a
higher percentage of the period, the peak output voltage is lower and, therefore, the output power at clipping is
lower for the same given supply rails and load.
Feedback Amplifier and Filter
The purpose of the feedback amplifier is to differentially sample the output and provide a single-ended feedback
signal to the error amplifier to close the feedback loop. The feedback is taken directly from the switching output
before the demodulating LC filter to avoid the phase shift caused by the output filter. The signal fed back is first
low pass filtered with a single pole or dual pole RC filter to remove the switching frequency and its harmonics.
The differential signal, derived from the bridge output, goes into the high input impedance instrumentation
amplifier that is used as the feedback amplifier. The instrumentation amplifier has an internally fixed gain of 1.
The use of an instrumentation amplifier serves two purposes. First, it's input are high impedance so it doesn't
load down the output stage. Secondly, an IA has excellent common-mode rejection when its gain setting
resistors are properly matched. This feature allows the IA to derive the true feedback signal from the differential
output, which aids in improving the system performance.
Figure 23. Feedback instrumentation Amplifier Schematic
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Error Amplifier
The purpose of the error amplifier is to sum the input audio signal with the feedback signal derived from the
output. This inverting amplifier's gain is externally configurable by resistors Rf and R1. The parallel feedback
capacitor and resistor form a low pass filter that limits the frequency content of the input audio signal and the
feedback signal. The pole of the filter is set by Equation 7.
fIP = 1/(2πRfCf)
(Hz)
(7)
On-Board Regulators
The LM4651 has its own internal supply regulators for both analog and digital circuits. Separate ±6V regulators
exist solely for the analog amplifiers, oscillator and PWM comparators. A separate voltage regulator powers the
digital logic that controls the protection, level shifting, and high−/low−side driver circuits. System performance is
enhanced by bypassing each regulator's output. The ±6V regulator outputs, labeled +6VBYP (pin 6) and −6VBYP
(pin 8) should be bypassed to ground. The digital regulator output, −VDDBYP (pins 20 & 21) should be bypassed to
−VEE (pins 22 & 23). The voltage level of −VDDBYP should be always be 6V closer to ground than the negative
rail, −VEE. As an example, if −VEE = −20V, then −VDDBYP should equal −14V. Recommended capacitor values
and type can be found in Figure 3.
APPLICATIONS HINTS
Introduction
Texas Instruments (TI) is committed to providing application information that assists our customers in obtaining
the best performance possible from our products. The following information is provided in order to support this
commitment. The reader should be aware that the optimization of performance was done using a reference PCB
designed by NSC and shown in Figure 25 through Figure 29. Variations in performance can occur because of
physical changes in the printed circuit board and the application. Therefore, the designer should know that
component value changes may be required in order to optimize performance in a given application. The values
shown in this data sheet can be used as a starting point for evaluation purposes. When working with high
frequency circuits, good layout practices are also critical to achieving maximum performance.
Input Pre-Amplifier with Subwoofer Filter
The LM4651 and LM4652 Class D solution is designed for low frequency audio applications where low gain is
required. This necessitates a pre−amplifier stage with gain and a low pass audio filter. An inexpensive input
stage can be designed using TI's LM833 audio operational amplifier and a minimum number of external
components. A gain of 10 (20dB) is recommended for the pre−amplifier stage. For a subwoofer application, the
pole of the low pass filter is normally set within the range of 60Hz − 180Hz. For a clean sounding subwoofer the
filter should be at least a second-order filter to sharply roll off the high frequency audio signals. A higher order
filter is recommended for stand-alone self-powered subwoofer applications. Figure 6 shows a simple input stage
with a gain of 10 and a second-order low pass filter.
Figure 24. Pre−amplifier Stage with Low Pass Filter
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Supply Bypassing
Correct supply bypassing has two important goals. The first is to ensure that noise on the supply lines does not
enter the circuit and become audible in the output. The second is to help stabilize an unregulated power supply
and provide current under heavy current conditions. Because of the two different goals multiple capacitors of
various types and values are recommended for supply bypassing. For noise de-coupling, generally small ceramic
capacitors (.001µF to .1µF) along with slightly larger tantalum or electrolytic capacitors (1µF to 10µF) in parallel
will do an adequate job of removing most noise from the supply rails. These capacitors should be placed as
close as possible to each IC's supply pin(s) using leads as short as possible. For supply stabilizing, large
electrolytic capacitors (3,300µF to 15,000µF) are needed. The value used is design and cost dependent.
High Frequency PCB Design
A double-sided PCB is recommended when designing a class D amplifier system. One side should contain a
ground plane with the power traces on the other side directly over the ground plane. The advantage is the
parasitic capacitance created between the ground plane and the power planes. This parasitic capacitance is very
small (pF) but is the value needed for coupling high frequency noise to ground. At high frequencies, capacitors
begin to act more like inductors because of lead and parasitic inductance in the capacitor. For this reason,
bypassing capacitors should be surface mount because of their low parasitic inductance. Equation 8 shows how
to determine the amount of power to ground plane capacitance.
C = εoεrA/d
(Farads)
where
•
here εo = 0.22479pF/in and εr = 4.1
(8)
A is the common PCB area and d is the distance between the planes. The designer should target a value of
100pF or greater for both the positive supply to ground capacitance and negative supply to ground capacitance.
Signal traces that cross over each other should be laid out at 90° to minimized any coupling.
Output Offset Voltage Minimization
The amount of DC offset voltage seen at the output with no input signal present is already quite good with the
LM4651/52. With no input signal present the system should be at 50% duty cycle. Any deviation from 50% duty
cycle creates a DC offset voltage seen by the load. To completely eliminate the DC offset, a DC voltage divider
can be used at the input to set the DC offset to near zero. This is accomplished by a simple resistor divider that
applies a small DC voltage to the input. This forces the duty cycle to 50% when there is no input signal. The
result is a LM4651 and LM4652 system with near zero DC offset. The divider should be a 1.8MΩ from the +6V
output (pin 6) to the input (other side of 25k, R1). R1 acts like the second resistor in the divider. Also use a 1µF
input capacitor before R1 to block the DC voltage from the source. R1 and the 1µF capacitor create a high pass
filter with a 3dB point at 6.35Hz. The value of ROFFSET is set according to the application. Variations in switching
frequency and supply voltage will change the amount of offset voltage requiring a different value than stated
above. The value above (1.8MΩ) is for ±20V and a switching frequency of 125kHz.
Output Stage Filtering
As common with Class D amplifier design, there are many trade-offs associated with different circuit values. The
output stage is not an exception. Texas Insturments has found good results with a 50µF inductor and a 5µF
Mylar capacitor (see Figure 3) used as the output LC filter. The two-pole filter contains three components; L1 and
CBYP because the LM4651 and LM4652 have a bridged output. The design formula for a bridge output filter is fC
= 1/{2π[L1(2CBYP + C1)]½} (Hz).
A common mistake is to connect a large capacitor between ground and each output. This applies only to singleended applications. In bridge operation, each output sees CBYP. This causes the extra factor of 2 in the formula.
The alternative to CBYP is a capacitor connected between each output, VO, and VO2, and ground. This alternative
is, however, not size or cost efficient because each capacitor must be twice CBYP's value to achieve the same
filter cutoff frequency. The additional small value capacitors connected between each output and ground (C1)
help filter the high frequency from the output to ground . The recommended value for C1 is 0.1µF to 1µF or 2% to
20% of CBYP."
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Modulation Frequency Optimization
Setting the modulation frequency depends largely on the application requirements. To maximize efficiency and
output power a lower modulation frequency should be used. The lower modulation frequency will lower the
amount of loss caused by switching the output MOSFETs increasing the efficiency a few percent. A lower
switching frequency will also increase the peak output power before clipping because the over modulation
protection time is a smaller percentage of the total period. Unfortunately, the lower modulation frequency has
worse THD+N performance when the output power is below 10 watts. The recommended switching frequency to
balance the THD+N performance, efficiency and output power is 125kHz to 145kHz.
THD+N Measurements and Out of Audio Band Noise
THD+N (Total Harmonic Distortion plus Noise) is a very important parameter by which all audio amplifiers are
measured. Often it is shown as a graph where either the output power or frequency is changed over the
operating range. A very important variable in the measurement of THD+N is the bandwidth limiting filter at the
input of the test equipment.
Class D amplifiers, by design, switch their output power devices at a much higher frequency than the accepted
audio range (20Hz - 20kHz). Switching the outputs makes the amplifier much more efficient than a traditional
Class A/B amplifier. Switching the outputs at high frequency also increases the out-of-band noise. Under normal
circumstances this out-of-band noise is significantly reduced by the output low pass filter. If the low pass filter is
not optimized for a given switching frequency, there can be significant increase in out-of-band noise.
THD+N measurements can be significantly affected by out-of-band noise, resulting in a higher than expected
THD+N measurement. To achieve a more accurate measurement of THD, the bandwidth at the input of the test
equipment must be limited. Some common upper filter points are 22kHz, 30kHz, and 80kHz. The input filter limits
the noise component of the THD+N measurement to a smaller bandwidth resulting in a more real-world THD+N
value.
The output low pass filter does not remove all of the switching fundamental and harmonics. If the switching
frequency fundamental is in the measurement range of the test equipment, the THD+N measurement will include
switching frequency energy not removed by the output filter. Whereas the switching frequency energy is not
audible, it's presence degrades the THD+N measurement. Reducing the bandwidth to 30kHz and 22kHz reveals
the true THD performance of the Class D amplifier. Increasing the switching frequency or reducing the cutoff
frequency of the output filter will also reduce the level of the switching frequency fundamental and it's harmonics
present at the output. This is caused by a switching frequency that is higher than the output filter cutoff frequency
and, therefore, more attenuation of the switching frequency.
In-band noise is higher in switching amplifiers than in linear amplifiers because of increased noise from the
switching waveform. The majority of noise is out of band (as discussed above), but there is also an increase of
audible noise. The output filter design (order and location of poles) has a large effect on the audible noise level.
Power supply voltage also has an effect on noise level. The output filter removes a certain amount of the
switching noise. As the supply increases, the attenuation by the output fiter is constant. However, the switching
waveform is now much larger resulting in higher noise levels.
THERMAL CONSIDERATIONS
Heat Sinking
The choice of a heat sink for the output FETs in a Class D audio amplifier is made such that the die temperature
does not exceed TJMAX and activate the thermal protection circuitry under normal operating conditions. The heat
sink should be chosen to dissipate the maximum IC power which occurs at maximum output power for a given
load. Knowing the maximum output power, the ambient temperature surrounding the device, the load and the
switching frequency, the maximum power dissipation can be calculated. The additional parameters needed are
the maximum junction temperature and the thermal resistance of the IC package (θJC, junction to case), both of
which are provided in the Absolute Maximum Ratings and Operating Ratings sections above.
It should be noted that the idea behind dissipating the power within the IC is to provide the device with a low
resistance to convection heat transfer such as a heat sink. Convection cooling heat sinks are available
commercially and their manufacturers should be consulted for ratings. It is always safer to be conservative in
thermal design.
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Proper IC mounting is required to minimize the thermal drop between the package and the heat sink. The heat
sink must also have enough metal under the package to conduct heat from the center of the package bottom to
the fins without excessive temperature drop. A thermal grease such as Wakefield type 120 or Thermalloy
Thermacote should be used when mounting the package to the heat sink. Without some thermal grease, the
thermal resistance θCS (case to sink) will be no better than 0.5°C/W, and probably much worse. With the thermal
grease, the thermal resistance will be 0.2°C/W or less. It is important to properly torque the mounting screw.
Over tightening the mounting screw will cause the package to warp and reduce the contact area with the heat
sink. It can also crack the die and cause failure of the IC. The recommended maximum torque applied to the
mounting screw is 40 inch-lbs. or 3.3 foot-lbs.
Determining Maximum Power Dissipation
Power dissipation within the integrated circuit package is a very important parameter. An incorrect maximum
power dissipation (PD) calculation may result in inadequate heat sinking, causing thermal shutdown circuitry to
operate intermittently. There are two components of power dissipation in a class D amplifier. One component of
power dissipation in the LM4652 is the RDS(ON) of the FET times the RMS output current when operating at
maximum output power. The other component of power dissipation in the LM4652 is the switching loss. If the
output power is high enough and the DC resistance of the filter coils is not minimized then significant loss can
occur in the output filter. This will not affect the power dissipation in the LM4652 but should be checked to be
sure that the filter coils with not over heat.
The first step in determining the maximum power dissipation is finding the maximum output power with a given
voltage and load. Refer to the graph Output Power verses Supply Voltage to determine the output power for
the given load and supply voltage. From this power, the RMS output current can be calculated as IOUTRMS =
SQRT(POUT/RL). The power dissipation caused by the output current is PDOUT = (IOUTRMS)2 * (2 * RDS(ON)). The
value for RDS(ON) can be found from the Electrical Characteristics for LM4652 table above. The percentage of loss
due to the switching is calculated by Equation 9:
%LOSSSWITCH = (tr+ tf + TOVERMOD) * fSW
(9)
tr, tf and TOVERMOD can be found in the Electrical Characteristics for LM4651 and Electrical Characteristics for
LM4652 sections above. The system designer determines the value for fSW (switching frequency). Power
dissipation caused by switching loss is found by Equation 10. POUTMAX is the 1% output power for the given
supply voltage and the load impedance being used in the application. POUTMAX can be determined from the graph
Output Power vs. Supply Voltage in the Typical Performance Characteristics section above.
PDSWITCH = (%LOSSSWITCH * POUTMAX) / (1−%LOSSSWITCH)
(Watts)
(10)
PDMAX for the LM4652 is found by adding the two components (PDSWITCH + PDOUT) of power dissipation together.
Determining the Correct Heat Sink
Once the LM4652's power dissipation known, the maximum thermal resistance (in °C/W) of a heat sink can be
calculated. This calculation is made using Equation 11 and is based on the fact that thermal heat flow
parameters are analogous to electrical current flow properties.
PDMAX = (TJMAX − TAMBIENTMAX) / θJA (Watts)
where
•
θJA = θJC + θCS + θSA
(11)
Since we know θJC, θCS, and TJMAX from the Absolute Maximum Ratings and Operating Ratings sections above
(taking care to use the correct θJC for the LM4652 depending on which package type is being used in the
application) and have calculated PDMAX and TAMBIENTMAX, we only need θSA, the heat sink's thermal resistance.
The following equation is derived from Equation 11:
θSA = [(TJMAX − TAMBIENTMAX) / PDMAX] − θJC − θCS
(12)
Again, it must be noted that the value of θSA is dependent upon the system designer's application and its
corresponding parameters as described previously. If the ambient temperature surrounding the audio amplifier is
higher than TAMBIENTMAX, then the thermal resistance for the heat sink, given all other parameters are equal, will
need to be lower.
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Example Design of a Class D Amplifier
The following is an example of how to design a class D amplifier system for a power subwoofer application
utilizing the LM4651 and LM4652 to meet the design requirements listed below:
• Output Power, 1% THD 125W
• Load Impedance 4Ω
• Input Signal level 3V RMS (max)
• Input Signal Bandwidth 10Hz − 150Hz
• Ambient Temperature 50°C (max)
Determine the Supply Voltage
From the graph Output Power verses Supply voltage at 1% THD the supply voltage needed for a 125 watt, 4Ω
application is found to be ±20V.
Determine the Value for ROSC(Modulation Frequency)
The oscillation frequency is chosen to obtain a satisfactory efficiency level while also maintaining a reasonable
THD performance. The modulation frequency can be chosen using the Clipping Power Point and Efficiency
verses Switching Frequency graph. A modulation frequency of 125kHz is found to be a good middle ground for
THD performance and efficiency. The value of the resistor for ROSC is found from Equation 6 to be 3.9 kΩ.
Determine the Value for RSCKT (Circuit Limit)
The current limit is internally set as a failsafe to 10 amps. The inductor ripple current and the peak output current
must be lower than 10 amps or current limit protection will turn on. A typical 4Ω load driven by a filter using 50µH
inductors does not require more than 10A. The current limit will have to be increased when loads less than 4Ω
are used to acheive higher output power. With RSCKT equal to 100kΩ, the current limit is 10A.
Determine the Value for RDLY (Dead Time Control)
The delay time or dead time is set to the recommended value so RDLY equals 5kΩ. If a higher bandwidth of
operation is desired, RDLY should be a lower value resistor. If a zero value for RDLY is desired, connect the
LM4651's pin 17 to GND.
Determine the Value of L1, CBYP, C1, Rfl1 Rfl2, Cfl1 Cfl2, Rf, Cf (the Output and Feedback Filters)
All component values show in Figure 3, are optimized for a subwoofer application. Use the following guidelines
when changing any component values from those shown. The frequency response of the output filter is
controlled by L1 and CBYP. Refer to the Application Information section titled Output Stage Filtering for a detailed
explanation on calculating the correct values for L1 and CBYP.
C1 should be in the range of 0.1µF to 1µF or 2 - 20% of CBYP.
Rfl1 and Rfl2 are found by the ratio Rfl1 = 10Rfl2.
A lower ratio can be used if the application is for lower output voltages than the 125Watt, 4Ω solution show here.
The feedback RC filter's pole location should be higher than the output filter pole. The reason for two capacitors
in parallel instead of one larger capacitor is to reduce the possible EMI from the feedback traces. Cfl1 is placed
close as possible to the output of the LM4652 so that an audio signal is present on the feedback trace instead of
a high frequency square wave. Cfl2 is then placed as close as possible to the feedback inputs (pins 14, 19) of the
LM4651 to filter off any noise picked up by the feedback traces. The combination lowers EMI and provides a
cleaner audio feedback signal to the LM4651. Rf should be in range of 100kΩ to1MΩ. Cf controls the bandwidth
of the error signal and should be in the range of 100pF to 470pF.
Determine the Value for CSTART (Start Up Delay)
The start-up delay is chosen to be 1 second to ensure minimum pops or clicks when the amplifier is powered up.
Using Equation 2, the value of CSTART is 11.7µF. A standard value of 10µF is used.
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Determine the Value of Gain, R1, and R2
The gain is set to produce a 125W output at no more than 1% distortion with a 3VRMS input. A dissipation of
125W in a 4Ω load requires a 22.4VRMS signal. To produce this output signal, the LM4651/LM4652 amplifier
needs an overall closed-loop gain of 22.4VRMS/3VRMS, or 7.5V/V (17.5db). Equation 13 shows all the variables
that affect the system gain.
Gain = {[R2/(R1 + 100)] x [(Rfl1 + Rfl2)/Rfl2] − [R2/(R1 + 100)] + 0.5} + [(VCC - 20) * 0.0175] (V/V)
(13)
The values for RfI1, RfI2, and Rf were found in the Determine the Value of the Filters section above and shown
in Figure 1. Therefore, RfI1 = 620kΩ, RfI2 = 62kΩ and Rf = 390kΩ. The value of VCC was also found as the first
step in this example to be ±20V. Inserting these values into Equation 13 and reducing gives the equation below:
R2 = 0.7(R1 + 100)
(14)
The input resistance is desired to be 20kΩ so R1 is set to 20kΩ. R2 is then found to require a value of 14.1kΩ.
Standard resistor values are 14.0kΩ giving a gain of 7.43V/V or 14.3kΩ giving a gain of 7.58V/V.
Lowering R2 direcly affects the noise of the system. Changing R1 to increase gain with the lower value for R2 has
very little affect on the noise level. The percent change in noise is about what whould be expected with a higher
gain. The drawback to a lower R1 value is a larger CIN value, necessary to properly couple the lowest desired
signal frequencies. If a 20kΩ input impedance is not required, then the recommended values shown in Figure 3
should be used: with R1's value set to 4.7kΩ and then using a value of 3.4kΩ for R2 for a gain of 7.5V/V.
Determine the Needed Heat Sink
The only remaining design requirement is a thermal design that prevents activating the thermal protection
circuitry. Use Equation 9, Equation 10, and Equation 11 to calculate the amount of power dissipation for the
LM4652. The appropriate heat sink size, or thermal resistance in °C/W, will then be determined.
Equation 9 determines the percentage of loss caused by the switching. Use the typical values given in the
Electrical Characteristics for LM4651 and Electrical Characteristics for LM4652 tables for the rise time, fall time
and over modulation time:
%Loss = (25ns+26ns+350ns) * 125kHz
where
•
%Loss = 5.0%
(15)
This switching loss causes a maximum power dissipation, using Equation 10, of:
PDSWITCH = (5.0% * 125W) / (1−5.0%)
where
•
PDSWITCH = 6.6W
(16)
Next the power dissipation caused by the RDS(ON) of the output FETs is found by multiplying the output current
times the RDS(ON). Again, the value for RDS(ON) is found from the Electrical Characteristics for LM4652 table
above. The value for RDS(ON) at 100°C is used since we are calculating the maximum power dissipation.
IOUTRMS = SQRT(125watts/4Ω) = 5.59 amps
where
•
•
PRDS(ON) = (5.59A)2 * (0.230Ω*2)
PRDS(ON) = 14.4W
(17)
The total power dissipation in the LM4652 is the sum of these two power losses giving:
PDTOTAL = 6.6W + 14.4W = 21W
(18)
The value for Maximum Power Dissipation given in the System Electrical Characteristics for LM4651 and
LM4652 is 22 watts. The difference is due to approximately 1 watt of power loss in the LM4651. The above
calculations are for the power loss in the LM4652.
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Lastly, use Equation 11 to determine the thermal resistance of the LM4652's heat sink. The values for θJC and
TJMAX are found in the Operating Ratings and the Absolute Maximum Ratings section above for the LM4652. The
value of θJC is 2°C/W for the isolated (NDB) package or 1°C/W for the non-isolated (NDL) package. The value for
TJMAX is 150°C. The value for θCS is set to 0.2°C/W since this is a reasonable value when thermal grease is
used. The maximum ambient temperature from the design requirements is 50°. The value of θSA for the isolated
(NDB) package is:
θSA = [(150°C − 50°C)/21W] − 2°C/W − 0.2°C/W
where
•
θSA = 2.5°C/W
(19)
and for the non-isolated (NDL) package without a mica washer to isolate the heat sink from the package:
θSA = [(150°C − 50°C)/21W] − 1°C/W − 0.2°C/W
where
•
θSA = 3.5°C/W
(20)
To account for the use of a mica washer simply subtract the thermal resistance of the mica washer from θSA
calculated above.
Table 1. RECOMMENDATIONS FOR CRITICAL EXTERNAL COMPONENTS
Circuit Symbol
Suggested
Value
Suggested Type
CfI1
330pF
Ceramic Disc
CfI2
100pF
Ceramic Disc
Cf
470pF
Ceramic Disc
Supplier/Contact Information
Supplier Part #
CB2
1.0µF - 10µF
Resin Dipped Solid Tantalum
CB1 & CBT
0.1µF
Monolithic Ceramic
CB3
0.001µF - 0.1µF
Monolithic Ceramic
C2
0.1µF - 1.0µF
Metallized Polypropylene or Polyester
Film
CBYP
1.0µF - 10µF
Metallized Polypropylene or Polyester
Film
Bishop Electronics Corp.
(562) 695 - 0446
http://www.bishopelectronics.com/
BEC-9950
A11A-50V
CBYP
1.0µF - 10µF
Metallized Polypropylene or Polyester
Film
Nichicon Corp.
(847) 843-7500
http://www.nichicon-us.com/
QAF2Exx
or
QAS2Exx
D1
1A, 50V
Fast Schottky Diode
L1
25µH, 5A
High Current Toroid Inductor
(with header)
J.W. Miller
(310) 515-1720
http://www.jwmiller.com/
6702
L1
47µH, 5A
High Saturation Open Core
(Vertical Mount Power Chokes)
CoilCraft
(847) 639-6400
http://www.coilcraft.com/
PCV-0473-05
L1
50µH, 5.6A
High Saturation Flux Density
Ferrite Rod
J.W. Miller
(310) 515-1720
http://www.jwmiller.com/
5504
L1
68µH, 7.3A
High Saturation Flux Density
Ferrite Rod
J.W. Miller
(310) 515-1720
http://www.jwmiller.com/
5512
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Figure 25. Reference PCB Schematic
Figure 26. Reference PCB Silk Screen Layer
22
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Figure 27. Reference PCB Silk Screen and Solder Mask Layers
Figure 28. Reference PCB Top Layer
Figure 29. Reference PCB Bottom Layer
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Table 2.
BILL OF MATERIALS FOR REFERENCE PCB
Symbol
Value
Tolerance
Type
# per
Board
RFL1
620kΩ
1%
1/8 - 1/4 watt
2
RFL2
62kΩ
1%
1/8 - 1/4 watt
2
RFL3
0Ω
1%
1/8 - 1/4 watt
2
RF
1MΩ
1%
1/8 - 1/4 watt
1
R1
4.7kΩ
1%
1/8 - 1/4 watt
1
R2
4.7kΩ
1%
1/8 - 1/4 watt
1
RLP
2.2kΩ
1%
1/8 - 1/4 watt
1
ROFFSET
0
1%
1/8 - 1/4 watt
0
RDLY
5.1kΩ
10%
1/8 - 1/4 watt
1
RSCKT
39kΩ
10%
1/8 - 1/4 watt
1
ROSC
6.8kΩ
10%
1/8 - 1/4 watt
1
Supplier/Comment
Part #
Shorting Jumper
** NOT USED **
Can also use as a 5.6kΩ resistor
All Caps. are Radial lead except CBYP, C1.
Symbol
Value
Tolerance
Type
Voltage
# per
Board
Supplier/Comment
Part #
CIN
1µF
10%
Metal Polyester
100V
1
Digi-Key (800) 344-4539
EF1105-ND
CLP
0.47µF
10%
Metal Polyester
25V
1
Digi-Key (800) 344-4539
EF1474-ND
CF
470pF
5%
Ceramic Disc
25V
1
Digi-Key (800) 344-4539
1321PH-ND
CFL1
0
5%
Ceramic Disc
25V
0
** NOT USED **
1319PH-ND
CFL2
100pF
5%
Ceramic Disc
25V
2
Digi-Key (800) 344-4539
1313PH-ND
CBT
0.1µF
10% - 20%
Monolithic
Ceramic
100V
2
Digi-Key (800) 344-4539
P4924-ND
CB1
0.1µF
10% - 20%
Monolithic
Ceramic
100V
6
Digi-Key (800) 344-4539
P4924-ND
CB2
1µF
10%
Tantalum Radial
lead
35V
6
Digi-Key (800) 344-4539
P2059-ND
CB3
0.001µF
10% - 20%
Monolithic
Ceramic
100V
3
Digi-Key (800) 344-4539
P4898-ND
CB4
47µF
10% - 20%
Electrolytic Radial
16V
1
Digi-Key (800) 344-4539
P914-ND
1.5µF
10%
Tantalum Radial
lead
25V
1
Digi-Key (800) 344-4539
P2044-ND
C1
0
10%
Metal Polyester
25V
0
** NOT USED **
C2
1µF
10%
Metal Polyester
25V
2
Digi-Key (800) 344-4539
EF1105-ND
CBYP
4.7µF
10% - 20%
Metal Polyester
50V
1
Digi-Key (800) 344-4539
EF1475-ND
CSBY1
4,700µF
20%
Electrolytic Radial
25V
2
Digi-Key (800) 344-4539
P5637A-ND
CSBY2
0.1µF
20%
Ceramic Disc
25V
2
Digi-Key (800) 344-4539
P4201-ND
CSBY3
0
10% - 20%
Mylar Axial lead
50V
0
** NOT USED **
CStart
One or more pairs of coils from the list below is included with the reference PCB.
Symbol
Value
Tolerance
Type
Voltage
# per
Board
Supplier/Comment
Part #
L1
25µH
15%
High Current
Toroid with
Header
5.5 amp
2
J.W. Miller (310) 515-1720
6702
L1
47µH
10%
Ferrite Bobbin
Core
5.0 amp
2
CoilCraft (847) 639-6400
http://www.coilcraft.com
PVC-2-473-05
L1
50µH
10%
Ferrite Core
5.6 amp
2
J.W. Miller (310) 515-1720
5504
# per
Board
Supplier/Comment
Part #
Symbol
S1
Standoffs
24
Description
(SPDT) on-on, switch for STBY
1
Mouser (800) 346-6873
1055-TA2130
Plastic Round, 0.875", 4-40
4
Newark (800) 463-9275
92N4905
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Table 2.
BILL OF MATERIALS FOR REFERENCE PCB (continued)
RCA Input
PCB Mount
Banana Jack Banana jack BLACK
Heat sink
D1
1
Mouser (800) 346-6873
16PJ097
5
Mouser (800) 346-6873
164-6218
Wakefield 603K, 2” high X 2” wide, ~ 7°C/W
1
Newark (800) 463-9275
58F537 (603K)
1A, 50Volt Schottky (40A surge current, 8.3mS)
4
Digi-Key (800) 344-4539
SR105CT-ND
Additional Formulas for Reference PCB:
Pole due to CIN:
f3dB = 1/[2π(R1 + RLP)CIN] or CIN = 1/[2π(R1 + RLP)f3dB]
Pole due to RLP and CLP:
f3dB = 1/[2π(R1 // RLP)CLP] or CLP = 1/[2π(R1 // RLP)f3dB]
where:
(R1 // RLP) = 1/[1/R1 + 1/RLP]
Gain for Reference PCB:
Gain = {[R2/(R1 + 100 + Rlp)] x [(Rfl1 + Rfl2)/Rfl2] − [R2/(R1 + 100 + Rlp)] + 0.5} + [(VCC - 20) * 0.0175]
FULL AUDIO BANDWIDTH OPERATION
There is nothing in the design of the LM4651/52 class D chipset that prevents full audio bandwidth (20 – 20kHz)
operation. For full bandwidth operation there are several external circuit changes required. Additional external
circuitry is helpful to achieve a complete solution with the best performance possible with the LM4651/52 class D
chipset. The additional sections and figures below detail the changes needed for either a 60W / 8Ω or 100W / 4Ω
(10% THD+N) complete solution using a +/-17V supply.
FILTERS
To achieve full bandwidth operation there are several filter points that must be modified. They are the output
filter, the feedback filters, the error amplifier filter and the input filter. If any of the filter points are too low there will
be large phase shifts in the upper audio frequencies reducing the resolution and clarity of the highs. For this
reason the frequency response of the system should be flat out to 20kHz. The mistake is often made to set the
–3dB point near 20kHz resulting in good bench performance but poor quality in listening test.
The output filter is made up of L1, L2, CBYP, CF1, CF2 (see Figure 31). The output filter design is determined by the
load impedance along with the frequency response. The filter must have a 3dB point beyond 20kHz and a Q
factor close to 0.707 for best performance. The output filter is the only filter that changes with the load
impedance (See Table 2 for values). Standard inductor values were used for both 4Ω and 8Ω filters.
The feedback filters and error amplifier filters will interact with the output filter if the individual pole locations of
each are too close together. The feedback filter point is moved by reducing the value of CFL1, CFL3 to 50pF
putting the feedback filter points approximately 5kHz higher than the output filter point. The error amplifier filter
point is determined by Equation 7. Reducing the value of CF to 390pF gave the best results.
The input filter in the typical application is a simple passive, single pole RC filter. For improved performance an
active two pole filter was added as discussed below.
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PRE-AMPLIFIER AND INPUT FILTER
For a complete solution and best performance a pre-amplifier is required. With the addition of a pre-amplifier the
gain of the class D stage can be greatly reduced to improve performance. The pre-amplifier gain is set to 10V/V
allowing for low gain on the class D stage with total system gain high enough to be a complete solution from line
level (1VRMS) sources. Without the pre-amplifier stage the class D stage must have much higher gain and will
result in decreased performance in the form of much higher THD.
With an extra op. amp. available on the other side of the LM833N the passive RC input filter is changed to an
active two pole filter. The input filter does not noticeable increase THD performance but will help maintain a flat
frequency response as the Q of the output filter changes with load impedance. A real speaker load impedance
varies with frequency changing the Q of the output filter. The input filter is recommended to maintain flat
response. For the pre-amplifier and input filter stage the circuit in Figure 6 was used with the complete input
stage shown in Figure 12.
SWITCHING FREQUENCY
A switching frequency from 75kHz to 125kHz is adequate for subwoofer applications. A lower switching
frequency has higher efficiency and higher output power at the start of clipping. For a full audio bandwidth
application a higher switching frequency is needed. The switching frequency must be increased not only for
waveform resolution for the higher audio frequencies but also to decrease the noise floor. A switching frequency
of 175kHz was used for the performance graphs shown below. The Audio Precision AUX-0025 Switching
Amplifier Measurement Filter was placed before the input to the Audio Precision unit for the THD+N graphs
below.
TYPICAL PERFORMANCE FOR FULL RANGE APPLICATION
26
Frequency Response
±17V, fSW = 175kHz, POUT = 5W = 0dB
RL = 8Ω, No Filters
THD+N vs Frequency
±17V, fSW = 175kHz, POUT = 1W & 25W
RL = 8Ω, 30kHz BW
THD+N vs Output Power
±17V, fSW = 175kHz
RL = 8Ω, 30kHz BW
Frequency Response
±17V, fSW = 175kHz, POUT = 5W = 0dB
RL = 4Ω, No Filters
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THD+N vs Frequency
±17V, fSW = 175kHz, POUT = 1W & 50W
RL = 4Ω, 30kHz BW
THD+N vs Output Power
±17V, fSW = 175kHz
RL = 4Ω, 30kHz BW
Figure 30. Input Pre-Amplifier And Filter Schematic
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Figure 31. Full Audio Bandwidth Schematic
28
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FULL AUDIO BANDWIDTH REFERENCE BOARD ARTWORK
Figure 32. Composite Top View
Figure 33. Composite Bottom View
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Figure 34. Silk Screen Layer
Figure 35. Top Layer
30
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Figure 36. Bottom Layer
Table 3. BILL OF MATERIALS FOR FULL AUDIO BANDWIDTH REFERENCE PCB
Symbol
Value
Tolerance
Type
RFL1, RFL2
620kΩ
1%
1/8 – 1/4 Watt
RFL3, RFL4
62kΩ
1%
1/8 – 1/4 Watt
RF
1MΩ
1%
1/8 – 1/4 Watt
R1
10kΩ
1%
1/8 – 1/4 Watt
R2
3.3kΩ
1%
1/8 – 1/4 Watt
Supplier/ Comment
ROFFSET
Part #
** NOT USED **
RDLY
5.1kΩ
5%
RSCKT
39kΩ
5%
1/8 – 1/4 Watt
1/8 – 1/4 Watt
ROSC
20kΩ
20%
Trim Potentiometer
RG1, RG2, RG3,
RG4
3.3Ω
5%
1/8 – 1/4 Watt
RTSD
100kΩ
5%
1/8 – 1/4 Watt
RPA
10kΩ
1%
1/8 – 1/4 Watt
Ri
1kΩ
1%
1/8 – 1/4 Watt
RLP1, RLP2
2.7kΩ
1%
1/8 – 1/4 Watt
RIN
47kΩ
5%
1/8 – 1/4 Watt
RV1, RV2
750kΩ
5%
Mouser (800) 346–6873
323–409H-20K
1/4 Watt
All Capacitors are
Radial lead
Symbol
Value
Tolerance
Type
Voltage
Supplier/Comment
Part #
CIN
1µF
10%
Metal Polyester
100V
Digi-Key (800) 344–4539
EF1105–ND
CLP1
0.0022µF
10%
Ceramic Disc
25V
Digi-Key (800) 344–4539
P4053A-ND
CLP2
0.001µF
10%
Ceramic Disc
25V
Digi-Key (800) 344–4539
P4049A-ND
CBT1, CBT2
0.1µF
20%
Monolithic Ceramic
100V
Digi-Key (800) 344–4539
P4924–ND
CF
390pF
10%
Metal Polyester
25V
Digi-Key (800) 344–4539
P4932–ND
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Table 3. BILL OF MATERIALS FOR FULL AUDIO BANDWIDTH REFERENCE PCB (continued)
CFL1, CFL3
47pF
10%
Metal Polyester
50V
CFL2
P4845–ND
** NOT USED **
CSTART
1.5μF
10%
Tantalum Radial lead
25V
Digi-Key (800) 344–4539
P2044–ND
CB1, CB2
0.001μF
20%
Monoilthic Ceramic
100V
Digi-Key (800) 344–4539
P4898–ND
CB3 – CB12
0.1μF
20%
Monolithic Ceramic
100V
Digi-Key (800) 344–4539
P4924–ND
CS1 – CS5
1μF
10%
Tantalum Radial lead
35V
Digi-Key (800) 344–4539
P2059–ND
CVD1
0.001μF
20%
Monolithic Ceramic
100V
Digi-Key (800) 344–4539
P4898–ND
CVD2
47μF
20%
Electrolytic Radial
16V
Digi-Key (800) 344–4539
P914–ND
CF1, CF2
0.1μF
10%
Metal Polyester
25V
Digi-Key (800) 344–4539
EF1104–ND
CBYP (4Ω)
0.47μF
10%
Metal Polyester
50V
Digi-Key (800) 344–4539
EF1474–ND
CBYP (8Ω)
0.22μF
10%
Metal Polyester
50V
Digi-Key (800) 344–4539
EF1224–ND
CSBY1 , CSBY2
4,700μF
20%
Electrolytic Radial
25V
Digi-Key (800) 344–4539
P10289-ND
CSBY3 , CSBY4
1,000μF
20%
Electrolytic Radial
25V
Digi-Key (800) 344–4539
P10279–ND
CSBY5 , CSBY6
0.1μF
Ceramic Disc
25V
Digi-Key (800) 344–4539
P4201–ND
CSBY7, CSBY8
47μF
20%
Electrolytic Radial
16V
Digi-Key (800) 344–4539
P914–ND
Symbol
Value
Tolerance
Type
Rating
Supplier/Comment
Part #
PVC–2–103–05
20%
L1, L2 (4Ω)
10μH
10%
Ferrite Bobbin Core
5.0 amp
CoilCraft (847) 639–6400
http://www.coilcraft.com
L1, L2 (8Ω)
22μH
10%
Ferrite Bobbin Core
5.0 amp
CoilCraft (847) 639–6400
http://www.coilcraft.com
PVC–2–223–05
Symbol
Description
Supplier/Comment
Part #
S1
(SPDT) on-on,
switch for STBY
Mouser (800) 346–6873
1055–TA2130
D1 – D4
1A, 50V Schottky
(40A surge
current, 8.3ms)
Digi-Key (800) 344–4539
SR105CT-ND
ZDV1, ZDV2
12V, 500mW
Zener diode
Digi-Key (800) 344–4539
1N5242
Standoffs
J1, J2, J3, J4
32
Digi-Key (800) 344–4539
Plastic Round,
0.875”, 4–40
Newark (800) 463–9275
92N4905
Banana jack RED
Mouser (800) 346–6873
164–6219
J5
Banana jack
BLACK
Mouser (800) 346–6873
164–6218
J6
RCA jack, PCB
mount
Mouser (800) 346–6873
16PJ097
U1
Dual audio Op.
Amp.
Texas Instruments
LM833N
U2
Integrated Class D
controller and
amplifier
Texas Instruments
LM4651N
U3
H-Bridge Power
MOSFET
Texas Instruments
LM4652
Heat sink
Wakefield 603K, 2”
high x 2” wide,
∼7°C/W
Newark (800) 463–9275
58F537 (603K)
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REVISION HISTORY
Changes from Revision G (April 2013) to Revision H
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 31
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