Maxim MAX747EPD High-efficiency pwm, step-down p-channel dc-dc controller Datasheet

19-0171; Rev 1; 9/93
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
________________________Applications
Notebook Power Supplies
Personal Digital Assistants
Battery-Operated Equipment
Cellular Phones
5V to 3.3V Green PC Applications
____________________________Features
♦ 90% to 95% Efficiency for 50mA to 2.5A
Output Currents
♦ 4V to 15V Input Voltage Range
♦ Low 800µA Supply Current
♦ 0.6µA Shutdown Current
♦ Drives External P-Channel FETs
♦ Cycle-by-Cycle Current Limiting
♦ 2V ±1.5% Accurate Reference Output
♦ Adjustable Soft-Start
♦ Precision Comparator for Power-Fail or
Low-Battery Warning
______________Ordering Information
PART
TEMP. RANGE
PIN-PACKAGE
MAX747CPD
0°C to +70°C
14 Plastic DIP
MAX747CSD
MAX747C/D
MAX747EPD
MAX747ESD
MAX747MJD
0°C to +70°C
0°C to +70°C
-40°C to +85°C
-40°C to +85°C
-55°C to +125°C
14 Narrow SO
Dice*
14 Plastic DIP
14 Narrow SO
14 CERIDIP
* Contact factory for dice specifications.
__________Typical Operating Circuit
INPUT
6V TO 15V
V+
100µF
AV+
5OmΩ
MAX747
__________________Pin Configuration
CS
ON/OFF
SHDN
EXT
P
5OµH
OUTPUT
5V
2.3A
TOP VIEW
430µF
LBI
14 LBO
1
SS
2
13 GND
MAX747
REF
3
SHDN
4
11 EXT
FB
5
10 AGND
CC
6
9
CS
AV+ 7
8
OUT
12 V+
OUT
LOW-BATTERY
DETECTOR
INPUT
LBI
REF
SS
0.1µF
LBO
CC FB AGND GND
LOW-BATTERY
DETECTOR
OUTPUT
DIP/SO
™ Dual-Mode and Idle-Mode are trademarks of Maxim Integrated Products.
________________________________________________________________ Maxim Integrated Products
Call toll free 1-800-998-8800 for free samples or literature.
1
MAX747
_______________General Description
The MAX747 high-efficiency, high-current, step-down
controller drives external P-channel FETs. It provides
90% to 95% efficiency from a 6V supply with load
currents ranging from 50mA up to 2.5A. It uses a
pulse-width-modulating (PWM) current-mode control
scheme to provide precise output regulation and low
output noise. The MAX747’s 4V to 15V input voltage
range, a fixed 5V/adjustable (Dual-Mode™) output, and
a current limit set with an external resistor make this
device ideal for a wide range of applications.
High efficiency is maintained with light loads due to a
proprietary dual-control (Idle-Mode™) scheme that
minimizes switching losses by reducing the switching
frequency at light loads. The low 800µA quiescent
current and ultra-low 0.6µA shutdown current further
extend battery life.
External components are protected by the MAX747’s
cycle-by-cycle current limit. The MAX747 also features a
2V ±1.5% reference, a comparator for low-battery
detection or level translating, as well as soft-start and
shutdown capability.
The MAX746, discussed in a separate data sheet,
functions similarly to the MAX747, but it drives N-channel
logic level FETs on the high side.
MAX747
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
ABSOLUTE MAXIMUM RATINGS
Supply Voltage V+, AV+ to GND ..............................-0.3V to 17V
AGND to GND..........................................................-0.3V to 0.3V
All Other Pins................................................-0.3V to (V+ + 0.3V)
Reference Current (IREF) ....................................................±2mA
Continuous Power Dissipation (TA = +70°C)
Plastic DIP (derate 10.00mW/°C above +70°C) ..........800mW
SO (derate 8.33mW/°C above +70°C) .........................667mW
CERDIP (derate 9.09mW/°C above +70°C) .................727mW
Operating Temperature Ranges:
MAX747C_D .......................................................0°C to +70°C
MAX747E_D.....................................................-40°C to +85°C
MAX747MJD ..................................................-55°C to +125°C
Junction Temperature
MAX747C_D/E_D .........................................................+150°C
MAX747MJD ...............................................................+175°C
Storage Temperature Range .............................-65°C to +160°C
Lead Temperature (soldering, 10sec) .............................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied.
Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V+ = 10V, ILOAD = 0mA, IREF = 0mA, TA = TMIN to TMAX, unless otherwise noted.)
PARAMETER
Input Voltage Range
Output Voltage
Feedback Voltage
Line Regulation
Load Regulation
Efficiency
OUT Leakage Current
FB Input Logic Low
FB Input Leakage Current
Reference Voltage
Reference Load Regulation
Soft-Start Source Current
Soft-Start Fault Current
Supply Current
Oscillator Frequency
Maximum Duty Cycle
CS Amp ILIM Threshold
EXT Output High
EXT Output Low
EXT Sink Current
EXT Source Current
CC Impedance
LBI Threshold Voltage
LBO Output Voltage Low
LBI Input Leakage Current
LBO Output Leakage Current
SHDN Input Voltage Low
SHDN Input Voltage High
SHDN Input Leakage Current
2
SYMBOL
CONDITIONS
V+
For regulated outputs
V+ = 6V to 15V, 0V < V+ - CS < 0.125V, FB = 0V
VOUT
(includes line and load regulation)
V+ - CS = 0V, external MAX747C
feedback mode
MAX747E/M
V+ = 6V to 15V, FB = 0V
V+ = 4V to 15V, external feedback mode
0V < V+ - CS < 0.125V
Circuit of Figure 1, ILOAD = 0.5A to 2.5A
VOUT = 5V
For dual-mode switchover
FB = 2V
MAX747C
VREF
IREF = 0µA
MAX747E/M
IREF = 0µA to 100µA
SS = 0V
SS = 2V
Operating, V+ = 15V
Operating, V+ = 10V
Shutdown mode
MAX747C
fOSC
MAX747E/M
V+ = 6V
VLIMIT
V+ - CS
IEXT = -1mA (sourcing)
IEXT = 1mA (sinking)
VEXT = 7.5V
VEXT = 2.5V
VTH
LBI falling
MAX747C
MAX747EM
MIN
4
TYP
MAX
15
UNITS
V
4.85
5.08
5.25
V
1.96
1.95
2.00
2.00
0.05
2.04
2.05
V
1.3
91
50
1.97
1.96
100
85
80
91
125
V+ – 0.1
0.1
2.00
2.00
9
1
500
0.95
0.8
0.6
100
100
96
150
80
40
100
2.03
2.04
20
1.3
20
115
120
175
0.25
1.97
1.96
110
170
24
2.00
2.00
ISINK = 0.5mA
LBI = 2.5V
V+ = 15V, LBO = 15V, LBI = 2.5V
VIL
VIH
0.1
2.5
2.03
2.04
0.4
100
1
0.4
2.0
SHDN = 10V
0.1
_______________________________________________________________________________________
100
%V
%
%
µA
mV
nA
V
mV
µA
µA
mA
µA
kHz
%
mV
V
V
mA
mA
kΩ
V
V
nA
µA
V
V
nA
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
SUPPLY CURRENT vs.
SUPPLY VOLTAGE
0.9
SUPPLY CURRENT (mA)
3
2
ENTIRE
CIRCUIT
1
0.8
0.7
SCHOTTKY DIODE
LEAKAGE EXCLUDED
0
-50 -25
0
25
50
75
5
100 125
7
9
11
13
2
VIN = 12V
1
15
VIN = 6V
VIN = 9V
0
0.01
0.6
1
0.1
10
TEMPERATURE (°C)
SUPPLY VOLTAGE (V)
OUTPUT CURRENT (A)
PEAK INDUCTOR CURRENT vs.
OUTPUT CURRENT (VOUT = 3.3V)
EFFICIENCY vs. OUTPUT CURRENT
(VOUT = 5V)
EFFICIENCY vs. OUTPUT CURRENT
(VOUT = 3.3V)
VIN = 6V
100
MAX747-TOC8
100
MAX747-TOC6
3
MAX1747-TOC7
-75
VIN = 6V
1
VIN = 6V
EFFICIENCY (%)
VIN = 9V
VIN = 9V
90
EFFICIENCY (%)
2
VIN = 12V
80
90
VIN = 9V
VIN = 12V
80
VIN = 5V
70
0.01
10
1
0.1
OUTPUT CURRENT (A)
1.2
PEAK
INDUCTOR
CURRENT
6
0.8
CONTINUOUS
CONDUCTION
REGION
2
0.4
0.4
0.6
0.8
1.0
OUTPUT CURRENT (A)
1.2
1.4
10
15
DISCONTINUOUS
CONDUCTION
REGION
PEAK INDUCTOR
CURRENT
VOUT = 5V
L = 50µH
RSENSE = 50mΩ
13
11
2.0
1.6
1.2
9
CONTINUOUS
CONDUCTION
REGION
7
0.8
PEAK INDUCTOR CURRENT (A)
1.6
DISCONTINUOUS
CONDUCTION
REGION
1
0.1
OUTPUT CURRENT (A)
CONTINUOUS-CONDUCTION MODE BOUNDARY
AND CORRESPONDING PEAK INDUCTOR CURRENT (VOUT = 5V)
2.0
PEAK INDUCTOR CURRENT (A)
VOUT = 3.3V
L = 33µH
RSENSE = 50mΩ
MAX747-TOC3
18
10
70
0.01
10
OUTPUT CURRENT (A)
CONTINUOUS-CONDUCTION MODE BOUNDARY
AND CORRESPONDING PEAK INDUCTOR CURRENT (VOUT = 3.3V)
14
1
MAX747-TOC4
0.1
SUPPLY VOLTAGE (V)
0
0.01
SUPPLY VOLTAGE (V)
PEAK INDUCTOR CURRENT (A)
3
MAX747-TOC5
MAX747-TOC2
VIN = 9V
VOUT = 5V
SUPPLY CURRENT (mA)
1.0
MAX747-TOC1
4
PEAK INDUCTOR CURRENT vs.
OUTPUT CURRENT (VOUT = 5V)
PEAK INDUCTOR CURRENT (A)
SUPPLY CURRENT vs.
TEMPERATURE
0.4
5
0.5
0.7
0.9
1.1
1.3
OUTPUT CURRENT (A)
_______________________________________________________________________________________
3
MAX747
__________________________________________Typical Operating Characteristics
(Circuit of Figure 1, V+ = 9V, TA = +25°C, unless otherwise noted.)
____________________________Typical Operating Characteristics (continued)
DISCONTINUOUS-CONDUCTION IDLE-MODE WAVEFORMS
MAX747-SCOPE2
MAX747-SCOPE1
CONTINUOUS-CONDUCTION MODE WAVEFORMS
a
a
b
b
c
c
20µs/div
5µs/div
V+ = 9V, IOUT = 125mA
a) EXT VOLTAGE, 10V/div
b) INDUCTOR CURRENT, 200mA/div
c) VOUT RIPPLE, 50mV/div
V+ = 9V, IOUT = 2.5A
a) EXT VOLTAGE, 10V/div
b) INDUCTOR CURRENT, 1A/div
c) VOUT RIPPLE, 50mV/div
LOAD-TRANSIENT RESPONSE
MAX747-SCOPE5
LINE-TRANSIENT RESPONSE
MAX747-SCOPE4
a
a
b
b
100µs/div
5ms/div
V+ = 9V, COUT = 430µF
a) LOAD CURRENT, 0.1A TO 2.5A, 1A/div
b) VOUT RIPPLE, 100mV/div
IOUT = 2.0A
a) V+ = 6V to 12V, 5V/div
b) VOUT RIPPLE, 100mV/div
MODERATE LOAD, IDLE-MODE WAVEFORMS
MAX747-SCOPE3
MAX747
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
a
b
c
5µs/div
V+ = 9V, IOUT = 560mA
a) EXT VOLTAGE, 5V/div
b) INDUCTOR CURRENT, 0.5A/div
c) VOUT RIPPLE 100mV/div
4
_______________________________________________________________________________________
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
PIN
NAME
FUNCTION
1
LBI
Input to the internal low-battery comparator. Tie to V+ or GND if not used.
2
SS
Soft-start limits start-up surge currents. On power-up, it charges the soft-start capacitor, slowly raising the
peak current limit to the level set by the sense resistor.
3
REF
4
SHDN
5
FB
Feedback input for adjustable-output operation. Connect to GND for fixed +5V output. Use a resistor
divider network to adjust the output voltage. See the section Setting the Output Voltage.
6
CC
Compensation capacitor. AC compensation input for the error amplifier. Connect a capacitor between CC
and GND for fixed +5V output operation. See Compensation Capacitor section.
7
AV+
Quiet supply voltage for sensitive analog circuitry. A bypass capacitor is not required for AV+.
8
OUT
Output voltage sense input. Connects to internal resistor divider. Leave unconnected for adjustable output.
Bypass to AGND with a 0.1µF capacitor close to the IC.
9
CS
Negative input to the current-sense amplifier. Connect the current-sense resistor (RSENSE) from V+ to CS.
10
AGND
11
EXT
Power MOSFET gate drive output that swings between V+ and GND. EXT is not protected against short
circuits to V+ or AGND.
12
V+
High-current supply voltage for the output driver
13
GND
High-current ground return for the output driver
14
LBO
Low-battery output is an open-drain output that goes low when LBI is less than 2V. Connect to V+ through
a pull-up resistor. Leave floating if not used. LBO is disabled in shutdown mode.
2V reference output that can source 100µA for external loads. Bypass with 0.22µF. The reference is
disabled in shutdown mode.
Active-high TTL/CMOS logic-level input. In shutdown mode, VOUT = 0V and the supply current is reduced
to 20µA.
Quiet analog ground
____________________Getting Starting
_______________Detailed Description
Figure 1a shows the 5V output 11.4W standard
application circuit and Figure 1b shows the 3.3V output
7.5W standard application circuit. Most applications
will be served by these circuits. To learn more about
component selection for particular applications, refer to
the Design Procedure section. To learn more about the
operation of the MAX747, refer to the Detailed
Description.
The MAX747 monolithic, CMOS, step-down switchmode power-supply controller drives external
P-channel FETs. It uses a unique current-mode pulsewidth-modulating (PWM) control scheme that results in
high efficiency over a wide range of load currents, tight
output voltage regulation, excellent load- and linetransient response, and low noise. Efficiency at light
loads is further enhanced by a proprietary Idle-Mode
switching control scheme that skips oscillator cycles in
order to reduce switching losses.
_______________________________________________________________________________________
5
MAX747
______________________________________________________________Pin Description
MAX747
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
VIN
(7.5V TO15V)
C2
100µF
VIN
(4.5V TO 15V)
C3
0.1µF
R2
R2
R3
100k
14
1
2
C4
0.1µF 3
C5
0.22µF 5
4
10
14
1
N.C.
LBI
RSENSE
50mΩ
R1
C6
470pF 6
MAX747
CC
CS
9
11
SS
EXT
REF
Q1
SI9405DY
P
L1
50µH
5V
C1 @ 2.3A
D1 430µF
NSQ03A03
FB
SHDN
AGND
C7
0.1µF
8
C1
430µF
Figure 1a. +5V Standard Application Circuit
Operating Principle
Figure 2 is the MAX747 block diagram. The MAX747
regulates using an inner current-feedback loop and an
outer voltage-feedback loop. The current loop is
stabilized by a slope compensation scheme and the
voltage loop is stabilized by the dominant pole formed
by the filter output capacitor and the load.
Discontinuous-/ContinuousConduction Modes
The MAX747 operates in continuous-conduction mode
(CCM) under heavy loads, but operates in
discontinuous-conduction mode (DCM) at light loads,
making it ideal for variable load applications. In DCM,
the inductor current starts and ends at zero on each
cycle. In CCM, the inductor current never returns to zero.
It is composed of a small AC component superimposed
on a DC level, which results in higher load-current
capability and lower output noise. Output noise is
reduced because the inductor does not exhibit the
ringing that occurs when the inductor current reaches
zero, and because there is a smaller AC component in
the inductor-current waveform (see inductor waveforms
in the Typical Operating Characteristics section). Note
12
V+ AV+ 7
LBO
LBI
OUT
RSENSE
50mΩ
R1
6
N.C.
C4
0.1µF 2
3
C5
0.22µF
4
10
OUT 8
GND
13
6
R3
100k
12
V+ AV+ 7
LBO
C3
0.1µF
C2
100µF
MAX747
CC
CS
9
11
SS
EXT
REF
Q1
SI9405DY
P
L1
33µH
3.3V
@ 2.3A
D1
NSQ03A03
SHDN
AGND
GND
13
FB
R5
13k
5
C6
2.7nF
R4
20k
C1
880µF
Figure 1b. +3.3V Standard Application Circuit
that to transfer equal amounts of energy to the load in
one cycle, the peak current level for the discontinuous
waveform must be much larger than the continuous
waveform peak current.
Slope Compensation
Stability of the inner current-feedback loop is provided
by a slope-compensation scheme that adds a ramp
signal to the current-sense amplifier output. Ideal slope
compensation can be achieved by adding a linear
ramp with the same slope as the declining inductor
current to the rising inductor current-sense voltage.
Therefore, the inductor must be scaled to the currentsense resistor value.
Overcompensation adds a pole to the outer voltagefeedback loop response that degrades loop stability.
This may cause voltage-mode pulse-frequencymodulation instead of PWM operation. Undercompensation results in inner current-feedback loop
instability, and may cause the inductor current to
staircase. Ideal matching between the sense resistor
and inductor is not required. The matching can be
±30% or more.
_______________________________________________________________________________________
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
EXT
MAX747
LBO
V+
LBI
LOW-BATTERY
COMPARATOR
+2V
REFERENCE
N
100kHz
OSCILLATOR
REF
OUT
ERROR
AMPLIFIER
60k
EXT
CONTROL
CC
40k
PWM
COMPARATOR
DUAL-MODE
COMPARATOR
FB
SHDN
100mV
AV+
CURRENT-SENSE
AMPLIFIER
CS
SLOPE
COMPENSATION
RAMP
VRAMP
IDLE-MODE
COMPARATOR
Σ
50mV
CURRENT-LIMIT
COMPARATOR
SS
SOFT-START
CIRCUITRY
AGND
GND
Figure 2. Block Diagram
The Oscillator and EXT Control
The switching frequency is nominally 100kHz and the
duty cycle varies from 5% to 96%, depending on the
input/output voltage ratio. EXT, which provides the gate
drive for the external P-FET, is switched between V+
and GND at the switching frequency. EXT is controlled
by a unique two-comparator control scheme composed
of a PWM comparator and an idle-mode comparator
(Figure 2). The PWM comparator determines the cycleby-cycle peak current with heavy loads, and the
light-load comparator sets the light-load peak current.
As VOUT begins to drop, EXT goes low and remains low
until both comparators trip. With heavy loads, the idlemode comparator trips quickly, and the PWM control
comparator determines the EXT on-time; with light
loads, the idle-mode comparator sets the EXT on-time.
_______________________________________________________________________________________
7
MAX747
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
PEAK CURRENT LIMIT (A)
MAX747-FIG3
3
RSENSE = 50mΩ
Figure 3 shows how the peak current limit increases as
the voltage on SS rises for two RSENSE values.
Shutdown Mode
When SHDN is high, the MAX747 enters shutdown
mode. In this mode, the internal biasing circuitry
(including EXT) is turned off, VOUT drops to 0V, and the
supply current drops to 0.6µA (20µA max). This
excludes external component leakage, which may add
several microamps to the shutdown supply current for
the entire circuit. SHDN is a TTL/CMOS logic-level
input. Connect SHDN to GND for normal operation.
2
V+ –VCS = 150mV
1
RSENSE = 100mΩ
Low-Battery Detector
0
0
1
2
3
4
SOFT-START VOLTAGE (V)
Figure 3. Peak Current Limit vs. Soft-Start Voltage
With decreasing loads, as the inductor current becomes
discontinuous, traditional PWM converters continue to
switch at a fixed frequency, decreasing light-load efficiency.
However, the MAX747’s idle-mode comparator increases
the peak inductor current, allowing more energy to be
transferred per cycle. Since fewer cycles are required, the
switching frequency is reduced. This keeps the external PFET off for longer periods, minimizing switching losses and
increasing efficiency.
The light-load output noise spectrum widens due to variable
switching frequency in idle-mode, but output ripple remains
low. Using the Typical Operating Circuit, with a 9V input and
a 125mA load current, output ripple is less than 40mV.
Soft-Start and Current Limiting
The MAX747 draws its highest current at power-up. If The
power source to the MAX747 cannot provide this initial
elevated current, the circuit may not function correctly. For
example, after prolonged use, a battery’s increased series
resistance may prevent it from providing adequate initial
surge currents when the MAX747 is brought out of
shutdown. Using Soft-Start (SS) minimizes the possibility of
overloading the incoming supply at power-up by gradually
increasing the peak current limit. Connect an external
capacitor from SS to ground to reduce the initial peak
currents drawn from the supply.
The steady-state SS pin voltage is typically 3.8V. On
power-up, SS sources 1µA until the SS voltage reaches
3.8V. The current-limit comparator inhibits EXT switching
until the SS voltage reaches 1.8V. The maximum current
limit is set by:
IPK =
8
VLIMIT
150mV (typ)
=
RSENSE
RSENSE
The MAX747 provides a low-battery comparator that
compares the voltage on LBI to the reference voltage.
LBO, an open-drain output, goes low when the LBI
voltage is below VREF. Use a resistor-divider network
as shown in Figure 4 to set the trip voltage (V TRIP) to
the desired level. In this circuit, LBO goes low when
V+ ≤ VTRIP. LBO is high impedance in shutdown mode.
__________________Design Procedure
Setting the Output Voltage
The MAX747’s output voltage can be set to 5V by
grounding FB, or adjusted from 2V to 14V using
external resistors R4 and R5, configured as shown in
Figure 5. Select feedback resistor R4 from the 10kΩ to
1MΩ range. R5 is given by:

V
R5 = (R4)  OUT − 1

 2V
Selecting RSENSE
First, approximate the peak current assuming IPK is
(1.1)(ILOAD), where ILOAD is the maximum load current.
Once all component values have been determined, the
actual peak current is given by:
 VOUT  
VOUT 
IPK = ILOAD + 
 1 −
(2L)
(f
)
VIN 

OSC 

Next, determine the value of RSENSE such that:
RSENSE =
VLIMIT (MIN)
IPK
=
125mV
IPK
For example, to obtain 5V at 3A, IPK = 3.3A and RSENSE =
125mV/3.3A = 38mΩ.
The sense resistor should have a power rating greater
than (IPK2)(RSENSE) (with an adequate safety margin).
With a 3A load current, IPK = 3.3A and RSENSE = 38mΩ.
The power dissipated by the resistor (assuming an 80%
_______________________________________________________________________________________
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller


VOUT + VDIODE
Duty cycle (%) = 
 (100%)
V
+
−
V
+
V

SW
DIODE 
where VSW is the voltage drop across the external PFET and sense resistor, and can be approximated as
(ILOAD)[RDS(ON) + RSENSE].
Inductor Selection
Once the sense resistor value is determined, the
inductor is determined from the following equation. The
value of inductor L ensures proper slope
compensation. Continuing with the above example,
(RSENSE ) (VOUT(MAX) )
L =
(VRAMP(MAX) ) (fOSC )
=
(38mΩ) (5V)
= 38µH
(50mV) (100kHz)
Although 38µH is the calculated value, the component
used may have a tolerance of ±30% or more. Make
sure the inductor’s saturation current rating (the current
at which the core begins to saturate and the
inductance starts to fall) exceeds the peak current set
by RSENSE.
Inductors with molypermalloy powder (MPP), Kool Mµ,
or ferrite are recommended. Inexpensive iron powder
core inductors are not suitable due to their increased
core losses. MPP and Kool Mµ cores have low
permeability, allowing larger currents.
For highest efficiency, use a coil with low DC
resistance. To minimize radiated noise, use a toroid,
pot core, or shielded coil.
External P-FET Selection
To ensure the external P-FET is fully on, use logic-level,
or low threshold P-FETs when the minimum input
voltage is less than 8V.
When selecting the P-FET, three important parameters
to note are total gate charge (Q g ), on resistance
(RDS(ON)), and reverse transfer capacitance (CRSS).
Qg, the total gate charge, includes all capacitances
associated with charging the gate. Use the typical Qg
value for best results; the maximum value is usually
overspecified since it is a guaranteed limit and not the
measured value. The typical total gate charge should
be ≤ 50nC. Larger numbers mean that EXT may not be
able to adequately drive the gate. EXT sink/source
capability (IEXT) is typically 140mA.
There are two losses associated with the P-FET’s power
dissipation: I 2 R losses and switching losses. CCM
power dissipation (PD) is approximated by:
(
[
)
]
2
PD = Duty Cycle IPK  RDS(ON) +


(
) ( ) (fOSC ) 

2
  V +  CRSS IPK

IEXT





where the duty cycle is approximated by VOUT/V+, fOSC =
100kHz, and RDS(ON) and CRSS are given in the data
sheet of the chosen P-FET. In the equation, RDS(ON) is
assumed to be constant, but is actually a function of
temperature. Note that the equation does not account
for losses incurred by charging and discharging the
VIN
VIN
12
12
R2
1
MAX747
LBI
V+
…TO VOUT OR VIN
V+
R3
100k
LBO
14
5
...to VOUT
C6*
OUT
8
N.C.
GND
GND
13
R2 = R1
( VVTRIP
-1)
TH
VTH = 2.0V
Figure 4. Input Voltage Monitor Circuit
R4
MAX747
LOW-BATTERY
OUTPUT
R1
R5
FB
13
R4 = 10kΩ TO 1MΩ
VOUT
R5 = R4
-1
2V
(
)
* SEE COMPENSATION CAPACITOR SECTION
Figure 5. Adjustable Output Circuit
_______________________________________________________________________________________
9
MAX747
duty cycle) is 331mW. Metal film resistors are
recommended. Do not use wire-wound resistors because
their inductance will adversely affect circuit operation.
Determine the duty cycle for CCM from the following
equation:
MAX747
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
gate capacitance, because that energy is dissipated
by the gate-drive circuitry, not the P-FET.
The Standard Application Circuit (Figure 1a, 1b) uses
an 8-pin Si9405DY surface-mount P-FET that has 0.1Ω
on resistance with a 10V VGS. Optimum efficiency is
obtained when the voltage at the drain swings between
the supply rails (within a few hundred mV).
Diode Selection
The MAX747’s high switching frequency demands a
high-speed rectifier. Schottky diodes are recommended.
Ensure that the Schottky diode average current rating
exceeds the load current level.
Capacitor Selection
Output Filter Capacitor
The output filter capacitor C1 should have a low
effective series resistance (ESR), and its capacitance
should remain fairly constant over temperature. This is
especially true when in CCM, since the output filter
capacitor and the load form the dominant pole that
stabilizes the loop. 430µF is adequate for load currents
up to 2.3A in Figure 1a. At low input/output
differentials, it may be necessary to use much larger
output filter capacitors to maintain adequate loadtransient response. See the AC Stability with Low
Input/Output Differentials section.
Sprague 595D surface-mount solid tantalum capacitors
and Sanyo OS-CON through-hole capacitors are
recommended due to their extremely low ESR. OS-CON
capacitors are particularly useful at low temperatures.
For best results when using other capacitors, increase
the output filter capacitor’s size or use capacitors in
parallel to reduce ESR.
Input Bypass Capacitor
The input bypass capacitor C2 reduces peak currents
drawn from the voltage source, and also reduces noise
at the voltage source caused by the MAX747’s fast
switching action (this is especially important when other
circuitry is operated from the same source). The input
capacitor ripple current rating must exceed the RMS
input current.
IRMS = RMS AC input current
 V

OUT (VIN − VOUT ) 
= ILOAD 
VIN




For load currents up to 2.5A, 100µF (C2) in parallel with
a 0.1µF (C3) is adequate. Smaller bypass capacitors
may be acceptable for lighter loads. The input voltage
source impedance determines the capacitor size
10
required at the V+ input. As with the output filter
capacitor, a low-ESR capacitor (Sanyo OS-CON,
Sprague 595D, or equivalent) is recommended for
input bypassing.
Soft-Start and Reference Capacitors
A typical value for the soft-start capacitor C4 is 0.1µF,
which provides a 380ms ramp to full current limit. Use
values in the 0.001µF and 1µF range. The nominal time
for C4 to reach its steady-state value is given by:
t SS (sec) = (C4) (3.8 × 106 )
Note that tSS does not equal the time it takes for the
MAX747 to power up, although it does affect start-up
time. Start-up time is also a function of the input voltage
and load current. With a 2.5A load current, a 7V input
voltage, and a 0.1µF soft-start capacitor, power-up
takes typically 360ms.
Bypass REF with a 0.22µF capacitor (C5).
Compensation Capacitor
With a fixed +5V output, connect the compensation
capacitor (C6) between CC and GND to optimize
transient response. Appropriate compensation is
determined by the ESR of the output filter capacitor
(C1) and the feedback voltage-sense resistor network.
270pF is adequate for applications where V+ ≤ 9V.
Over the full input voltage range, increase C6 to 470pF.
C6 also depends on the load current, so for light loads,
C6’s value can be reduced. If appropriate
compensation is not obtained using 470pF, use the
following equations to determine C6:
For fixed 5V output operation,
C6 =
(C1) (ESRC1)
24kΩ
For adjustable-output operation, FB becomes the
compensation input pin and CC is left unconnected.
Connect C6 between FB and GND in parallel with R4
(Figure 5). C6 is determined by:
C6 =
(C1) (ESRC1)
R4 II R5
For example, with a fixed 5V output, C1 = 330µF and
an ESRC1 of 0.04Ω (at a 100kHz frequency),
C6 =
(C1) (ESRC1)
= 783pF
24kΩ
______________________________________________________________________________________
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
VIN
 (V
− VREF ) 
R2 = R1  TRIP

VREF


Connect a pull-up resistor (e.g., 100kΩ) between LBO
and VOUT (Figure 4).
__________Applications Information
Layout Considerations
Due to high current levels and fast switching
waveforms, which radiate noise, proper MAX747 PC
board layout is essential. Protect sensitive analog
grounds by using a star ground configuration. Use an
adequate ground plane and minimize ground noise by
connecting GND, the anode of the steering Schottky
diode, the input bypass capacitor ground lead, and the
output filter capacitor ground lead to a single point
(star ground configuration). Also, minimize lead lengths
to minimize stray capacitance, trace resistance, and
radiated noise. Place bypass capacitor C3 as close as
possible to V+ and GND.
AV+ and CS are the inputs to the differential-input
current-sense amplifier. Use a Kelvin connection
across the sense resistor as shown in Figure 6. Note
that even though AV+ also functions as the supply
voltage for sensitive analog circuitry, a separate AV+
bypass capacitor should not be used. By not using a
capacitor, any noise appearing at the CS input will also
appear at the AV+ input and will appear as a commonmode signal to the current-sense amplifier. A separate
AV+ capacitor causes the noise to appear only on one
input, and this differential noise will be amplified,
adversely affecting circuit operation.
Similarly, CC (or FB in adjustable-output operation) is a
sensitive input that should not be shorted to any node.
Avoid shorting CC when probing the circuit, as this
may damage the device.
Switching Waveforms
A region exists between CCM and DCM where the
inductor current operates in both modes, as shown in
the Idle-Mode Moderate current EXT waveform in the
Typical Operating Characteristics . As the output
voltage varies, it is fed back into CC and the duty cycle
is adjusted to compensate for this change. The switch
is considered off when V EXT ≤ the P-FET’s V GS
threshold voltage. Once the switch is off, the voltage at
EXT is pulled to V+ and the P-FET drain voltage is a
Schottky diode drop below GND. However, in this “in-
V+ AV+
KELVIN SENSE
CONNECTION
RSENSE
MAX747
CS
EXT
P
L1
VOUT
Figure 6. Kelvin Connection for Current-Sense Amplifier
between” mode (due to the changing duty cycle
inherent with DCM), when the device is at maximum
duty cycle, EXT turns off at V+ - V GS . But it is not
always pulled to V+ because the switch sometimes
turns on again after a minimum off-time before EXT can
be pulled to V+. The result is short spikes that appear
on the EXT waveform in the Typical Operating
Characteristics.
AC Stability with Low
Input/Output Differentials
At low input/output differentials, the inductor current
cannot slew quickly to respond to load changes, so the
output filter capacitor must hold up the voltage as the
load transient is applied. In Figure 1a’s circuit, for
V+ = 6.5V, increase the output filter capacitor to 700µF
(Sprague 595D low-ESR capacitors) to obtain a
transient response less than 250mV with a load step
from 200mA to 2.5A. For V+ = 6V and V OUT = 5V,
increase the output filter capacitor to approximately
1000µF. As V+ increases, the device will no longer be
operating near full duty cycle with light loads, allowing
it to adjust to full duty cycle when the load transient is
applied and, in turn, allowing smaller output filter
capacitors to be used.
Dual-Mode Operation
The MAX747 is designed in either fixed-output mode
(5V-output, FB = GND) or in adjustable mode (FB = 2V)
using a resistor divider. It is not designed to be
switched from one mode to another when powered up;
however, in adjustment mode, switching between two
different resistor dividers is acceptable.
______________________________________________________________________________________
11
MAX747
Setting the Low-Battery Detector Voltage
Select R1 between 10kΩ and 1MΩ.
MAX747
High-Efficiency PWM, Step-Down
P-Channel DC-DC Controller
Additional Notes
When probing the MAX747 circuit, avoid shorting AV+
to GND (the two pins are adjacent to each other) as
this may cause the IC to malfunction due to large
ground currents. Also, the MAX747 may continue to
operate with AV+ disconnected, but erratic switching
waveforms will appear at EXT. Finally, due to its fast
switching and high drive capability requirements, EXT
is a low-impedance point that is not short-circuit
protected. Therefore, do not short EXT to any node
(including AGND and V+, which are adjacent to EXT)
to prevent damaging the device.
___________________Chip Topography
MAX747
LBI LBO GND
SS
V+
Table 1. Component Suppliers
EXT
REF
SUPPLIER
PHONE
FAX
INDUCTORS
Coiltronics
(305) 781-8900
(305) 782-4163
Gowanda
(716) 532-2234
(716) 532-2702
Sumida USA
(708) 956-0666
(708) 956-0702
Sumida Japan
81-3-3607-511
81-3-3607-5428
Kemet
(803) 963-6300
(803) 963-6322
Matsuo
(714) 969-2491
(714) 960-6492
Nichicon
(708) 843-7500
(708) 843-2798
Sprague
(603) 224-1961
(603) 224-1430
Sanyo USA
(619) 661-6322
Sanyo Japan
81-3-3837-6242
United Chemi-Con
(714) 255-9500
SHDN
0.130"
(3.30mm)
CAPACITORS
AGND
FB CC
AV+ OUT
CS
0.080"
(2.03mm)
(714) 255-9400
DIODES
SUBSTRATE CONNECTED TO V+;
TRANSISTOR COUNT: 508.
Motorola
(800) 521-6274
Nihon USA
(805) 867-2555
(805) 867-2698
Nihon Japan
81-3-3494-7411
81-3-3494-7414
Harris
(407) 724-3739
(407) 724-3937
International Rectifier
(213) 772-2000
(213) 772-9028
Siliconix
(408) 988-8000
(408) 727-5414
(512) 992-7900
(512) 992-3377
POWER TRANSISTORS
RESISTORS
IRC
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
12 __________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 (408) 737-7600
© 1993 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
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