MPS MPQ4561DQ-AEC1 1.5a, 2mhz, 55v step-down converter available in aec-q100 Datasheet

MPQ4561-AEC1
1.5A, 2MHz, 55V
Step-Down Converter
Available in AEC-Q100
DESCRIPTION
FEATURES
The MPQ4561 is a high-frequency, step-down,
switching regulator with an integrated internal
high-side high voltage power MOSFET. It
provides up to a 1.5A output with current mode
control for fast loop response and easy
compensation.

The wide 3.8V-to-55V input range accommodates
a variety of step-down applications, including
those in automotive input environment. A 12µA
shutdown mode supply current makes it suitable
for battery-powered applications.







A scaled-down switching frequency in light-load
conditions provides high power-conversion
efficiency over a wide load range while reducing
switching and gate driver losses.
The frequency foldback prevents inductor current
runaway during startup and thermal shutdown
provides reliable, fault tolerant operation.
By switching at 2MHz, the MPQ4561 is able to
prevent EMI (Electromagnetic Interference) noise
problems, such as those found in AM radio and
ADSL applications.



Guaranteed Industrial/Automotive
Temperature Range Limits
Wide 3.8V to 55V Operating Input Range
300mΩ Internal Power MOSFET
Up to 2MHz Programmable Switching
Frequency
140μA Quiescent Current
Stable with Ceramic Capacitor
External Soft-Start
Up to 95% Efficiency
Output Adjustable from 0.8V to 52V
Available in QFN10 (3mmx3mm) package
AEC-Q100 Qualified
APPLICATIONS





High-Voltage Power Conversion
Automotive Systems
Industrial Power Systems
Distributed Power Systems
Battery Powered Systems
All MPS parts are lead-free and adhere to the RoHS directive. For MPS green
status, please visit MPS website under Quality Assurance.
“MPS” and “The Future of Analog IC Technology” are Registered Trademarks of
Monolithic Power Systems, Inc.
The MPQ4561 is available in a 3mm x 3mm
QFN10 package with an exposed pad.
TYPICAL APPLICATION
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
ORDERING INFORMATION
Part Number
MPQ4561DQ*
MPQ4561DQ-AEC1**
Package
Top Marking
9C
9C
QFN10 (3x3mm)
Junction Temperature (TJ)
–40°C to +125°C
* For Tape & Reel, add suffix –Z (e.g. MPQ4561DQ–AEC-Z)
For RoHS compliant packaging, add suffix –LF (e.g. MPQ4561DQ–AEC-LF–Z)
** Available End Sept. 2011
PACKAGE REFERENCE
ABSOLUTE MAXIMUM RATINGS (1)
Thermal Resistance
Supply Voltage (VIN).................... –0.3V to +60V
Switch Voltage (VSW)......... –0.5V to (VIN + 0.5V)
BST to SW .................................... –0.3V to +5V
All Other Pins ................................ –0.3V to +5V
(2)
Continuous Power Dissipation (TA = 25°C)
............................................................2.5W
Junction Temperature .............................. 150°C
Lead Temperature ................................... 260°C
Storage Temperature .............. –65°C to +150°C
QFN10 (3x3mm) ..................... 50 ...... 12 ... °C/W
Recommended Operating Conditions
(3)
Supply Voltage VIN .......................... 3.8V to 55V
Output Voltage VOUT........................ 0.8V to 52V
Maximum Junction Temp. (TJ) ............... +125C
(4)
θJA
θJC
Notes:
1) Exceeding these ratings may damage the device.
2) The maximum allowable power dissipation is a function of the
maximum junction temperature TJ(MAX), the junction-toambient thermal resistance θJA, and the ambient temperature
TA. The maximum allowable continuous power dissipation at
any ambient temperature is calculated by PD(MAX)=(TJ(MAX)TA)/θJA. Exceeding the maximum allowable power dissipation
will cause excessive die temperature, and the regulator will go
into thermal shutdown. Internal thermal shutdown circuitry
protects the device from permanent damage.
3) The device is not guaranteed to function outside of its
operating conditions.
4) Measured on JESD51-7 4-layer board.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
ELECTRICAL CHARACTERISTICS
VIN = 12V, VEN = 2.5V, VCOMP = 1.4V, TJ= -40°C to +125°C, unless otherwise noted. Typical values
are at TJ=25°C.
Parameter
Symbol Condition
Feedback Voltage
Upper Switch On Resistance
(5)
VFB
4.5V < VIN < 55V
–40°C to +125°C
RDS(ON)
VBST – VSW = 5V
–40°C to +125°C
Upper Switch Leakage
VEN = 0V, VSW = 0V
–40°C to +125°C
(Duty Cycle ≤ 60%)
Current Limit
COMP to Current Sense
Transconductance
Error Amp Voltage Gain
Error Amp Transconductance
Error Amp Min Source current
Error Amp Min Sink current
VIN UVLO Threshold
VIN UVLO Hysteresis
(5)
Soft-Start Time
Oscillator Frequency
Min
Typ
Max
Units
0.766
0.795
0.829
V
200
300
475
mΩ
1.7
GCS
ICOMP = ±3µA
VFB = 0.7V
VFB = 0.9V
–40°C to +125°C
0V < VFB < 0.8V, CSS=10nF
RFREQ = 95kΩ
–40°C to +125°C
μA
1
2.4
2.5
3.3
4.5
A/V
400
120
10
-10
V/V
µA/V
µA
µA
3.0
3.6
0.35
1.6
0.7
A
V
V
ms
1
1.3
MHz
20
µA
Shutdown Supply Current
VEN < 0.3V
12
Quiescent Supply Current
Thermal Shutdown
Minimum Off Time
(5)
Minimum On Time
No load, VFB = 0.9V
Hysteresis = 20°C
140
150
100
100
130
µA
°C
ns
ns
EN Up Threshold
–40°C to +125°C
1.55
1.8
V
–40°C to +125°C
EN Threshold Hysteresis
1.3
320
mV
Note:
5) Not production tested. Specified by design and bench characterization.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
PIN FUNCTIONS
QFN
Pin #
Name
1, 2
SW
3
EN
4
COMP
5
FB
6
GND,
Exposed Pad
7
FREQ
8
SS
9
VIN
10
BST
Description
Switch Node. Output of the high-side switch. Requires a low VF Schottky rectifier to
ground. Place the rectifier close to the SW pins to reduce switching spikes..
Enable Input. Pull this pin below the specified threshold to shut the chip down. Pull it
above the specified threshold or leaving it floating to enable the chip.
Compensation. GM error amplifier output. Apply control-loop frequency compensation to
this pin.
Feedback. This is the input to the error amplifier. Connect an external resistive divider
connected between the output and GND. Compare to the internal +0.8V reference to set
the regulation voltage.
Ground. Connect as close as possible to the output capacitor and avoid high-current
switching paths. Connect the exposed pad to GND plane for optimal thermal
performance.
Switching Frequency Program Input. Connect a resistor from this pin to ground to set the
switching frequency.
Soft start programming mode. Connect a capacitor between SS and GND to set the soft
start time.
Input Supply. Supplies power to all the internal control circuitry, both BS regulators, and
the high-side switch. Place a decoupling capacitor to ground close to this pin to minimize
switching spikes.
Bootstrap. This is the positive power supply for the internal floating high-side MOSFET
driver. Connect a bypass capacitor between this pin and the SW pin.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
TYPICAL CHARACTERISTICS
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
TYPICAL CHARACTERISTICS (continued)
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS
VIN = 12V, VOUT =3.3V, C1 = 10µF, C2 = 22µF, L1 = 10µH and TA = +25°C, unless otherwise noted.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
TYPICAL PERFORMANCE CHARACTERISTICS (continued)
VIN = 12V, VOUT =3.3V, C1 = 10µF, C2 = 22µF, L1 = 10µH and TA = +25°C, unless otherwise noted.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
BLOCK DIAGRAM
Figure 1: Functional Block Diagram
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
OPERATION
The MPQ4561 is a programmable-frequency,
non-synchronous, step-down, switching regulator
with an integrated high-side high-voltage power
MOSFET. It provides a single highly efficient
solution with current mode control for fast loop
response and easy compensation. It features a
wide input voltage range, external soft-start
control for start-up ramp-up flexibility, and
precision current limiting. Its very low operational
quiescent current makes it suitable for battery
powered applications.
PWM Control Mode
At moderate to high output current, the MPQ4561
operates in a fixed frequency, peak current
control mode to regulate the output voltage. The
internal clock initiates a PWM cycle. The power
MOSFET turns on and remains on until its
current reaches the value set by the COMP
voltage. When the power MOSFET is off, it
remains off for at least 100ns before the next
cycle starts. If, in one PWM period, the power
MOSFET current does not reach the COMP set
current value, the power MOSFET remains on to
saves on a turn-off operation.
Pulse Skipping Mode
Under light load condition the switching
frequency drops zero to reduce the switching loss
and driving losses.
Error Amplifier
The error amplifier compares the FB pin voltage
with the internal reference (REF) and outputs a
current proportional to the difference between the
two. This output current then charges the
external compensation network to form the
COMP voltage, which controls the power
MOSFET current.
While in operation, the minimum COMP voltage
is clamped to 0.9V and its maximum is clamped
to 2.0V. COMP is internally pulled down to GND
in shutdown mode. Avoid pulling COMP up
beyond 2.6V.
Internal Regulator
The 2.6V internal regulator powers most of the
internal circuits. This regulator takes the VIN
input and operates in the full VIN range. When
VIN exceeds 3.0V, the output of the regulator is in
full regulation: When VIN is less than 3.0V, the
output drops to 0V.
Enable Control
The MPQ4558 has a dedicated enable control
pin (EN): An input voltage that exceeds an upper
threshold enables the chip, while a voltage the
drops below a lower threshold disables the chip.
Its falling threshold is precisely 1.2V, and its
rising threshold is 300mV higher, or 1.5V.
When floating, EN is pulled up to about 3.0V by
an internal 1µA current source to enable the chip.
Pulling it down requires a 1µA current.
When EN drops below 1.2V, the chip enters the
lowest shutdown current mode. When EN
exceeds 0V but remains below its rising
threshold, the chip is still in shutdown mode but
with a slightly higher shutdown current.
Under-Voltage Lockout
Under-voltage lockout (UVLO) protects the chip
from operating at an insufficient supply voltage.
The UVLO rising threshold is about 3.0V while its
falling threshold is a consistent 2.6V.
Thermal Shutdown
Thermal shutdown prevents the chip from
operating at exceedingly high temperatures.
When the silicon die temperature exceeds its
upper threshold, it shuts down the whole chip.
When the temperature falls below its lower
threshold, the chip is enabled again.
Floating Driver and Bootstrap Charging
An external bootstrap capacitor powers the
floating power MOSFET driver. This floating
driver has its own UVLO protection with a rising
threshold of 2.5V and a hysteresis of 300mV.
The driver’s UVLO is connected to the SS: If the
bootstrap voltage hits its UVLO, the soft-start
circuit resets. To prevent noise, there is 20µs
delay before the reset action. When the device
exits the bootstrap UVLO condition, the reset
turns off and then soft-start process resumes.
The dedicated internal bootstrap regulator
charges and regulates the bootstrap capacitor to
about 5V. When the voltage between the BST
and SW nodes falls below regulation, a PMOS
pass transistor connected from VIN to BST turns
on. The charging current path goes from VIN, to
BST and then to SW. The external circuit must
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
provide enough voltage headroom to facilitate
charging.
As long as VIN is sufficiently higher than SW, the
bootstrap capacitor will charge. When the power
MOSFET is ON, VIN is about equal to SW so the
bootstrap capacitor cannot charge. When the
external diode is on, the difference between VIN
and SW is at its largest, thus making it the best
period to charge. When there is no current in the
inductor, SW equals the output voltage VOUT so
the difference between VIN and VOUT can charge
the bootstrap capacitor.
Under higher duty-cycle operation conditions, the
time period available for bootstrap charging is
smaller so the bootstrap capacitor may not
sufficiently charge.
In case the internal circuit does not have
sufficient voltage and the bootstrap capacitor is
not charged, extra external circuitry can ensure
the bootstrap voltage is in the normal operational
region. Refer to the External Bootstrap Diode in
Application section.
The DC quiescent current of the floating driver is
about 20µA. Make sure the bleeding current at
the SW node is higher than this value, such that:
VO
 20μA
(R1  R 2)
Current-Comparator and Current Limit
A current-sense MOSFET accurately senses the
current in the power MOSFET. This signal is then
fed to the high speed current comparator for
current-mode–control purposes, which uses it as
one of its inputs with the COMP voltage. When
the power MOSFET turns on, the comparator is
first blanked until the end of the turn-on transition
to avoid noise issues. When the sensed current
exceeds the COMP voltage, the comparator
output is low and the power MOSFET turns off.
The cycle-by-cycle maximum current of the
internal power MOSFET is internally limited.
IO 
Short Circuit Protection
When the output is shorted to the ground, the
switching frequency folds back and the current
limit falls to reduce the short circuit current. When
the FB voltage equals 0V, the current limit falls to
about 50% of its full current limit. The FB voltage
reaches its 100% of its current limit when it
exceeds 0.4V
When the short-circuit FB voltage is low, the SS
drops by VFB and SS ≈ VFB + 100mV. If the short
circuit is removed, the output voltage recovers at
the SS rate. When FB is high enough, the
frequency and current limit return to normal
values.
Startup and Shutdown
If both VIN and VEN exceed their appropriate
thresholds, the chip starts. The reference block
starts first, generating stable reference voltage
and currents, and then the internal regulator is
enabled. The regulator provides stable supply for
the remaining circuits.
While the internal supply rail is up, an internal
timer blanks the power MOSFET OFF for about
50µs to avoid start-up glitches. When the internal
soft-start block is enabled, it first holds its SS
output low to ensure the other circuits are ready
and then slowly ramps up.
Three events can shut down the chip: EN low,
VIN low and thermal shutdown. In the shutdown
procedure, power MOSFET is turned off first to
avoid any fault triggering. The COMP voltage and
the internal supply rail are then pulled down.
Programmable Oscillator
An external resistor (RFREQ) connected from the
FREQ pin to GND sets the MPQ4558 oscillating
frequency. Calculate the value of RFREQ from:
RFREQ (k) 
100000
5
fS (kHz)
For fSW=500kHz, RFREQ=195kΩ.
Soft-Start
Soft-start prevents the converter output voltage
from overshooting during startup and short circuit
recovery phases. Internally the soft-start voltage
(VSS) is the voltage at SS pin offset by about 1V.
VSS is applied on the error amplifier in parallel
with the internal reference voltage REF. VSS or
REF (whichever is lower) controls the error
amplifier . When VSS ramps up from 0 to high, the
controller tries to regulate FB from zero to REF at
the VSS ramp-up pace.
A 5μA current source pulls up the SS pin. Given
the soft-start capacitor (CSS), the soft-start time is
about the time CSS voltage changes by 0.8V. So
the soft-start time can be estimated as:
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
t SS 
CSS  0.8 V
5μA
Figure 2: Recommend SS time vs SS
Capacitance
Figure 2 shows the soft-start time with different
external soft-start capacitance. The typical softstart capacitance is recommended from 5.6nF to
220nF.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
APPLICATION INFORMATION
COMPONENT SELECTION
Setting the Output Voltage
Set the output voltage with a resistor divider
between the output voltage and the FB pin. The
voltage divider drops the output voltage down to
the feedback voltage by the ratio:
VFB =VOUT 
ILP  ILOAD 

VOUT
V
 1  OUT
2  fS  L1 
VIN



Where ILOAD is the load current.
R2
Table 1 lists a number of suitable inductors from
various manufacturers. The choice the inductor
style mainly depends on the price vs. size
requirements and any EMI requirement.
R1+R2
Thus the output voltage is:
VOUT =VFB 
Choose an inductor that will not saturate under
the maximum inductor peak current. Calculate
the peak inductor current with:
R1+R2
R2
For example, for R2 = 10kΩ, R1 can be
determined by:
R1=12.5  (VOUT -0.8)(kΩ)
For example, for a 3.3V output voltage, R2 is
10kΩ, and R1 is 31.6kΩ.
Inductor
The inductor supplies constant current to the
output load while being driven by the switched
input voltage. A larger value inductor will result in
less ripple current that will lower the output ripple
voltage. However, a larger-valued inductor is
physically larger, has a higher series resistance,
or lower saturation current.
Output Rectifier Diode
The output rectifier diode supplies the current to
the inductor when the high-side switch is off. To
reduce losses due to the forward diode voltage
and recovery times, use a Schottky diode.
Choose a diode whose maximum reverse voltage
rating is greater than the maximum input voltage,
and whose current rating is greater than the
maximum load current. Table 2 lists example
Schottky diodes and manufacturers.
Generally, determine an appropriate inductance
value by selecting the peak-to-peak inductor
ripple current equal to approximately 30% of the
maximum switch current limit. Also, make sure
that the peak inductor current is below the
maximum switch current limit. Calculate the
inductance value with:
L1=
VOUT
fs  ΔIL
 (1-
VOUT
VIN
)
Where:

VOUT is the output voltage,

VIN is the input voltage,

fS is the switching frequency, and

∆IL is the peak-to-peak inductor ripple current.
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
Table 1: Inductor Selection Guide
Inductance
(µH)
Max DCR
(Ω)
Current Rating
(A)
Dimensions
3
L x W x H (mm )
7447789004
4.7
0.033
2.9
7.3x7.3x3.2
744066100
10
0.035
3.6
10x10x3.8
744771115
15
0.025
3.75
12x12x6
744771122
22
0.031
3.37
12x12x6
RLF7030T-4R7
4.7
0.031
3.4
7.3x6.8x3.2
SLF10145T-100
10
0.0364
3
10.1x10.1x4.5
SLF12565T-150M4R2
15
0.0237
4.2
12.5x12.5x6.5
SLF12565T-220M3R5
22
0.0316
3.5
12.5x12.5x6.5
FDV0630-4R7M
4.7
0.049
3.3
7.7x7x3
919AS-100M
10
0.0265
4.3
10.3x10.3x4.5
919AS-160M
16
0.0492
3.3
10.3x10.3x4.5
919AS-220M
22
0.0776
3
10.3x10.3x4.5
Part Number
Wurth Electronics
TDK
Toko
Table 2: Diode Selection Guide
Diodes
Voltage/
Current
Rating
Manufacturer
B290-13-F
90V, 2A
Diodes Inc.
B380-13-F
80V, 3A
Diodes Inc.
CMSH2-100M
100V, 2A
Central Semi
CMSH3-100MA
100V, 3A
Central Semi
Input Capacitor
The input current to the step-down converter is
discontinuous and therefore requires a capacitor
to supply the AC current to the step-down
converter while maintaining the DC input voltage.
Use capacitors with low equivalent series
resistance (ESR) for the best performance.
Ceramic capacitors are preferred, but tantalum or
low-ESR electrolytic capacitors may also suffice.
For simplification, choose the input capacitor with
an RMS current rating greater than half of the
maximum load current. The input capacitor (C1)
can be electrolytic, tantalum or ceramic.
When using electrolytic or tantalum capacitors,
include a small, high-quality ceramic capacitor—
i.e. 0.1μF—placed as close to the IC as possible.
When using ceramic capacitors, make sure that
they have enough capacitance to provide
sufficient charge to prevent excessive voltage
ripple at the input. The input voltage ripple
caused by the capacitance can be estimated by:
ΔVIN 

ILOAD
V
V 
 OUT   1  OUT 
fS  C1 VIN 
VIN 
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
Output Capacitor
The output capacitor (C2) maintains the DC
output voltage. Use ceramic, tantalum, or lowESR electrolytic capacitors for best results. Low
ESR capacitors are preferred to keep the output
voltage ripple low. The output voltage ripple can
be estimated by:
ΔVOUT

 V  
V
1
 OUT   1  OUT    RESR 

fS  L 
VIN  
8  fS  C2 
Where L is the inductor value and RESR is the
equivalent series resistance (ESR) value of the
output capacitor.
For ceramic capacitors, the impedance at the
switching frequency is dominated by the
capacitance. The output voltage ripple is mainly
caused by the capacitance. For simplification, the
output voltage ripple can be estimated by:
ΔVOUT 
 V 
VOUT
  1  OUT 
2
8  fS  L  C2 
VIN 
For tantalum or electrolytic capacitors, the ESR
dominates the impedance at the switching
frequency. For simplification, the output ripple
can be approximated as:
ΔVOUT 
VOUT 
V
  1  OUT
fS  L 
VIN

  RESR

The characteristics of the output capacitor also
affect the stability of the regulation system. The
MPQ4561 can be optimized for a wide range of
capacitance and ESR values.
Compensation Components
MPQ4558 employs current-mode control for easy
compensation and fast transient response. The
COMP pin controls the system stability and
transient response—the COMP pin is the output
of the internal error amplifier. A capacitor-resistor
combination in series sets a pole-zero
combination to control the characteristics of the
control system. The DC gain of the voltage
feedback loop is given by:
A VDC  RLOAD  GCS  A VEA 
VFB
VOUT
Where

AVEA is the error amplifier voltage gain,
400V/V,

GCS
is
the
current
transconductance, 4.5A/V, and

RLOAD is the load resistor value.
sense
The system has two poles of importance: One is
caused by the compensation capacitor (C3) and
the output resistor of error amplifier; the other is
caused by the output capacitor and the load
resistor. These poles are located at:
fP1 
GEA
2 π C3  A VEA
fP2 
1
2π  C2  RLOAD
Where,
GEA
is
the
transconductance, 120μA/V.
error
amplifier
The system has one zero of importance, from C3
and the compensation resistor (R3). This zero is
located at:
fZ1 
1
2 π C3  R3
The system may have another important zero if
the output capacitor has a large capacitance or a
high ESR value. The zero, due to the ESR and
the output capacitor value, is located at:
fESR 
1
2π  C2  RESR
In this case, a third pole set by the compensation
capacitor (C6) and R3 compensates for the effect
of the ESR zero on the loop gain. This pole is
located at:
fP 3 
1
2 π C6  R3
The compensation network shapes the converter
transfer function for a desired loop gain. The
feedback-loop unity gain at the system crossover
frequency is important: Lower crossover
frequencies result in slower line and load
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
transient responses, while higher crossover
frequencies can destabilize the system.
Generally, set the crossover frequency to
approximately 1/10 of the switching frequency
Table 3: Compensation Values for Typical
Output Voltage/Capacitor Combinations
VOUT
(V)
L (µH)
C2
(µF)
R3
(kΩ)
C3
(pF)
C7
(pF)
1.8
4.7
47
62
1000
47
2.5
4.7 - 6.8
22
36
680
None
3.3
6.8 -10
22
51
470
None
5
15 - 22
33
82
680
None
12
10
33
40.2
330
2
To optimize the compensation components for
conditions not listed in Table 3 use the following
procedure.
1. Choose the compensation resistor (R3) to set
the desired crossover frequency. Determine the
R3 value from the following equation:
R3 
2 π C2  fC VOUT

GEA  GCS
VFB
Where fC is the desired crossover frequency.
2. Choose C3 to achieve the desired phase
margin. For applications with typical inductor
values, setting the compensation zero (fZ1) below
¼ of the crossover frequency to provide sufficient
phase margin. Determine the C3 from the
following equation:
C3 
4
2 π R3  fC
3. Determine if C6 is required—if the ESR zero of
the output capacitor is located at less than half of
the switching frequency, or if the following
relationship is valid:
f
1
 S
2π  C2  RESR 2
If this is the case, then add C6 to set the pole fP3
at the location of the ESR zero. Determine the
C6 value by the equation:
C6 
C2  RESR
R3
High Frequency Operation
The MPQ4561 switching frequency can be
programmed up to 2MHz by an external resistor.
The minimum MPQ4561 ON-time of is typically
about 100ns. Pulse-skipping operation can be
seen more easily at higher switching frequencies
due to the minimum on time.
Since the internal bootstrap circuitry has higher
impedance that may not be adequate to charge
the
bootstrap
capacitor
during
each
(1-D)×tS charging period, add an external
bootstrap charging diode if the switching
frequency is about 2MHz (see External Bootstrap
Diode section for detailed implementation
information).
With higher switching frequencies, the inductive
reactance (XL) of the capacitor dominates, so
that the ESL of the input/output capacitor
determines the input/output ripple voltage at
higher switching frequencies. Because of this
ripple, use high-frequency ceramic capacitors for
the input decoupling capacitor and output the
filtering capacitor for high-frequency operation.
Layout becomes more important when the device
switches at higher frequencies. For best results,
place the input decoupling capacitor and the
catch diode as close to the MPQ4561 (VIN pin,
SW pin and PGND) as possible, with short and
wide traces. This can help to greatly reduce the
voltage spike on SW node, and lower the EMI
noise level as well.
Route the feedback trace as far from the inductor
and noisy power traces as possible. If possible,
run the feedback trace on the opposite side of
the PCB from the inductor with a ground plane
separating the two.
Place the compensation components close to the
MPQ4561. Do not place the compensation
components close to or under high dv/dt SW
node, or inside the high di/dt power loop. If this is
necessary, add a ground plane to isolate them.
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
Switching loss is expected to be increased at
high switching frequency.
To help to improve the thermal conduction, add
grid of thermal vias under the exposed pad. use
small vias (15mil barrel diameter) so that the
plating process fills the holes, thus aiding
conduction to the other side. Excessively large
holes can cause solder wicking during the reflow
soldering process. The typical pitch (distance
between the centers) between thermal vias is
typically 40mil.
External Bootstrap Diode
An external bootstrap diode may enhance the
regulator efficiency. For the cases described
below, add an external BST diode from 5V to the
BST pin:

There is a 5V rail available in the system;

VIN is no greater than 5V;

VOUT is between 3.3V and 5V;
This diode is also recommended for high duty
cycle operation (when VOUT / VIN > 65%)
applications.
The bootstrap diode can be a low cost one such
as IN4148 or BAT54.
Figure 3: External Bootstrap Diode
At no load or light load, the converter may
operate in pulse skipping mode to maintain the
output voltage in regulation: There is less time to
refresh the BST voltage. For sufficient gate
voltage under such operating conditions, chose
(VIN –VOUT) greater than 3V. For example, if the
VOUT is 3.3V, VIN needs exceed 3.3V+3V=6.3V to
maintain enough BST voltage at no load or light
load. To meet this requirement, EN pin can be
used to program the input UVLO voltage to
Vout+3V.
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
TYPICAL APPLICATION CIRCUITS
Figure 4: 1.8V Output Typical Application Schematic
Figure 5: 5V Output Typical Application Schematic
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
PCB Layout Guide
2. Place the bypass ceramic capacitors
close to the VIN pin.
PCB layout is very important to achieve stable
operation. Duplicate the EVB layout below for
optimal performance.
3. Use short and direct feedback
connections. Place the feedback
resistors and compensation components
as close to the chip as possible.
For changes, please follow these guidelines
and use Figure 6 as reference.
1. Keep the switching-current path short
and minimize the loop area formed by
the input capacitor, high-side MOSFET
and external switching diode.
4. Route the SW path away from sensitive
analog areas such as the FB path.
5. Connect
IN,
SW,
and
GND,
respectively, to a large copper area to
cool the chip to improve thermal
performance and long-term reliability
MPQ4561 Typical Application Circuit
GND
R5
R4
C3
FB
En
SW
SW
5
4
3
2
1
R1
COMP
R2
R3
L1
SW
C4
8
9
SS
Vin
10 BST
7 FREQ
6 GND
D1
R6
C5
C2
C1
Vin
GND
TOP Layer
GND
Vo
Bottom Layer
Figure 6: MPQ4561DQ Typical Application Circuit and PCB Layout Guide
MPQ4561 Rev. 1.12
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MPQ4561 – 1.5A, 2MHz, 55V STEP-DOWN CONVERTER
PACKAGE INFORMATION
QFN10 (3x3mm) (Exposed Pad)
NOTICE: The information in this document is subject to change without notice. Users should warrant and guarantee that third
party Intellectual Property rights are not infringed upon when integrating MPS products into any application. MPS will not
assume any legal responsibility for any said applications.
MPQ4561 Rev. 1.12
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5/24/2016
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20
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