LINER LT3506IFE-TR Dual monolithic 1.6a step-down switching regulator Datasheet

LT3506/LT3506A
Dual Monolithic 1.6A
Step-Down Switching Regulator
FEATURES
DESCRIPTIO
Wide Input Voltage Range, 3.6V to 25V
■ Two 1.6A Output Switching Regulators with Internal
Power Switches
■ Constant Switching Frequency
LT3506: 575kHz
LT3506A: 1.1MHz
■ Anti-Phase Switching Reduces Ripple
■ Accurate 0.8V Reference, ±1%
■ Independent Shutdown/Soft-Start Pins
■ Independent Power Good Indicators Ease Supply
Sequencing
■ Uses Small Inductors and Ceramic Capacitors
■ Small 16-Lead Thermally Enhanced 5mm × 4mm
DFN and TSSOP Surface Mount Packages
The LT®3506 is a dual current mode PWM step-down DC/DC
converter with internal 2A power switches. Both converters are synchronized to a single oscillator and run with
opposite phases, reducing input ripple current. The output
voltages are set with external resistor dividers, and each
regulator has independent shutdown and soft-start circuits.
Each regulator generates a power-good signal when its
output is in regulation, easing power supply sequencing
and interfacing with microcontrollers and DSPs.
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APPLICATIO S
■
■
■
■
■
■
Disk Drives
DSP Power Supplies
Wall Transformer Regulation
Distributed Power Regulation
DSL Modems
Cable Modems
The LT3506 switching frequency is 575kHz and the LT3506A
is 1.1MHz. These high switching frequencies allow the
use of tiny inductors and capacitors, resulting in a very
small dual 1.6A output solution. Constant frequency and
ceramic capacitors combine to produce low, predictable
output ripple voltage. With its wide input range of 3.6V to
25V, the LT3506 regulates a wide variety of power sources,
from 4-cell batteries and 5V logic rails to unregulated wall
transformers, lead acid batteries and distributed-power
supplies. Current mode PWM architecture provides fast
transient response with simple compensation components
and cycle-by-cycle current limiting. Frequency foldback
and thermal shutdown provide additional protection.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
TYPICAL APPLICATIO
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Efficiency
VIN
4.5V TO 25V
BOOST1
0.22µF
18.7k
1000pF
15k
D1
15k
6.4µH
VOUT2
3.3V
1.6A
SW2
SW1
47µF
90
BOOST2
0.22µF
4.7µH
VIN = 5V
1/2 BAT-54A
VIN1 VIN2
FB1
FB2
VC1
VC2
LT3506
33.2k
2200pF
10k
D2
10.7k
22µF
EFFICIENCY (%)
22µF
VOUT1
1.8V
1.6A
100
1/2 BAT-54A
VOUT = 3.3V
80
VOUT = 1.8V
70
60
RUN/SS1 RUN/SS2
100k
100k
PGOOD1
PGOOD2
1.5nF
PGOOD1
PGOOD2
GND
1.5nF
D1, D2: ON SEMI MBR5230LT3
3506 F01
50
0
0.5
1.0
IOUT (A)
1.5
2.0
3506 TA01b
3506afb
LT3506/LT3506A
Absolute Maximum Ratings
(Note 1)
VIN Voltage.................................................. –0.3V to 25V
BOOST Pin Voltage....................................................50V
BOOST Pin Above SW Pin..........................................25V
PG Pin Voltage...........................................................25V
RUN/SS, FB, VC Pins.................................................5.5V
Maximum Junction Temperature........................... 125°C
Operating Temperature Range (Note 2)
E Grade................................................. –40°C to 85°C
I Grade................................................ –40°C to 125°C
Storage Temperature Range.................... –65°C to 125°C
PI CO FIGURATIO
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TOP VIEW
TOP VIEW
BOOST1
1
16 FB1
BOOST1
1
16 FB1
SW1
2
15 VC1
SW1
2
15 VC1
VIN1
3
14 PG1
VIN1
3
14 PG1
VIN1
4
13 RUN/SS1
VIN1
4
VIN2
5
12 RUN/SS2
VIN2
5
17
17
13 RUN/SS1
12 RUN/SS2
VIN2
6
11 PG2
VIN2
6
11 PG2
SW2
7
10 VC2
SW2
7
10 VC2
BOOST2
8
9
BOOST2
8
9
FB2
DHD PACKAGE
16-LEAD PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W, θJC = 4.3°C/W
EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB
FB2
FE PACKAGE
16-LEAD PLASTIC TSSOP NARROW
TJMAX = 125°C, θJA = 45°C/W, θJC = 10°C/W
EXPOSED PAD (PIN 17) IS GND MUST BE SOLDERED TO PCB
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ORDER I FOR ATIO
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3506EDHD#PBF
LT3506AEDHD#PBF
LT3506IDHD#PBF
LT3506AIDHD#PBF
LT3506EFE#PBF
LT3506AEFE#PBF
LT3506IFE#PBF
LT3506AIFE#PBF
LT3506EDHD#TRPBF
LT3506AEDHD#TRPBF
LT3506IDHD#TRPBF
LT3506AIDHD#TRPBF
LT3506EFE#TRPBF
LT3506AEFE#TRPBF
LT3506IFE#TRPBF
LT3506AIFE#TRPBF
3506
3506A
3506
3506A
3506EFE
3506AEFE
3506IFE
3506AIFE
16-Lead (5mm x 4mm) Plastic DFN
16-Lead (5mm x 4mm) Plastic DFN
16-Lead (5mm x 4mm) Plastic DFN
16-Lead (5mm x 4mm) Plastic DFN
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
16-Lead Plastic TSSOP Narrow
–40°C to 85°C
–40°C to 85°C
–40°C to 125°C
–40°C to 125°C
–40°C to 85°C
–40°C to 85°C
–40°C to 125°C
–40°C to 125°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LT3506EDHD
LT3506EDHD#TR
3506
16-Lead (5mm x 4mm) Plastic DFN
–40°C to 85°C
LT3506AEDHD
LT3506AEDHD#TR
3506A
16-Lead (5mm x 4mm) Plastic DFN
–40°C to 85°C
LT3506IDHD
LT3506IDHD#TR
3506
16-Lead (5mm x 4mm) Plastic DFN
–40°C to 125°C
LT3506AIDHD
LT3506AIDHD#TR
3506A
16-Lead (5mm x 4mm) Plastic DFN
–40°C to 125°C
LT3506EFE
LT3506EFE#TR
3506EFE
16-Lead Plastic TSSOP Narrow
–40°C to 85°C
LT3506AEFE
LT3506AEFE#TR
3506AEFE
16-Lead Plastic TSSOP Narrow
–40°C to 85°C
LT3506IFE
LT3506IFE#TR
3506IFE
16-Lead Plastic TSSOP Narrow
–40°C to 125°C
LT3506AIFE
LT3506AIFE#TR
3506AIFE
16-Lead Plastic TSSOP Narrow
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3506afb
LT3506/LT3506A
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VBOOST = 8V, unless otherwise noted. (Note 2)
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
VIN(MIN)
Undervoltage Lockout
3.4
3.6
V
IINQ
Quiescent Current
IINSD
Shutdown Current
Not Switching
3.8
4.8
mA
VRUNSS = 0V
30
45
µA
VFB
Feedback Voltage
–40°C to 85°C, EDHD
–40°C to 85°C, EFE
–40°C to 125°C, IFE, IDHD
●
●
●
800
800
800
808
816
816
mV
mV
mV
IFB
FB Pin Bias Current
VFB = 800mV, VC = 0.4V
●
40
100
nA
VFB(REG)
Reference Line Regulation
VIN = 5V to 25V
gmEA
Error Amp GM
350
AV
Error Amp Voltage Gain
400
IVC
VC Source Current
VC Sink Current
VVC(THRESH)
●
792
784
784
0.005
VFB = 0.6V, VC = 0V
VFB = 1.2V, VC = 1100mV
UNITS
%/V
uMhos
30
30
µA
µA
VC Switching Threshold
0.7
V
VVC(CLAMP)
VC Clamp Voltage
1.9
V
fSW
Switching Frequency
LT3506
LT3506A
Switching Phase
(Note 5)
DC
Maximum Duty Cycle
LT3506
LT3506A
VFB(SWTHRESH)
Frequency Shift Threshold on FB
fFOLD
Foldback Frequency
VFB = 0V
ISW
Switch Current Limit
(Note 3)
VSW(SAT)
Switch VCESAT (Note 4)
ISW = 1A
ILSW
Switch Leakage Current
VBOOST(MIN)
Minimum Boost Voltage Above Switch
ISW = 1A
IBOOST
BOOST Pin Current
ISW = 1A
IRUN/SS
RUN/SS Current
VRUN/SS(THRESH)
RUN/SS Threshold
VFB(PGTHRESH)
VFB PG Threshold
VPG(LOW)
ILPG
500
1
89
78
575
1.1
650
1.2
180
Deg
93
88
%
%
0.4
V
170
2.0
kHz
MHz
2.6
kHz
3.6
210
A
mV
10
µA
1.5
2.5
V
20
30
mA
2.1
µA
0.8
V
VFB Rising
720
mV
PG Voltage Output Low
VFB = 640mV, IPG = 250µA
0.22
0.4
V
PG Pin Leakage
VPG = 2V
0.1
1
µA
0.3
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3506E/LT3506AE are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls. The LT3506I/LT3506AI are
guaranteed and tested over the full –40°C to 125°C operating temperature
range.
Note 3: Current limit is guaranteed by design and/or correlation to static
test. Slope compensation reduces current limit at high duty cycle.
Note 4: Switch VCESAT guaranteed by design.
Note 5: Switching phase is guaranteed by design.
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LT3506/LT3506A
TYPICAL PERFOR A CE CHARACTERISTICS
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Efficiency, VOUT = 1.8V (LT3506A)
100
80
70
60
30
0.2
0.4
0.6 0.8 1.0 1.2
OUTPUT CURRENT (A)
1.4
VIN = 5V
75
VIN = 12V
70
65
VIN = 25V
60
1.6
60
55
1.8
0.8
IOUT (A)
0.4
1.6
1.2
LOAD CURRENT (A)
LOAD CURRENT (A)
L = 1µH
L = 4.7µH
L = 3.3µH
1.4
1.2
L = 2.2µH
1.2
8
10 12
6
INPUT VOLTAGE (V)*
14
1.0
16
5
0
15
20
10
INPUT VOLTAGE (V)*
3506 G04
TA = 25°C
200
100
0
1.0
0.5
1.5
2.0
SW CURRENT (A)
3506 G06
Frequency vs Temperature
700
TA = 25°C
30
FREQUENCY (kHz)
650
2.0
MINIMUM
1.5
1.0
10
1.15
LT3506A
600
1.10
LT3506
550
FREQUENCY (MHz)
CURRENT LIMIT (A)
1.20
TYPICAL
2.5
BOOST CURRENT (mA)
TA = 25°C
300
0
25
Current Limit vs Duty Cycle
3.0
20
1.6
1.2
3506 G05
Boost Pin Current
40
0.8
IOUT (A)
0.4
Switch VCESAT
400
SLOPE COMPENSATION REQUIRES
L > 2.2µH FOR VIN < 7 WITH VOUT = 3.3V
TA = 25°C
1.6
1.4
4
0
3506 G03
SWITCH VOLTAGE (mV)
L = 2.2µH
2
VIN = 25V
65
Maximum Load Current,
VOUT = 3.3V (LT3506A)
TA = 25°C
0
70
3506 G02
L = 1.5µH
1.0
VIN = 15V
75
50
Maximum Load Current,
VOUT = 1.8V (LT3506A)
1.6
80
50
0
VIN = 8V
85
55
3506 G01
1.8
VOUT = 5V
95 L = 10µH (COOPER UP1B-100)
TA = 25°C
90
80
VIN = 4.5V
VIN = 12V
VIN = 25V
0
Efficiency, VOUT = 5V (LT3506)
100
85
50
40
Efficiency, VOUT = 3.3V (LT3506)
VOUT = 3.3V
95 L = 6.4µH (SUMIDA CR54-6R4)
TA = 25°C
90
EFFICIENCY (%)
EFFICIENCY (%)
VOUT = 1.8V
L = 4.7µH (COILCRAFT MSS6122-472MLB)
90 TA = 25°C
EFFICIENCY (%)
100
1.05
0.5
0
0
1.0
1.5
0.5
SWITCH CURRENT (A)
2.0
3506 G07
0
0
20
60
40
DUTY CYCLE (%)
80
100
3506 G08
500
–50 –25
0
25
50
75
TEMPERATURE (°C)
100
1.00
125
3506 G10
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LT3506/LT3506A
TYPICAL PERFOR A CE CHARACTERISTICS
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RUN/SS Thresholds vs
Temperature
3.0
1.4
2.5
1.2
RUNN/SS THRESHOLDS (V)
RUN/SS CURRENT (µA)
IRUN/SS vs Temperature
2.0
1.5
1.0
0.5
1.0
TO SWITCH
0.8
0.6
TO RUN
0.4
0.2
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
3506 G12
0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
3506 G13
PI FU CTIO S
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BOOST1 (Pin 1), BOOST2 (Pin 8): The BOOST pins are
used to provide drive voltages, higher than the input
voltage, to the internal bipolar NPN power switches. Tie
through a diode from VOUT or from VIN.
SW1 (Pin 2), SW2 (Pin 7): The SW pins are the outputs
of the internal power switches. Connect these pins to the
inductors, catch diodes and boost capacitors.
VIN1 (Pins 3, 4): The VIN1 pins supply current to the
LT3506’s internal regulator and to the internal power
switch connected to SW1. These pins must be locally
bypassed.
VIN2 (Pins 5, 6): The VIN2 pins supply current to the internal power switch connected to SW2 and must be locally
bypassed. Connect these pins directly to VIN1 unless power
for Channel 2 is coming from a different source.
RUN/SS1 (Pin 13), RUN/SS2 (Pin 12): The RUN/SS pins
are used to shut down the individual switching regulators and the internal bias circuits. They also provide a
soft-start function. To shut down either regulator, pull the
RUN/SS pin to ground with an open drain or collector.
Tie a capacitor from these pins to ground to limit switch
current during start-up. If neither feature is used, leave
these pins unconnected.
PG1 (Pin 14), PG2 (Pin 11): The Power Good pins are
the open collector outputs of an internal comparator. PG
remains low until the FB pin is within 10% of the final
regulation voltage. As well as indicating output regulation,
the PG pins can be used to sequence the two switching
regulators. These pins can be left unconnected. The PG
outputs are valid when VIN is greater than 3.4V and either
of the RUN/SS pins is high. The PG comparators are
disabled in shutdown.
VC1 (Pin 15), VC2 (Pin 10): The VC pins are the outputs of
the internal error amps. The voltages on these pins control
the peak switch currents. These pins are normally used
to compensate the control loops, but can also be used to
override the loops. Pull these pins to ground with an open
drain to shut down each switching regulator.
FB1 (Pin 16), FB2 (Pin 9): The LT3506 regulates each
feedback pin to 800mV. Connect the feedback resistor
divider taps to these pins.
Exposed Pad (Pin 17): The Exposed Pad of the package
provides both electrical contact to ground and good thermal
contact to the printed circuit board. The Exposed Pad must
be soldered to the circuit board for proper operation.
3506afb
LT3506/LT3506A
BLOCK DIAGRA
W
VIN
RUN/SS2
2µA
INT REG
AND REF
RUN/SS1
MASTER
OSC
CLK1
CLK2
2µA
VIN
VIN
CIN
0.75V
∑
SLOPE
R
C1
S
CLK
BOOST
D2
Q
C3
FOLDBACK
LOGIC
SW
L1
OUT
C1
D1
–
RC
CC
RUN/SS
+
R1
R2
+
CF
ERROR
AMP
FB
–
VC
800mV
ILIMIT
CLAMP
80mV
PG
+
GND
–
3506 F02
Figure 2. Block Diagram of the LT3506 with Associated External Components (1 of 2 Regulators Shown)
3506afb
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OPERATIO
LT3506/LT3506A
(Refer to the Block Diagram)
The LT3506 is a dual, constant frequency, current mode
buck regulator with internal 2A power switches. The two
regulators share common circuitry including voltage
reference and oscillator. In addition, the analog blocks
on both regulators share the VIN1 supply voltage, but are
otherwise independent. This section describes the operation of the LT3506.
If the RUN/SS (run/soft-start) pins are both tied to ground,
the LT3506 is shut down and draws 30μA from VIN1.
Internal 2μA current sources charge external soft-start
capacitors, generating voltage ramps at these pins. If either
RUN/SS pin exceeds 0.6V, the internal bias circuits turn
on, including the internal regulator, 800mV reference and
575kHz master oscillator. In this state, the LT3506 draws
1.8mA from VIN1, whether one or both RUN/SS pins are
high. Neither switching regulator will begin to operate
until its RUN/SS pin reaches ~0.8V. The master oscillator
generates two clock signals of opposite phase.
The two switchers are current mode, step-down regulators.
This means that instead of directly modulating the duty
cycle of the power switch, the feedback loop controls the
peak current in the switch during each cycle. This current mode control improves loop dynamics and provides
cycle-by-cycle current limit.
The Block Diagram in Figure 2 shows only one of the two
switching regulators. A pulse from the slave oscillator
sets the RS flip-flop and turns on the internal NPN bipolar
power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level
determined by the voltage at VC, current comparator C1
resets the flip-flop, turning off the switch. The current in
the inductor flows through the external Schottky diode,
and begins to decrease. The cycle begins again at the next
pulse from the oscillator. In this way the voltage on the VC
pin controls the current through the inductor to the output.
The internal error amplifier regulates the output voltage
by continually adjusting the VC pin voltage.
The threshold for switching on the VC pin is 0.75V, and an
active clamp of 1.9V limits the output current. The VC pin
is also clamped to the RUN/SS pin voltage. As the internal
current source charges the external soft-start capacitor,
the current limit increases slowly. Each switcher contains
an independent oscillator. This slave oscillator is normally
synchronized to the master oscillator. However, during
start-up, short-circuit or overload conditions, the FB pin
voltage will be near zero and an internal comparator gates
the master oscillator clock signal. This allows the slave
oscillator to run the regulator at a lower frequency. This
frequency foldback behavior helps to limit switch current
and power dissipation under fault conditions.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for efficient operation.
A power good comparator trips when the FB pin is at 90%
of its regulated value. The PG output is an open collector
transistor that is off when the output is in regulation, allowing an external resistor to pull the PG pin high. Power
good is valid when the LT3506 is enabled (either RUN/SS
pin is high) and VIN is greater than ~3.4V.
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LT3506/LT3506A
APPLICATIO S I FOR ATIO
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FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
maximum input voltage is ~8V with VOUT=0.8V. Note that
this is a restriction on the operating input voltage; the
circuit will tolerate transient inputs up to the absolute
maximum rating.
R1 = R2(VOUT/0.8 – 1)
Inductor Selection and Maximum Output Current
The parallel combination of R1 and R2 should be 10k or
less to avoid bias current errors. Reference designators
refer to the Block Diagram in Figure 2.
A good first choice for the inductor value is:
Input Voltage Range
where VD is the voltage drop of the catch diode (~0.4V)
and L is in μH. With this value the maximum load current
will be ~1.6A, independent of input voltage. The inductor’s
RMS current rating must be greater than your maximum
load current and its saturation current should be about 30%
higher. To keep efficiency high, the series resistance (DCR)
should be less than 0.1W. Table 1 lists several vendors and
types that are suitable. Of course, such a simple design
guide will not always result in the optimum inductor for
your application. A larger value provides a slightly higher
maximum load current, and will reduce the output voltage ripple. If your load is lower than 1.6A, then you can
decrease the value of the inductor and operate with higher
ripple current. This allows you to use a physically smaller
inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the
simple rule above, then the maximum load current will
depend on input voltage. There are several graphs in the
Typical Performance Characteristics section of this data
sheet that show the maximum load current as a function
of input voltage and inductor value for several popular
output voltages. Also, low inductance may result in discontinuous mode operation, which may be acceptable,
but further reduces maximum load current. For details of
maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50%(VOUT/VIN < 0.5), there
is a minimum inductance required to avoid subharmonic
oscillations. See Application Note 19 for detailed information on subharmonic oscillations. The following discussion
assumes continuous inductor current.
The minimum input voltage is determined by either the
LT3506’s minimum operating voltage of ~3.6V, or by its
maximum duty cycle. The duty cycle is the fraction of
time that the internal switch is on and is determined by
the input and output voltages:
DC = (VOUT + VD)/(VIN – VSW + VD)
where VD is the forward voltage drop of the catch diode
(~0.4V) and VSW is the voltage drop of the internal switch
(~0.3V at maximum load). This leads to a minimum input
voltage of:
VIN(MIN) = (VOUT + VD)/DCMAX - VD + VSW
with DCMAX = 0.89 (0.78 for the LT3506A).
A more detailed analysis includes inductor loss and the
dependence of the diode and switch drop on operating
current. A common application where the maximum duty
cycle limits the input voltage range is the conversion of 5V
to 3.3V. The maximum load current that the LT3506 can
deliver at 3.3V depends on the accuracy of the 5V input
supply. With a low loss inductor (DCR less than 80mW),
the LT3506 can deliver 1.2A for VIN > 4.7V and 1.6A for
VIN > 4.85V. The maximum input voltage is determined
by the absolute maximum ratings of the VIN and BOOST
pins and by the minimum duty cycle DCMIN = 0.08 (0.15
for the LT3506A):
VIN(MAX) = (VOUT + VD)/DCMIN – VD + VSW.
This limits the maximum input voltage to ~21V with VOUT
= 1.2V and ~15V with VOUT = 0.8V. For the LT3506A the
L = 2 • (VOUT + VD) for the LT3506
L = (VOUT + VD) for the LT3506A
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LT3506/LT3506A
APPLICATIO S I FOR ATIO
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The current in the inductor is a triangle wave with an
average value equal to the load current. The peak switch
current is equal to the output current plus half the peak-topeak inductor ripple current. The LT3506 limits its switch
current in order to protect itself and the system from
overload faults. Therefore, the maximum output current
that the LT3506 will deliver depends on the current limit,
the inductor value and the input and output voltages. L
is chosen based on output current requirements, output
voltage ripple requirements, size restrictions and efficiency
goals. When the switch is off, the inductor sees the output
voltage plus the catch diode drop. This gives the peak-topeak ripple current in the inductor:
ΔIL = (1 – DC)(VOUT + VD)/(L • f)
where f is the switching frequency of the LT3506 and L
is the value of the inductor. The peak inductor and switch
current is
ISWPK = ILPK = IOUT + ΔIL/2.
To maintain output regulation, this peak current must be
less than the LT3506’s switch current limit ILIM. ILIM is at
least 2A at low duty cycle and decreases linearly to 1.7A
at DC = 0.8. The maximum output current is a function of
the chosen inductor value:
IOUT(MAX) = ILIM – ΔIL/2 = 2A • (1 – 0.21 • DC) – ΔIL/2
If the inductor value is chosen so that the ripple current
is small, then the available output current will be near
the switch current limit. One approach to choosing the
inductor is to start with the simple rule given above, look
at the available inductors, and choose one to meet cost or
space goals. Then use these equations to check that the
LT3506 will be able to deliver the required output current.
Note again that these equations assume that the inductor
current is continuous. Discontinuous operation occurs
when IOUT is less than ΔIL/2 as calculated above.
Table 1. Inductors
Part Number
Value
(μH)
ISAT (A)
DCR (W)
Height
(mm)
CR43-3R3
3.3
1.44
0.086
3.5
CR43-4R7
4.7
1.15
0.109
3.5
CDC5d23-2R2
2.2
2.16
0.030
2.5
CDRH5D28-2R6
2.6
2.60
0.013
3.0
CDRH6D26-5R6
5.6
2.00
0.027
2.8
CDH113-100
10
2.00
0.047
3.7
DO1606T-152
1.5
2.10
0.060
2.0
DO1606T-222
2.2
1.70
0.070
2.0
DO1608C-332
3.3
2.00
0.080
2.9
DO1608C-472
4.7
1.50
0.090
2.9
DO1813P-682HC
6.8
2.20
0.080
5.0
SD414-2R2
2.2
2.73
0.061
1.35
SD414-6R8
6.8
1.64
0.135
1.35
UP1B-100
10
1.90
0.111
5.0
(D62F)847FY-2R4M
2.4
2.5
0.037
2.7
(D73LF)817FY2R2M
2.2
2.7
0.03
3.0
Sumida
Coilcraft
Cooper
Toko
Input Capacitor Selection
Bypass the input of the LT3506 circuit with a 4.7μF or
higher ceramic capacitor of X7R or X5R type. A lower
value or a less expensive Y5V type can be used if there is
additional bypassing provided by bulk electrolytic or tantalum capacitors. The following paragraphs describe the
input capacitor considerations in more detail. Step-down
regulators draw current from the input supply in pulses
with very fast rise and fall times. The input capacitor is
required to reduce the resulting voltage ripple at the LT3506
and to force this very high frequency switching current
into a tight local loop, minimizing EMI. The input capaci-
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tor must have low impedance at the switching frequency
to do this effectively, and it must have an adequate ripple
current rating. With two switchers operating at the same
frequency but with different phases and duty cycles, calculating the input capacitor RMS current is not simple.
However, a conservative value is the RMS input current for
the channel that is delivering most power (VOUT • IOUT).
This is given by:
IINRMS = IOUT
VOUT • ( VIN − VOUT )
VIN
<
IOUT
2
and is largest when VIN = 2VOUT (50% duty cycle). As
the second, lower power channel draws input current,
the input capacitor’s RMS current actually decreases as
the out-of-phase current cancels the current drawn by
the higher power channel. Considering that the maximum
load current from a single channel is ~1.6A, RMS ripple
current will always be less than 0.8A.
The high frequency of the LT3506 reduces the energy
storage requirements of the input capacitor, so that the
capacitance required is less than 22μF (less than 10μF
for the LT3506A). The combination of small size and low
impedance (low equivalent series resistance or ESR) of
ceramic capacitors makes them the preferred choice.
The low ESR results in very low voltage ripple and the
capacitors can handle plenty of ripple current. They are also
comparatively robust and can be used in this application
at their rated voltage. X5R and X7R types are stable over
temperature and applied voltage, and give dependable
service. Other types (Y5V and Z5U) have very large temperature and voltage coefficients of capacitance, so they
may have only a small fraction of their nominal capacitance
in your application. While they will still handle the RMS
ripple current, the input voltage ripple may become fairly
large, and the ripple current may end up flowing from
your input supply or from other bypass capacitors in your
system, as opposed to being fully sourced from the local
input capacitor.
An alternative to a high value ceramic capacitor is a lower
value along with a larger electrolytic capacitor, for example
a 1μF ceramic capacitor in parallel with a low ESR tantalum
capacitor. For the electrolytic capacitor, a value larger than
22mF (10mF for the LT3506A) will be required to meet the
10
ESR and ripple current requirements. Because the input
capacitor is likely to see high surge currents when the input
source is applied, tantalum capacitors should be surge
rated. The manufacturer may also recommend operation
below the rated voltage of the capacitor. Be sure to place
the 1μF ceramic as close as possible to the VIN and GND
pins on the IC for optimal noise immunity.
A final caution is in order regarding the use of ceramic
capacitors at the input. A ceramic input capacitor can
combine with stray inductance to form a resonant tank
circuit. If power is applied quickly (for example by plugging the circuit into a live power source) this tank can ring,
doubling the input voltage and damaging the LT3506. The
solution is to either clamp the input voltage or dampen the
tank circuit by adding a lossy capacitor in parallel with the
ceramic capacitor. For details, see Application Note 88.
Output Capacitor Selection
The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores
energy in order satisfy transient loads and to stabilize the
LT3506’s control loop. Because the LT3506 operates at a
high frequency, you don’t need much output capacitance.
Also, the current mode control loop doesn’t require the
presence of output capacitor series resistance (ESR). For
these reasons, you are free to use ceramic capacitors to
achieve very low output ripple and small circuit size.
Estimate output ripple with the following equations:
VRIPPLE = ΔIL/(8 • f • COUT) for ceramic capacitors, and
VRIPPLE = ΔIL • ESR for electrolytic capacitors (tantalum
and aluminum);
where ΔIL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low, and the
RMS current rating of the output capacitor is usually not
of concern.
Another constraint on the output capacitor is that it
must have greater energy storage than the inductor; if the stored energy in the inductor is transferred
to the output, you would like the resulting voltage
step to be small compared to the regulation voltage. For a 5% overshoot, this requirement becomes
COUT > 10L(ILIM/VOUT)2.
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Finally, there must be enough capacitance for good transient
performance. The last equation gives a good starting point.
Alternatively, you can start with one of the designs in this
data sheet and experiment to get the desired performance.
This topic is covered more thoroughly in the section on
loop compensation.
For 5V and 3.3V outputs with greater than 1A output, a
22μF 6.3V ceramic capacitor (X5R or X7R) at the output
results in very low output voltage ripple and good transient response. For lower voltages, 22μF is adequate but
increasing COUT will improve transient performance. For
the LT3506A, 10μF of output capacitance is sufficient at
VOUT between 3.3V and 5V. Other types and values can be
used. The following discusses tradeoffs in output ripple
and transient performance.
The high performance (low ESR), small size and robustness of ceramic capacitors make them the preferred type
for LT3506 applications. However, all ceramic capacitors
are not the same. As mentioned above, many of the
higher value capacitors use poor dielectrics with high
temperature and voltage coefficients. In particular, Y5V
and Z5U types lose a large fraction of their capacitance
with applied voltage and temperature extremes. Because
the loop stability and transient response depend on the
value of COUT, you may not be able to tolerate this loss.
Use X7R and X5R types.
You can also use electrolytic capacitors. The ESRs of most
aluminum electrolytics are too large to deliver low output
ripple. Tantalum and newer, lower ESR organic electrolytic
capacitors intended for power supply use are suitable,
and the manufacturers will specify the ESR. The choice of
capacitor value will be based on the ESR required for low
ripple. Because the volume of the capacitor determines
its ESR, both the size and the value will be larger than a
ceramic capacitor that would give similar ripple performance. One benefit is that the larger capacitance may give
better transient response for large changes in load current.
Table 2 lists several capacitor vendors.
Table 2. Low-ESR Surface Mount Capacitors
VENDOR
TYPE
SERIES
Taiyo-Yuden
Ceramic
AVX
Ceramic
Tantalum
Kemet
Tantalum
Tantalum
Organic
Aluminum
Organic
T491, T494, T495, T520
Sanyo
Tantalum or Aluminum
Organic
POSCAP
Panasonic
Aluminum
Organic
SP
CAP
TDK
Ceramic
TPS
A700
Catch Diode
The catch diode (D1 in Figure 2) must have a reverse voltage rating greater than the maximum input voltage. The
average current of the catch diode is given by:
IDAVE=IOUT(1-DCMIN)
A Schottky diode with a 1A average forward current rating
will suffice for most applications. The ON Semiconductor
MBRM120LT3 (20V) and MBRM130LT3 (30V) are good
choices; they have a tiny package with good thermal properties. Many vendors have suitable surface mount versions of
the 1N5817 (20V) and 1N5818 (30V) 1A Schottky diodes
such as the Microsemi UPS120.
Applications with large step down ratios and high output
currents may have more than 1A of average diode current.
The ON Semiconductor MBRS230LT3 or International Rectifier 20BQ030 (both 2A, 30V) would be good choices.
BOOST Pin Considerations
The capacitor and diode tied to the BOOST pin generate
a voltage that is higher than the input voltage. In most
cases a 0.1μF capacitor and fast switching diode (such
as the CMDSH-3 or FMMD914) will work well. Figure 3
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shows three ways to arrange the boost circuit. The BOOST
pin must be more than 2.5V above the SW pin for full
efficiency. For outputs of 3.3V and higher the standard
circuit (Figure 3a) is best. For outputs between 2.8V and
3.3V, use a small Schottky diode (such as the BAT-54).
For lower output voltages the boost diode can be tied to
the input (Figure 3b). The circuit in Figure 3a is more efficient because the BOOST pin current comes from a lower
voltage source. Finally, as shown in Figure 3c, the anode
of the boost diode can be tied to another source that is
at least 3V. For example, if you are generating 3.3V and
1.8V and the 3.3V is on whenever the 1.8V is on, the 1.8V
boost diode can be connected to the 3.3V output. In any
case, you must also be sure that the maximum voltage at
the BOOST pin is less than the maximum specified in the
Absolute Maximum Ratings section.
The boost circuit can also run directly from a DC voltage
that is higher than the input voltage by more than 3V,
as in Figure 3d. The diode is used to prevent damage to
the LT3506 in case VINB is held low while VIN is present.
The circuit saves several components (both BOOST pins
can be tied to D2). However, efficiency may be lower and
dissipation in the LT3506 may be higher. Also, if VINB is
absent, the LT3506 will still attempt to regulate the output,
but will do so with very low efficiency and high dissipation
because the switch will not be able to saturate, dropping
1.5V to 2V in conduction.
D2
D2
C3
BOOST
VIN
VIN
VOUT
SW
VIN
VIN
SW
VBOOST – VSW ≅ VIN
MAX VBOOST ≅ 2VIN
VBOOST – VSW ≅ VOUT
MAX VBOOST ≅ VIN + VOUT
(3b)
(3a)
D2
D2
VINB
>VIN + 3V
VINB > 3V
BOOST
BOOST
C3
LT3506
VIN
VOUT
GND
GND
VIN
C3
BOOST
LT3506
LT3506
LT3506
SW
VOUT
VIN
VIN
GND
SW
VOUT
GND
VBOOST – VSW ≅ VINB
MAX VBOOST ≅ VINB + VIN
MINIMUM VALUE FOR VINB = 3V
MAX VBOOST – VSW ≅ VINB
MAX VBOOST ≅ VINB
MINIMUM VALUE FOR VINB = VIN + 3V
(3c)
3506 F03
(3d)
Figure 3. Generating the Boost Voltage
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The minimum input voltage of an LT3506 application is
limited by the minimum operating voltage (<3.6V) and by
the maximum duty cycle as outlined above. For proper
start-up, the minimum input voltage is also limited by
the boost circuit. If the input voltage is ramped slowly,
or the LT3506 is turned on with its RUN/SS pin when the
output is already in regulation, then the boost capacitor
may not be fully charged. Because the boost capacitor is
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load generally goes to
zero once the circuit has started. The plots below show
the minimum load current to start and to run as a function
of input voltage for 3.3V and 5V outputs. In many cases
the discharged output capacitor will present a load to the
switcher which will allow it to start. The plots show the
worst-case situation where VIN is ramping very slowly.
Use a Schottky diode (such as the BAT-54) for the lowest
start-up voltage.
Minimum Input Voltage,
VOUT = 3.3V (LT3506A)
5.5
VIN TO START
4.5
4.0
VIN TO RUN
3.0
0.001
TA = 25°C
DBOOST = 1N5817
VIN TO START
6.5
BOOST DIODE
TIED TO OUTPUT
3.5
7.0
TA = 25°C
DBOOST = 1N5817
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
5.0
Minimum Input Voltage,
VOUT = 5V (LT3506A)
BOOST DIODE
TIED TO OUTPUT
6.0
5.5
VIN TO RUN
5.0
BOOST DIODE
TIED TO INPUT
0.01
0.1
ILOAD (A)
1
3506 G14
4.5
0.001
BOOST DIODE
TIED TO INPUT
0.01
0.1
1
ILOAD (A)
3506 G15
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Frequency Compensation
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data sheet
that is similar to your application and tune the compensation network to optimize the performance. Stability should
then be checked across all operating conditions, including
load current, input voltage and temperature. The LT1375
data sheet contains a more thorough discussion of loop
compensation and describes how to test the stability using
a transient load. Figure 4 shows an equivalent circuit for the
LT3506 control loop. The error amp is a transconductance
amplifier with finite output impedance. The power section,
consisting of the modulator, power switch and inductor,
is modeled as a transconductance amplifier generating an
output current proportional to the voltage at the VC pin.
Note that the output capacitor integrates this current, and
that the capacitor on the VC pin (CC) integrates the error
amplifier output current, resulting in two poles in the loop.
In most cases a zero is required and comes from either the
output capacitor ESR or from a resistor in series with CC.
This simple model works well as long as the value of the
inductor is not too high and the loop crossover frequency
is much lower than the switching frequency. A phase lead
capacitor (CPL) across the feedback divider may improve
the transient response.
330umhos
1MΩ
GND
+
Frequency compensation is provided by the components
tied to the VC pin. Generally a capacitor and a resistor in
series to ground determine loop gain. In addition, there
is a lower value capacitor in parallel. This capacitor is not
part of the loop compensation but is used to filter noise
at the switching frequency.
VSW
ERROR
AMPLIFIER
OUTPUT
R1
ESR
VFB
800mV
VC
RC
CPL
FB
–
The LT3506 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3506 does not require the ESR of the output capacitor
for stability so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size.
LT3506
CURRENT MODE
POWER STAGE
gmp
2.4A/V
C1
R2
CF
C1
+
POLYMER
OR
TANTALUM
CERAMIC
CC
3506 F04
Figure 4. Circuit Model for Frequency Compensation
Soft-Start and Shutdown
The RUN/SS (Run/Soft-Start) pins are used to place the
individual switching regulators and the internal bias circuits
in shutdown mode. They also provide a soft-start function.
To shut down either regulator, pull the RUN/SS pin to ground
with an open-drain or collector. If both RUN/SS pins are
pulled to ground, the LT3506 enters its shutdown mode
with both regulators off and quiescent current reduced to
~30μA. Internal 2μA current sources pull up on each pin.
If either pin reaches ~0.6V, the internal bias circuits start
and the quiescent current increases to ~3.5mA.
If a capacitor is tied from the RUN/SS pin to ground, then
the internal pull-up current will generate a voltage ramp on
this pin. This voltage clamps the VC pin, limiting the peak
switch current and therefore input current during start-up.
A good value for the soft-start capacitor is COUT/10,000,
where COUT is the value of the output capacitor.
The RUN/SS pins can be left floating if the shutdown
feature is not used. They can also be tied together with a
single capacitor providing soft-start. The internal current
sources will charge these pins to ~2.5V.
The RUN/SS pins provide a soft-start function that limits
peak input current to the circuit during start-up. This helps
to avoid drawing more current than the input source can
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supply or glitching the input supply when the LT3506 is
enabled. The RUN/SS pins do not provide an accurate
delay to start or an accurately controlled ramp at the output
voltage, both of which depend on the output capacitance
and the load current. However, the power good indicators
can be used to sequence the two outputs, as described
below.
Power Good Indicators
The PG pin is the open collector output of an internal
comparator. PG remains low until the FB pin is within
10% of the final regulation voltage. Tie the PG pin to any
supply with a pull-up resistor that will supply less than
250μA. Note that this pin will be open when the LT3506 is
placed in shutdown mode (both RUN/SS pins at ground)
regardless of the voltage at the FB pin. Power good is valid
when the LT3506 is enabled (either RUN/SS pin is high)
and VIN is greater than ~2.4V.
Output Sequencing
The PG and RUN/SS pins can be used to sequence the
two outputs. Figure 5 shows several circuits to do this. In
each case channel 1 starts first. Note that these circuits
sequence the outputs during start-up. When shut down
the two channels turn off simultaneously. In Figure 5a, a
larger capacitor on RUN/SS2 delays channel 2 with respect
to channel 1. The soft-start capacitor on RUN/SS2 should
be at least twice the value of the capacitor on RUN/SS1.
A larger ratio may be required, depending on the output
capacitance and load on each channel. Make sure to test
the circuit in the system before deciding on final values
for these capacitors. The circuit in Figure 5b requires the
fewest components, with both channels sharing a single
soft-start capacitor. The power good comparator of channel 1 disables channel 2 until output 1 is in regulation. For
independent control of channel 2, use the circuit in Figure
5c. The capacitor on RUN/SS1 is smaller than the capaciRUN/SS1
RUN/SS1
OFF ON
1nF
LT3506
OFF ON
1nF
VC2
LT3506
RUN/SS2
GND
RUN/SS2 PG1
GND
2.2nF
(5a) Channel 2 is Delayed
(5b) Fewest Components
RUN/SS1
OFF ON
OFF2 ON2
1nF
LT3506
PG1
RUN/SS1
OFF ON
1nF
RUN/SS2
GND
1.5nF
RUN/SS2
GND
1.5nF
(5c) Independent Control of Channel 2
LT3506
PG1
3506 F05
(5d) Doesn't Work !
Figure 5. Sequencing the Outputs
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tor on RUN/SS2. This allows the LT3506 to start up and
enable its power good comparator before RUN/SS2 gets
high enough to allow channel 2 to start switching. Channel
2 only operates when it is enabled with the external control
signals and output 1 is in regulation. The circuit in Figure
5a leaves both power good indicates free. However, the
circuits in Figures 5b and 5c have another advantage. As
well as sequencing the two outputs at start-up, they also
disable channel 2 if output 1 falls out of regulation (due
to a short circuit or a collapsing input voltage).
PARASITIC DIODE
D4
SW
VOUT
LT3506
3506 F06
Figure 6. Shorted Input Protection
and needs to generate 12V and 2.5V, it would be more
efficient to generate the 2.5V output from the 5V supply
and the 12V output from the 18V supply. The LT3506 can
step down 18V to 2.5V, but the efficiency would be lower
than stepping down from 5V to 2.5V.
Finally, be aware that the circuit in Figure 5d does not
work, because the power good comparators are disabled in
shutdown. When the system is placed in shutdown mode
by pulling down on RUN/SS1, then output 1 will go low,
PG1 will pull down on RUN/SS2, and the LT3506 will enter
its low current shutdown state. This disables PG1, and
RUN/SS2 ramps up again to enable the LT3506. The circuit
will oscillate and pull extra current from the input.
This feature can also be used when the maximum stepdown ratio is exceeded. In this case, VIN2 can be tied to
VOUT1 for applications requiring high VIN to VOUT ratios. A
dual step-down application steps down the input voltage
(VIN1) to the highest output voltage then uses that voltage
to power the second channel (VIN2). VOUT1 must be able
to provide enough current for its output plus the average
current drawn from VOUT2. Note that the VOUT1 must be
above minimum input voltage for VIN2 when the second
channel starts to switch. Delaying the second channel can
be accomplished by either using independent soft-start
capacitors or sequencing with the PG1 output. The Two
Stage Step-Down circuit in the Applications section shows
an example of the latter approach.
Multiple Input Supplies
The internal supplies of the LT3506 operate from VIN1. It is
possible to supply VIN2 from a different source, provided
VIN1 is above the minimum supply level whenever VIN2 is
present. This could be used when a system has two primary supplies available. It is more efficient to generate the
desired outputs with the lowest step-down ratio possible.
For example, if a system has 18V and 5V power available
VIN
VIN
VIN
VIN
SW
GND
SW
GND
(7a)
VIN
IC1
(7b)
VSW
C1
L1
SW
D1
GND
C2
3506 F07
(7c)
Figure 7. Subtracting the Current when the Switch is ON (a) From the Current when the Switch in OFF (b) Reveals the Path
of the High Frequency Switching current (c) Keep This Loop Small. The Voltage on the SW and BOOST Nodes will also be
Switched; Keep these Nodes as Small as Possible. Finally, Make Sure the Circuit is Shielded with a Local Ground Plane.
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Shorted Input Protection
If the inductor is chosen so that it won’t saturate excessively, the LT3506 will tolerate a shorted output. There is
another situation to consider in systems where the output
will be held high when the input to the LT3506 is absent.
If the VIN and one of the RUN/SS pins are allowed to float,
then the LT3506’s internal circuitry will pull its quiescent
current through its SW pin. This is fine if your system can
tolerate a few mA of load in this state. With both RUN/SS
pins grounded, the LT3506 enters shutdown mode and the
SW pin current drops to ~30μA. However, if the VIN pin
is grounded while the output is held high, then parasitic
diodes inside the LT3506 can pull large currents from the
PIN 1
TOP MARK
VOUT1
VIA TO LOCAL GROUND PLANE
VIA TO VIN
output through the SW pin and the VIN pin. A Schottky
diode in series with the input to the LT3506 will protect
the LT3506 and the system from a shorted or reversed
input, as shown in Figure 6.
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board (PCB) layout. Figure 7
shows the high-di/dt paths in the buck regulator circuit.
Note that large, switched currents flow in the power switch,
the catch diode and the input capacitor. The loop formed by
these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components,
and tie this ground plane to system ground at one location,
ideally at the ground terminal of the output capacitor C2.
Additionally, the SW and BOOST nodes should be kept as
small as possible. Figure 8 shows recommended component placement with trace and via locations.
Thermal Considerations
GND
VOUT2
3506 F08
Figure 8. A Good PCB Layout Ensures Proper Low EMI Operation
The PCB must also provide heat sinking to keep the LT3506
cool. The exposed metal on the bottom of the package
must be soldered to a ground plane. This ground should
be tied to other copper layers below with thermal vias;
these layers will spread the heat dissipated by the LT3506.
Place additional vias near the catch diodes. Adding more
copper to the top and bottom layers and tying this copper to the internal planes with vias can reduce thermal
resistance further. With these steps, the thermal resistance from die (or junction) to ambient can be reduced to
qJA = 43°C/W.
The power dissipation in the other power components—
catch diodes, boost diodes and inductors, cause additional
copper heating and can further increase what the IC sees
as ambient temperature. See the LT1767 data sheet’s
Thermal Considerations section.
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Single, Low-Ripple 3.2A Output
The LT3506 can generate a single, low-ripple 3.2A output
if the outputs of the two switching regulators are tied
together and share a single output capacitor. By tying the
two FB pins together and the two VC pins together, the
two channels will share the load current. There are several
advantages to this two-phase buck regulator. Ripple currents at the input and output are reduced, reducing voltage ripple and allowing the use of smaller, less expensive
capacitors. Although two inductors are required, each will
be smaller than the inductor required for a single-phase
regulator. This may be important when there are tight
height restrictions on the circuit. The Typical Applications
section shows circuits with maximum heights of 1.4mm,
1.8mm and 2.1mm.
There is one special consideration regarding the two phase
circuit. When the difference between the input voltage and
output voltage is less than 2.5V, then the boost circuits may
prevent the two channels from properly sharing current.
If, for example, channel 1 gets started first, it can supply
the load current, while channel 2 never switches enough
current to get its boost capacitor charged. In this case,
channel 1 will supply the load until it reaches current limit,
the output voltage drops, and channel 2 gets started. The
solution is to generate a boost supply generated from
either SW pin that will service both BOOST pins. The low
profile, single output 5V to 3.3V converter shown in the
Typical Applications section shows how to do this.
Other Linear Technology Publications
Application notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design note 100 shows
how to generate a dual (+ and –) output supply using a
buck regulator
3506afb
18
LT3506/LT3506A
TYPICAL APPLICATIO S
U
1.8V and 1.2V Outputs with Sequencing
VIN
4.5V TO 21V
D3a
D3b
22µF
VIN1 VIN2
BOOST1
L1
4.7µH
VOUT1
1.8V
1.5A
LT3506
0.22µF
18.7k
1500pF
D1
47µF
15k
BOOST2
L2
3.3µH
0.22µF
SW1
SW2
FB1
FB2
VC1
VC2
VOUT2
1.2V
1.5A
16.2k
1000pF
D2
15k
20k
68µF
32.4k
RUN/SS1 PGOOD1
4.7nF
RUN/SS2
100k
3506 TA01
PGOOD2
GND
PGOOD
D1, D2: ON SEMICONDUCTOR MBRS230LT3
D3: BAT-54A
L1: COILCRAFT MSS6122-472
L2: TDK SLF7028-3R3M
OUTPUT CURRENTS CAN INCREASE TO 1.6A WHEN VIN>12V.
1.8V and ±5V Outputs
VOUT2
VIN
7V TO 25V
D3a
22µF
L1
4.7µH
D1
22µF
BOOST2
LT3506
0.22µF
1500pF
15k
47k
VIN1 VIN2
18.7k
47µF
D4
D3b
BOOST1
VOUT1
1.8V
1.5A
VOUT3
–5V
0.3A
0.22µF
SW1
SW2
FB1
FB2
VC1
VC2
2.2µF
L2
4.7µH
VOUT2
5V
0.6A
69.8k
1500pF
D2
15k
15k
22µF
13.3k
RUN/SS1
4.7nF
100k
PGOOD2 RUN/SS2
PGOOD1
PGOOD
D1: ON SEMICONDUCTOR MBRS230LT3
D2: ON SEMICONDUCTOR MBRM130LT3
D3: BAT-54A
D4: ON SEMICONDUCTOR MBR0530
3506 TA02
2.2nF
GND
L1: COILCRAFT MSS6122-472
L2: COILTRONICS CTX5-1A
IOUT3 SHOULD NEVER EXCEED 1/2 OF IOUT2.
SEE DESIGN NOTE 100 FOR DETAILS ON
GENERATING DUAL OUTPUTS USING A BUCK
REGULATOR.
3506afb
19
LT3506/LT3506A
TYPICAL APPLICATIO S
U
Low Ripple, Low Profile 1.2V, 3A Converter, Maximum Height = 2mm
VIN
4.5V TO 21V
D3a
22µF
D3b
VIN1 VIN2
BOOST1
LT3506
20k
1000pF
VOUT
L1
4.1µH
0.22µF
VC1
SW1
VC2
BOOST2
D1
L2
4.1µH
0.22µF
4.7nF
RUN/SS1
100k
SW2
D2
RUN/SS2
PGOOD
VOUT2
1.2V
3A
PGOOD1
FB1
PGOOD2
FB2
16.2k
GND
32.4k
D1, D2: DIODES, INC. B230A
D3: BAT-54A
L1, L2: SUMIDA CDRH5D18-4R1
68µF
3506 TA03
Two Stage Step Down, Up to 25V Input to 1.2V Output
VIN
8V TO 25V
D3a
D3b
22µF
VIN1 VIN2
BOOST1
VOUT1
5V
1A
L1
10µH
LT3506
0.22µF
69.8k
1500pF
D1
47µF
13.3k
BOOST2
L2
2.2µH
0.22µF
SW1
SW2
FB1
FB2
VC1
VC2
VOUT2
1.2V
1.5A
16.2k
1000pF
15k
D2
20k
32.4k
68µF
100k
RUN/SS1 PGOOD1
4.7nF
RUN/SS2
PGOOD2
GND
D1, D2: ON SEMICONDUCTOR MBRS230LT3
D3: BAT-54A
3506 TA04
PGOOD
L1: COOPER UP1B-100
L2: COOPER UP0.4C-2R2
3506afb
20
LT3506/LT3506A
TYPICAL APPLICATIO S
U
Low Ripple, Low Profile 0.8V, 3A Converter, Maximum Height = 1mm
VIN
3.6V TO 8V
D3a
22µF
D3b
VIN1 VIN2
BOOST1
LT3506AEDHD
20k
1000pF
VOUT
L1
1.5µH
0.1µF
VC1
SW1
VC2
BOOST2
D1
L2
1.5µH
0.1µF
4.7nF
RUN/SS1
100k
SW2
D2
RUN/SS2
PGOOD
VOUT
0.8V
3A
PGOOD1
FB1
PGOOD2
FB2
10k
GND
68µF
D1, D2: DIODES, INC. DFLS230L
D3: BAT-54AT
L1, L2: COILCRAFT LPO6610-152ML
3506 TA05
Low Profile 1.8V and 1.3V Outputs with Sequencing, Maximum Height = 1.2mm
VIN
4.5V TO 10V
D3a
10µF
D3b
VIN1 VIN2
BOOST1
L1
2.2µH
VOUT1
1.8V
1.5A
LT3506AEDHD
0.1µF
18.7k
1.5nF
D1
22µF
BOOST2
15k
SW1
SW2
FB1
FB2
VC1
VC2
10k
L2
2.2µH
0.1µF
VOUT2
1.3V
1.6A
17.4k
1.5nF
D2
15k
47µF
28k
RUN/SS1 PGOOD1
4.7nF
100k
RUN/SS2
3506 TA06
PGOOD2
PGOOD
GND
D1, D2: DIODES, INC. DFLS230L
D3: BAT-54AW
L1, L2: COILCRAFT LPS4012-222
3506afb
21
LT3506/LT3506A
Package Description
DHD Package
16-Lead Plastic DFN (5mm × 4mm)
(Reference LTC DWG # 05-08-1707)
0.70 ±0.05
4.50 ±0.05
3.10 ±0.05
2.44 ±0.05
(2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
4.34 ±0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
5.00 ±0.10
(2 SIDES)
R = 0.20
TYP
4.00 ±0.10
(2 SIDES)
9
R = 0.115
TYP
0.40 ± 0.10
16
2.44 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(SEE NOTE 6)
PIN 1
NOTCH
8
0.200 REF
1
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
0.00 – 0.05
(DHD16) DFN 0504
4.34 ±0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WJGD-2) IN JEDEC
PACKAGE OUTLINE MO-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3506afb
22
LT3506/LT3506A
Package Description
FE Package
16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BA
4.90 – 5.10*
(.193 – .201)
2.74
(.108)
2.74
(.108)
16 1514 13 12 1110
6.60 ±0.10
4.50 ±0.10
9
2.74
(.108)
2.74 6.40
(.108) (.252)
BSC
SEE NOTE 4
0.45 ±0.05
1.05 ±0.10
0.65 BSC
1 2 3 4 5 6 7 8
RECOMMENDED SOLDER PAD LAYOUT
4.30 – 4.50*
(.169 – .177)
0.09 – 0.20
(.0035 – .0079)
0.50 – 0.75
(.020 – .030)
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
3. DRAWING NOT TO SCALE
0.25
REF
1.10
(.0433)
MAX
0° – 8°
0.65
(.0256)
BSC
0.195 – 0.30
(.0077 – .0118)
TYP
0.05 – 0.15
(.002 – .006)
FE16 (BA) TSSOP 0204
4. RECOMMENDED MINIMUM PCB METAL SIZE
FOR EXPOSED PAD ATTACHMENT
*DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.150mm (.006") PER SIDE
3506afb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3506/LT3506A
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LT1765
25V, 2.75A (IOUT), 1.25MHz, High
Efficiency Step-Down DC/DC Converter
VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, S8, TSSOP16E Packages
LT1766
60V, 1.2A (IOUT), 200kHz, High
Efficiency Step-Down DC/DC Converter
VIN: 5.5V to 60V, VOUT(MIN) = 1.2V, IQ = 2.5mA, TSSOP16/TSSOP16E Packages
LT1767
25V, 1.2A (IOUT), 1.25MHz, High
Efficiency Step-Down DC/DC Converter
VIN: 3V to 25V, VOUT(MIN) = 1.2V, IQ = 1mA, MS8, MS8E Packages
LT1940/LT1940L
Dual Monolithic 1.4A, 1.1MHz StepDown Switching Regulator
VIN: 3.6V to 25V, VOUT(MIN) = 1.25V, IQ = 3.8mA, TSSOP16E Packages
LTC3407/LTC3407-2
Dual 600mA/800mA, 1.5MHz,
Synchronous Step-Down Regulator
VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40mA, MSE Package
LT3493
1.2A, 750kHz Step-Down Switching
Regulator in 2mm × 3mm DFN
VIN: 3.6V to 36V, VOUT(MIN) = 0.78V, IQ = 1.9mA, 2mm × 3mm DFN Package
LT3505
1.2A, 3MHz Step-Down Switching
Regulator in 3mm × 3mm DFN
VIN: 3.6V to 36V, VOUT(MIN) = 0.8V, IQ = 2mA, DFN or MSE10 Package
LTC3548
Dual 800mA and 400mA, 2.25MHz,
Synchronous Step-Down Regulator
VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, 3mm × 3mm DFN or MSE10 Package
LTC3549
Dual 300mA, 2.25MHz, Synchronous
Step-Down Regulator
VIN: 2.5V to 5.5V, VOUT(MIN) = 0.6V, IQ = 40µA, 3mm × 3mm DFN Package
LTC3701
Two Phase, Dual, 500kHz, Constant
Frequency, Current Mode, High
Efficiency Step-Down DC/DC Controller
VIN: 2.5V to 10V, VOUT(MIN) = 0.8V, IQ = 460µA, SSOP-16 Package
LTC3736
Dual Two Phase, No RSENSE™,
Synchronous Controller with Output
Tracking
VIN: 2.75V to 9.8V, VOUT(MIN) = 0.6V, IQ = 300µA, 4mm × 4mm QFN or SSOP-24
Packages
LTC3737
Dual Two Phase, No RSENSE DC/DC
Controller with Output Tracking
VIN: 2.75V to 9.8V, VOUT(MIN) = 0.6V, IQ = 220µA, 4mm × 4mm QFN or SSOP-24
Packages
No RSENSE is a trademark of Linear Technology Corporation.
3506afb
24 Linear Technology Corporation
LT 0807 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
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 LINEAR TECHNOLOGY CORPORATION 2006
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