LINER LTC1772HS6 Constant frequency current mode step-down dc/dc controller in sot-23 Datasheet

LTC1772
Constant Frequency
Current Mode Step-Down
DC/DC Controller in SOT-23
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DESCRIPTIO
FEATURES
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The LTC®1772 is a constant frequency current mode stepdown DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltage lockout feature that shuts down the LTC1772
when the input voltage falls below 2.0V.
High Efficiency: Up to 94%
High Output Currents Easily Achieved
Wide VIN Range: 2.5V to 9.8V
Constant Frequency 550kHz Operation
Burst Mode® Operation at Light Load
Low Dropout: 100% Duty Cycle
Tiny 6-Lead SOT-23 Package
0.8V Reference Allows Low Output Voltages
Current Mode Operation for Excellent Line and Load
Transient Response
Low Quiescent Current: 270µA
Shutdown Mode Draws Only 8µA Supply Current
±2.5% Reference Accuracy
The LTC1772 provides a ±2.5% output voltage accuracy
and consumes only 270µA of quiescent current. For
applications where efficiency is a prime consideration, the
LTC1772 is configured for Burst Mode operation, which
enhances efficiency at low output current.
To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously in
dropout (100%dutycycle).In shutdown, the device draws
a mere 8µA. High constant operating frequency of 550kHz
allows the use of a small external inductor.
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APPLICATIO S
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One or Two Lithium-Ion-Powered Applications
Cellular Telephones
Wireless Modems
Portable Computers
Distributed 3.3V, 2.5V or 1.8V Power Systems
Scanners
The LTC1772 is available in a small footprint 6-lead
SOT-23.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
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TYPICAL APPLICATIO
C1
10µF
10V
1
ITH/RUN PGATE
6
2
220pF
3
GND
VFB
VIN
SENSE –
C1: TAIYO YUDEN LMK325BJ106K-T
C2A: SANYO 6TPA47M
C2B: AVX 0805ZC105KAT1A
D1: MOTOROLA MBRM120T3
L1: MURATA LQN6C-4R7
M1: FAIRCHILD FDC638P
R1: IRC LRC-LR1206-01-R030F
5
4
D1
+
C2A
47µF
6V
VIN = 3.3V
90
L1
M1 4.7µH
LTC1772
10k
100
C2B
1µF
10V
VOUT
2.5V
2A
174k
80.6k
EFFICIENCY (%)
R1
0.03Ω
Efficiency vs Load Current
VIN
2.5V
TO 9.8V
VIN = 4.2V
80
VIN = 6V
70
VIN = 9.8V
VIN = 8.4V
60
50
VOUT = 2.5V
40
1772 F01a
1
10
100
1000
LOAD CURRENT (mA)
10000
1772 F01b
Figure 1. High Efficiency, High Output Current 2.5V/2A Regulator
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LTC1772
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PACKAGE/ORDER INFORMATION
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(Note 1)
Input Supply Voltage (VIN).........................– 0.3V to 10V
SENSE–, PGATE Voltages ............. – 0.3V to (VIN + 0.3V)
VFB, ITH /RUN Voltages .............................– 0.3V to 2.4V
PGATE Peak Output Current (<10µs) ......................... 1A
Storage Ambient Temperature Range ... – 65°C to 150°C
Operating Temperature Range
LTC1772CS6 ........................................... 0°C to 70°C
LTC1772ES6 (Note 2) ........................ – 40°C to 85°C
LTC1772IS6 (Note 2) ......................... – 40°C to 85°C
LTC1772HS6 (Notes 2,3) ................. – 40°C to 140°C
Junction Temperature (Note 3) ............................. 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
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ABSOLUTE MAXIMUM RATINGS
TOP VIEW
ITH/RUN 1
GND 2
VFB 3
6 PGATE
5 VIN
4 SENSE –
ORDER PART NUMBER
LTC1772CS6
LTC1772ES6
LTC1772IS6
LTC1772HS6
S6 PART MARKING
S6 PACKAGE
6-LEAD PLASTIC SOT-23
θJA = 230°C/ W
LTIL
LTIM
LTB7
LTBRY
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
Input DC Supply Current
Normal Operation
Sleep Mode
Shutdown
UVLO
Typicals at VIN = 4.2V (Note 4)
2.4V ≤ VIN ≤ 9.8V
2.4V ≤ VIN ≤ 9.8V
2.4V ≤ VIN ≤ 9.8V, VITH /RUN = 0V
VIN < UVLO Threshold
MIN
TYP
MAX
UNITS
270
230
8
6
420
370
22
10
µA
µA
µA
µA
Undervoltage Lockout Threshold
VIN Falling (LTC1772C)
VIN Falling (LTC1772E, LTC1772I, LTC1772H)
VIN Rising
●
●
1.60
1.55
1.85
2.00
2.00
2.10
2.30
2.35
2.40
V
V
V
Shutdown Threshold (at ITH /RUN)
(LTC1772C)
(LTC1772E, LTC1772I, LTC1772H)
●
●
0.20
0.15
0.35
0.35
0.50
0.55
V
V
Start-Up Current Source
VITH/RUN = 0V
Regulated Feedback Voltage
(Note 5) (LTC1772C)
(Note 5) (LTC1772E, LTC1772I, LTC1772H)
0.25
0.5
0.85
µA
●
●
0.780
0.770
0.800
0.800
0.820
0.830
V
V
Output Voltage Line Regulation
2.4V ≤ VIN ≤ 9.8V (Note 5)
0.05
mV/V
Output Voltage Load Regulation
ITH/RUN Sinking 5µA (Note 5)
ITH/RUN Sourcing 5µA (Note 5)
2.5
2.5
mV/µA
mV/µA
VFB Input Current
(Note 5)
Overvoltage Protect Threshold
Measured at VFB
0.820
Overvoltage Protect Hysteresis
10
50
nA
0.860
0.895
V
20
500
550
120
mV
Oscillator Frequency
VFB = 0.8V
VFB = 0V
Gate Drive Rise Time
CLOAD = 3000pF
Gate Drive Fall Time
CLOAD = 3000pF
40
ns
Peak Current Sense Voltage
(Note 6)
120
mV
40
650
kHz
kHz
ns
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LTC1772
ELECTRICAL CHARACTERISTICS
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1772E is guaranteed to meet specifications from 0°C to
70°C. Specifications over the –40°C to 85°C operating temperature range
are assured by design, characterization and correlation with statistical
process controls. The LTC1772I is guaranteed to meet specified
performance from –40°C to 85°C. The LTC1772H is guaranteed to meet
specified performance from –40°C to 140°C.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA°C/W)
Operation at high junction temperatures degrades operating lifetimes.
Operating lifetimes at junction temperatures greater than 125°C is derated
to 1000 hours.
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC1772 is tested in a feedback loop that servos VFB to the
output of the error amplifier.
Note 6: Peak current sense voltage is reduced dependent on duty cycle to
a percentage of value as given in Figure 2.
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TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage
vs Temperature
10
VIN = 4.2V
8
810
805
800
795
790
785
780
775
–55 –35 –15 5 25 45 65 85 105 125 145
TEMPERATURE (°C)
2.10
6
2.06
4
2.02
2
0
–2
–4
1.86
1.78
1.74
–55 –35 –15 5 25 45 65 85 105 125 145
TEMPERATURE (°C)
1772 G03
1772 G02
Shutdown Threshold
vs Temperature
550
VIN = 4.2V
TA = 25°C
120
1.90
–8
Maximum (VIN – SENSE –) Voltage
vs Duty Cycle
130
1.94
1.82
–10
–55 –35 –15 5 25 45 65 85 105 125 145
TEMPERATURE (°C)
VIN FALLING
1.98
–6
1772 G01
510
VIN = 4.2V
470
110
TRIP VOLTAGE (V)
TRIP VOLTAGE (mV)
VFB VOLTAGE (mV)
815
2.14
VIN = 4.2V
TRIP VOLTAGE (V)
820
NORMALIZED FREQUENCY (%)
825
Undervoltage Lockout Trip
Voltage vs Temperature
Normalized Frequency
vs Temperature
100
90
80
70
430
390
350
310
270
230
60
50
190
20
30
40
50 60 70 80
DUTY CYCLE (%)
90
100
1772 G04
150
–55 –35 –15 5 25 45 65 85 105 125 145
TEMPERATURE (°C)
1772 G05
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LTC1772
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PIN FUNCTIONS
ITH/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run control input. The current comparator threshold
increases with this control voltage. Nominal voltage range
for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V
causes the device to be shut down. In shutdown all
functions are disabled and the PGATE pin is held high.
VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output.
SENSE – (Pin 4): The Negative Input to the Current Comparator.
VIN (Pin 5): Supply Pin. Must be closely decoupled to GND
Pin 2.
PGATE (Pin 6): Gate Drive for the External P-Channel
MOSFET. This pin swings from 0V to VIN.
GND (Pin 2): Ground Pin.
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FUNCTIONAL DIAGRA
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VIN
SENSE –
5
4
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+
ICMP
–
VIN
RS1
SLOPE
COMP
OSC
PGATE
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
R
Q
S
6
–
FREQ
FOLDBACK
+
SHORT-CIRCUIT
DETECT
0.3V
OVP
BURST
CMP
+
0.15V
SLEEP
–
VIN
EAMP
+
–
VREF
+
60mV
+
VREF
0.8V
0.5µA
VFB
+
–
1 ITH/RUN
3
VIN
VIN
0.3V
–
0.35V
VOLTAGE
REFERENCE
+
SHDN
CMP
VREF
0.8V
–
GND
SHDN
UV
2
UNDERVOLTAGE
LOCKOUT
1.2V
1772 FD
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LTC1772
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OPERATIO
(Refer to Functional Diagram)
Main Control Loop
The LTC1772 is a constant frequency current mode switching regulator. During normal operation, the external
P-channel power MOSFET is turned on each cycle when
the oscillator sets the RS latch (RS1) and turned off when
the current comparator (ICMP) resets the latch. The peak
inductor current at which ICMP resets the RS latch is
controlled by the voltage on the ITH/RUN pin, which is the
output of the error amplifier EAMP. An external resistive
divider connected between VOUT and ground allows the
EAMP to receive an output feedback voltage VFB. When the
load current increases, it causes a slight decrease in VFB
relative to the 0.8V reference, which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.
The main control loop is shut down by pulling the ITH/RUN
pin low. Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.35V, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
As the external compensation network continues to charge
up, the corresponding output current trip level follows,
allowing normal operation.
Comparator OVP guards against transient overshoots
> 7.5% by turning off the external P-channel power
MOSFET and keeping it off until the fault is removed.
Burst Mode Operation
The LTC1772 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
set as if VITH/RUN = 1V (at low duty cycles) even though
the voltage at the ITH/RUN pin is at a lower value. If the
inductor’s average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the
ITH/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the external MOSFET. The sleep signal
goes low when the ITH/RUN voltage goes above 0.925V
and the LTC1772 resumes normal operation. The next
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.
Dropout Operation
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means that
the external P-channel MOSFET will remain on for more
than one oscillator cycle since the inductor current has not
ramped up to the threshold set by EAMP. Further reduction in input supply voltage will eventually cause the
P-channel MOSFET to be turned on 100%, i.e., DC. The
output voltage will then be determined by the input voltage
minus the voltage drop across the MOSFET, the sense
resistor and the inductor.
Undervoltage Lockout
To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorporated into the LTC1772. When the input supply voltage
drops below approximately 2.0V, the P-channel MOSFET
and all circuitry is turned off except the undervoltage block,
which draws only several microamperes.
Short-Circuit Protection
When the output is shorted to ground, the frequency of the
oscillator will be reduced to about 120kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s frequency will gradually increase to its designed rate when
the feedback voltage again approaches 0.8V.
Overvoltage Protection
As a further protection, the overvoltage comparator in the
LTC1772 will turn the external MOSFET off when the
feedback voltage has risen 7.5% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 20mV.
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LTC1772
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OPERATIO
(Refer to Functional Diagram)
Slope Compensation and Inductor’s Peak Current
110
100
The inductor’s peak current is determined by:
VITH – 0.7
10(RSENSE )
when the LTC1772 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves
in Figure 2.
SF = IOUT/IOUT(MAX) (%)
IPK =
90
80
70
60
50
IRIPPLE = 0.4IPK
AT 5% DUTY CYCLE
IRIPPLE = 0.2IPK
AT 5% DUTY CYCLE
40
30
20
VIN = 4.2V
10
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1772 F02
Figure 2. Maximum Output Current vs Duty Cycle
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APPLICATIONS INFORMATION
The basic LTC1772 application circuit is shown in Figure 1.
External component selection is driven by the load requirement and begins with the selection of L1 and RSENSE
(= R1). Next, the power MOSFET, M1 and the output diode
D1 are selected followed by CIN (= C1) and COUT (= C2).
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator
determines the inductor’s peak current. The output current the LTC1772 can provide is given by:
IOUT =
0.12
I
− RIPPLE
RSENSE
2
where IRIPPLE is the inductor peak-to-peak ripple current
(see Inductor Value Calculation section).
A reasonable starting point for setting ripple current is
IRIPPLE = (0.4)(IOUT). Rearranging the above equation, it
becomes:
RSENSE =
1
for Duty Cycle < 40%
(10)(IOUT )
However, for operation that is above 40% duty cycle, slope
compensation effect has to be taken into consideration to
select the appropriate value to provide the required amount
of current. Using Figure 2, the value of RSENSE is:
RSENSE =
SF
(10)(IOUT )(100)
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.
The inductance value also has a direct effect on ripple
current. The ripple current, IRIPPLE, decreases with higher
inductance or frequency and increases with higher VIN or
VOUT. The inductor’s peak-to-peak ripple current is given
by:
IRIPPLE =
VIN − VOUT ⎛ VOUT + VD ⎞
⎜
⎟
f(L) ⎝ VIN + VD ⎠
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LTC1772
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APPLICATIONS INFORMATION
where f is the operating frequency. Accepting larger values
of IRIPPLE allows the use of low inductances, but results in
higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
IRIPPLE = 0.4(IOUT(MAX)). Remember, the maximum IRIPPLE
occurs at the maximum input voltage.
In Burst Mode operation on the LTC1772, the ripple
current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peakto-peak ripple current must not exceed:
0.03
IRIPPLE ≤
RSENSE
This implies a minimum inductance of:
LMIN =
VIN − VOUT ⎛ VOUT + VD ⎞
⎜
⎟
⎛ 0.03 ⎞ ⎝ VIN + VD ⎠
f⎜
⎟
⎝ RSENSE ⎠
(Use VIN(MAX) = VIN)
A smaller value than L MIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
Inductor Core Selection
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are very space efficient,
especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.
Power MOSFET Selection
An external P-channel power MOSFET must be selected
for use with the LTC1772. The main selection criteria for
the power MOSFET are the threshold voltage VGS(TH) and
the “on” resistance RDS(ON), reverse transfer capacitance
CRSS and total gate charge.
Since the LTC1772 is designed for operation down to low
input voltages, a sublogic level threshold MOSFET (RDS(ON)
guaranteed at VGS = 2.5V) is required for applications that
work close to this voltage. When these MOSFETs are used,
make sure that the input supply to the LTC1772 is less than
the absolute maximum VGS rating, typically 8V.
The required minimum RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications
that may operate the LTC1772 in dropout, i.e., 100% duty
cycle, at its worst case the required RDS(ON) is given by:
RDS(ON)
DC= 100%
=
PP
(IOUT(MAX) )2 (1+ δp)
where PP is the allowable power dissipation and δp is the
temperature dependency of RDS(ON). (1 + δp) is generally
given for a MOSFET in the form of a normalized RDS(ON) vs
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.
In applications where the maximum duty cycle is less than
100% and the LTC1772 is in continuous mode, the RDS(ON)
is governed by:
RDS(ON) ≅
PP
(DC )IOUT 2 (1+ δp)
where DC is the maximum operating duty cycle of the
LTC1772.
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LTC1772
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APPLICATIONS INFORMATION
Output Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches VOUT
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safely handle IPEAK at close to 100% duty cycle. Therefore,
it is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.
Under normal load conditions, the average current conducted by the diode is:
⎛V −V ⎞
ID = ⎜ IN OUT ⎟ IOUT
⎝ VIN + VD ⎠
The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as:
VF ≈
PD
ISC(MAX)
where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
CIN and COUT Selection
In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (VOUT + VD)/
(VIN + VD). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used. The maximum RMS capacitor current is given by:
1/ 2
VOUT (VIN − VOUT )]
[
CIN Required IRMS ≈ IMAX
VIN
This formula has a maximum value at VIN = 2VOUT, where
IRMS = IOUT /2. This simple worst-case condition is commonly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC1772, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:
⎛
1 ⎞
∆VOUT ≈ IRIPPLE ⎜ ESR +
⎟
⎝
4 fC OUT ⎠
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
Nichicon PL series and Panasonic SP.
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LTC1772
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APPLICATIONS INFORMATION
Low Supply Operation
Efficiency Considerations
Although the LTC1772 can function down to approximately 2V, the maximum allowable output current is
reduced when VIN decreases below 3V. Figure 3 shows the
amount of change as the supply is reduced down to 2V.
Also shown in Figure 3 is the effect of VIN on VREF as VIN
goes below 2.3V.
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
where η1, η2, etc. are the individual losses as a percentage of input power.
NORMALIZED VOLTAGE (%)
105
VREF
100
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1772 circuits: 1) LTC1772 DC bias current,
2) MOSFET gate charge current, 3) I2R losses and 4)
voltage drop of the output diode.
VITH
95
90
85
80
75
2.0
2.2
2.4
2.6
2.8
INPUT VOLTAGE (V)
3.0
1772 F03
Figure 3. Line Regulation of VREF and VITH
Setting Output Voltage
The LTC1772 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4). By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:
⎛ R2⎞
VOUT = 0.8 ⎜ 1 + ⎟
⎝ R1⎠
For most applications, an 80k resistor is suggested for R1.
To prevent stray pickup, locate resistors R1 and R2 close
to LTC1772.
VOUT
LTC1772
VFB
Efficiency = 100% – (η1 + η2 + η3 + ...)
R2
3
R1
1772 F04
Figure 4. Setting Output Voltage
1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.
2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge dQ moves from VIN to ground.
The resulting dQ/dt is a current out of VIN which is
typically much larger than the DC supply current. In
continuous mode, IGATECHG = f(Qp).
3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but
is “chopped” between the P-channel MOSFET (in series with RSENSE) and the output diode. The MOSFET
RDS(ON) plus RSENSE multiplied by duty cycle can be
summed with the resistances of L and RSENSE to obtain
I2R losses.
4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the
load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of
1772fb
9
LTC1772
U
U
W
U
APPLICATIONS INFORMATION
0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.
5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(VIN)2IO(MAX)CRSS(f)
Other losses including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
Foldback Current Limiting
As described in the Output Diode Selection, the worst-case
dissipation occurs with a short-circuited output when the
diode conducts the current limit value almost continuously. To prevent excessive heating in the diode, foldback
current limiting can be added to reduce the current in
proportion to the severity of the fault.
Foldback current limiting is implemented by adding diodes
DFB1 and DFB2 between the output and the ITH/RUN pin as
shown in Figure 5. In a hard short (VOUT = 0V), the current
VOUT
LTC1772
R2
ITH /RUN VFB
will be reduced to approximately 50% of the maximum
output current.
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1772. These items are illustrated graphically in the
layout diagram in Figure 6. Check the following in your
layout:
1. Is the Schottky diode closely connected between ground
(Pin 2) and drain of the external MOSFET?
2. Does the (+) plate of CIN connect to the sense resistor
as closely as possible? This capacitor provides AC
current to the MOSFET.
3. Is the input decoupling capacitor (0.1µF) connected
closely between VIN (Pin 5) and ground (Pin 2)?
4. Connect the end of RSENSE as close to VIN (Pin 5) as
possible. The VIN pin is the SENSE + of the current
comparator.
5. Is the trace from SENSE – (Pin 4) to the Sense resistor
kept short? Does the trace connect close to RSENSE?
6. Keep the switching node PGATE away from sensitive
small signal nodes.
+
DFB1
R1
7. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1 and R2 must be
connected between the (+) plate of COUT and signal
ground.
DFB2
1772 F05
Figure 5. Foldback Current Limiting
1
ITH/RUN PGATE
CIN
LTC1772
RITH
2
GND
VIN
5
3
VFB
SENSE –
L1
RSENSE
0.1µF
CITH
VIN
+
6
4
VOUT
M1
+
D1
COUT
R1
BOLD LINES INDICATE HIGH CURRENT PATHS
R2
1772 F06
Figure 6. LTC1772 Layout Diagram (See PC Board Layout Checklist)
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10
LTC1772
U
TYPICAL APPLICATIO
LTC1772 High Efficiency, Small Footprint 3.3V to 1.8V/0.5A Regulator
1
R4
10k
C3
220pF
ITH/RUN PGATE
6
L1
M1 10µH
LTC1772
2
3
GND
VFB
VIN
SENSE –
5
VIN
3.3V
C1
10µF
10V
R1
0.15Ω
+
D1
4
C2
47µF
6V
C1: TAIYO YUDEN CERAMIC
L1: COILTRONICS UP1B-100
LMK325BJ106K-T
M1: Si3443DV
C2: SANYO POSCAP 6TPA47M R1: DALE 0.25W
D1: MOTOROLA MBRM120T3
VOUT
1.8V
0.5A
R2
100k
R3
80.6k
1772 TA02
U
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic TSOT-23
(Reference LTC DWG # 05-08-1636)
0.62
MAX
2.90 BSC
(NOTE 4)
0.95
REF
1.22 REF
3.85 MAX 2.62 REF
1.4 MIN
2.80 BSC
1.50 – 1.75
(NOTE 4)
PIN ONE ID
RECOMMENDED SOLDER PAD LAYOUT
PER IPC CALCULATOR
0.30 – 0.45
6 PLCS (NOTE 3)
0.95 BSC
0.80 – 0.90
0.20 BSC
0.01 – 0.10
1.00 MAX
DATUM ‘A’
0.30 – 0.50 REF
0.09 – 0.20
(NOTE 3)
1.90 BSC
S6 TSOT-23 0302
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DRAWING NOT TO SCALE
3. DIMENSIONS ARE INCLUSIVE OF PLATING
4. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
5. MOLD FLASH SHALL NOT EXCEED 0.254mm
6. JEDEC PACKAGE REFERENCE IS MO-193
1772fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LTC1772
U
TYPICAL APPLICATIONS
LTC1772 3.3V to 5V/1A Boost Regulator
R1
0.033Ω
VIN
3.3V
C1
47µF
16V
×2
L1
4.7µH
D1
U1
1
R4
10k
ITH/RUN PGATE
6
2
2
3
VIN
GND
SENSE –
VFB
C1: AVXTPSE476M016R0047
C2: AVXTPSE107M010R0100
D1: IR10BQ015
4
VOUT
5V
1A
C2
100µF
10V
×2
+
M1
3
LTC1772
C3
220pF
5
5
4
R2
422k
R3
80.6k
U1: FAIRCHILD NC7SZ04
L1: MURATA LQN6C-4R7
M1: Si9804
R1: DALE 0.25W
ALSO SEE LTC1872
FOR THIS APPLICATION
1772 TA03
LTC1772 5V/0.5A Flyback Regulator
R1
0.033Ω
1
R4
10k
ITH/RUN PGATE
6
M1
LTC1772
2
C3
220pF
3
VIN
GND
VFB
SENSE –
5
4
VIN
2.5V
TO 9.8V
C2
47µF
16V
×2
C5
150pF
R6 CERAMIC
100Ω
D1
VOUT
5V
0.5A
T1
R5
22Ω
C1: AVXTPSE476M016R0047
C2: AVXTPSE107M010R0100
D1: IR10BQ015
•
C4
10µH
100pF
CERAMIC
+
10µH
•
M1: Si9803
R1: DALE 0.25W
T1: COILTRONICS CTX10-4
C2
100µF
10V
×2
R2
52.3k
R3
10k
1772 TA04
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1147 Series
High Efficiency Step-Down Switching Regulator Controllers
100% Duty Cycle, 3.5V ≤ VIN ≤ 16V
LT1375/LT1376
1.5A, 500kHz Step-Down Switching Regulators
High Frequency, Small Inductor, High Efficiency
LTC1622
Low Input Voltage Current Mode Step-Down DC/DC Controller
VIN 2V to 10V, IOUT Up to 4.5A, Synchronizable to
750kHz Optional Burst Mode Operation, 8-Lead MSOP
LTC1624
High Efficiency SO-8 N-Channel Switching Regulator Controller
N-Channel Drive, 3.5V ≤ VIN ≤ 36V
TM
LTC1625
No RSENSE Synchronous Step-Down Regulator
97% Efficiency, No Sense Resistor
LTC1627
Low Voltage, Monolithic Synchronous Step-Down Regulator
Low Supply Voltage Range: 2.65V to 8V, IOUT = 0.5A
LTC1649
3.3V Input Synchronous Controller
No Need for 5V Supply, Uses Standard Logic Gate
MOSFETs; IOUT up to 15A
LTC1702
550kHz, 2 Phase, Dual Synchronous Controller
Two Channels; Minimum CIN and COUT, IOUT up to 15A
LTC1735
Single, High Efficiency, Low Noise Synchronous Switching Controller
High Efficiency 5V to 3.3V Conversion at up to 15A
LTC1771
Ultra-Low Supply Current Step-Down DC/DC Controller
10µA Supply Current, 93% Efficiency,
1.23V ≤ VOUT ≤ 18V; 2.8V ≤ VIN ≤ 20V
LTC1872
SOT-23 Step-Up Controller
2.5V ≤ VIN ≤ 9.8V; 550kHz; 90% Efficiency
No RSENSE is a trademark of Linear Technology Corporation.
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12
Linear Technology Corporation
LT/LT 0605 500 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 1999
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