LINER LT1680I High power dc/dc step-up controller Datasheet

LT1680
High Power DC/DC
Step-Up Controller
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DESCRIPTION
FEATURES
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The LT ®1680 is a high power, current mode switching
power supply controller optimized for boost topologies.
The IC drives N-channel MOSFET switches for DC/DC
converters in applications up to 60V input. A high current
gate drive output handles up to 10,000pF gate capacitance, enabling the construction of high power DC/DC
converters. Current sense common mode range up to 60V
allows current sensing to be referenced to the input
supply, eliminating the need for sense blanking circuits.
High Voltage: Operation Up to 60V Max
High Current: N-Channel Drive Handles Up to
10,000pF Gate Capacitance
Programmable Average Current Limiting
5V Reference Output with 10mA External
Loading Capability
Fixed Frequency Current Mode Operation
Oscillator Synchronizable Up to 200kHz
Undervoltage Lockout with Hysteresis
Programmable Start Inhibit for Power Supply
Sequencing and Protection
User Adjustable Slope Compensation
The LT1680 incorporates programmable average current
limiting allowing accurate limiting of DC current in the
magnetics, independent of ripple current . User adjustable
slope compensation provides stable operation at duty
cycles up to 90%.
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APPLICATIONS
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The LT1680 operating frequency is programmable and
can be synchronized up to 200kHz. Minimum off-time
operation provides switch protection. The IC also incorporates a soft start feature that is gated by both shutdown
and undervoltage lockout conditions.
High Power Single Board Systems
Distributed Power Converters
Industrial Control Systems
Lead-Acid Battery Back-Up Systems
Automotive and Heavy Equipment
, LTC and LT are registered trademarks of Linear Technology Corporation.
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TYPICAL APPLICATION
12V to 48V, 250W Boost
12VIN
10V TO 15V
24A (DC) RSENSE
0.005Ω
C2
1µF
Efficiency vs Output Power
100
CCT
1nF
LT1680
1
2
C3 2.2nF
C4 0.22µF
CVC 4.7nF
3
4
1N914
RVC 4.7k
5
6
7
8
C6 0.1µF
R7
2k
SL/ADJ
5VREF
CT
SYNC
IAVG
12VIN
SS
GATE
VC
PGND
SGND RUN/SHDN
VFB
VREF
SENSE –
16
15
+
14
CIN
680µF
25V
×4
C12
1µF
13
12
11
M1
IRFZ44
×3
MBR0520
C11 1nF
R6
75k
VOUT = 48V
95
R9
100k
10
9
SENSE +
L1
25µH
EFFICIENCY (%)
RCT
15k
+
90
85
80
D1
MBR20100CT
×2
+
L1: Kool Mµ®, 18T #14 ON 77314-A7
Kool Mµ IS A REGISTERED TRADEMARK OF MAGNETICS, INC.
VOUT
48V
5.2A
COUT
680µF
63V
1680 TA01
×3
75
0
50
150
200
100
OUTPUT POWER (W)
250
1680 TA02
1
LT1680
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ABSOLUTE MAXIMUM RATINGS
PACKAGE/ORDER INFORMATION
(Note 1)
Power Supply Voltage (12VIN) .................. – 0.3V to 20V
Sense Amplifier Input Common Mode ...... – 0.3V to 60V
GATE Pin Voltage....................... – 0.3V to 12VIN + 0.3V
RUN/SHDN Pin Voltage ......................... – 0.3V to 12VIN
All Other Pin Voltages ................................. – 0.3V to 7V
5V Reference Output Current ............................... 65mA
Operating Ambient Temperature Range
LT1680C .................................................. 0°C to 70°C
LT1680I .............................................. – 40°C to 85°C
Storage Temperature Range ................ – 65°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
ORDER PART
NUMBER
TOP VIEW
SL/ADJ
1
16 5VREF
CT
2
15 SYNC
IAVG
3
14 12VIN
SS
4
13 GATE
VC
5
12 PGND
SGND
6
11 RUN/SHDN
VFB
7
10 SENSE –
VREF
8
9
N PACKAGE
16-LEAD PDIP
LT1680CN
LT1680CSW
LT1680IN
LT1680ISW
SENSE +
SW PACKAGE
16-LEAD PLASTIC SO WIDE
TJMAX = 125°C, θJA = 75°C/ W (N)
TJMAX = 125°C, θJA = 90°C/ W (SW)
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
12VIN = 12V, VVC = 2V, VFB = VREF = 1.25V, CGATE = 3000pF, TA = 25°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
15
12
22
mA
mA
65
110
µA
1.25
1.35
V
Supply and Protection
I12VIN
DC Active Supply Current (Note 2)
Gate Output On
Gate Output Off
●
DC Standby Supply Current
VRUN < 0.5V
●
VRUN/SHDN Shutdown Rising Threshold
●
1.15
VSSHYST
Shutdown Threshold Hysteresis
15
mV
ISS
Soft Start Charge Current
●
5
8
14
µA
VUVLO
Undervoltage Lockout Threshold - Falling
Undervoltage Lockout Threshold - Rising
Undervoltage Lockout Hysteresis
●
●
●
8.20
9.75
9.95
200
9.00
9.35
350
V
V
mV
4.75
5
5.25
V
5V Reference
VREF5
IREF5
5V Reference Voltage
Line, Load and Temperature
●
5V Reference Line Regulation
10V ≤ 12VIN ≤ 15V
●
5V Reference Load Range - DC
Pulse
5V Reference Load Regulation
ISC
3
●
●
0 ≤ IREF5 ≤ 20mA
–1.25
●
5V Reference Short-Circuit Current
5
mV/V
10
20
mA
mA
–2
V/A
45
mA
Error Amplifier
VFB
Error Amplifier Reference Voltage
Measured at Feedback Pin
1.258
1.265
V
V
●
0.1
0.5
1.0
µA
3200
IFB
Feedback Input Current
gm
Error Amplifier Transconductance
●
1200
2000
AV
Error Amplifier Voltage Gain
●
1500
3000
V/V
IVC
Error Amplifier Source Current
Error Amplifier Sink Current
VFB – VREF = 500mV
●
●
200
280
275
400
µA
µA
Absolute VC Clamp Voltage
Measured at VC Pin
3.5
V
VVC
2
VFB = VREF
1.242
1.235
1.250
●
µmho
LT1680
ELECTRICAL CHARACTERISTICS
12VIN = 12V, VVC = 2V, VFB = VREF = 1.25V, CGATE = 3000pF, TA = 25°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
170
110
190
120
130
mV
mV
Error Amplifier
VSENSE
VIAVG
Peak Current Limit Threshold
Average Current Limit Threshold
Measured at Sense Inputs
Measured at Sense Inputs, VCMSENSE = 10V
Average Current Limit Threshold
Measured at IAVG Pin
2.5
V
15
V/V
●
●
Current Sense Amplifier
AV
Amplifier DC Gain
Measured at IAVG Pin
VOS
Amplifier Input Offset Voltage
2V < VCMSENSE < 60V,
SENSE + – SENSE – = 5mV
●
IBIAS
Input Bias Current
Sink (VCMSENSE > 5V)
Source (VCMSENSE = 0V)
●
●
fO ≤ 200kHz, RCT = 16.9k, CCT = 1000pF
●
●
–5
LT1680C
LT1680I
●
●
2.20
2.10
0.1
mV
45
700
75
1200
µA
µA
200
5
kHz
%
2.75
2.75
mA
mA
V
Oscillator
fO
Operating Frequency, Free Run
Frequency Programming Error
ICT
Timing Capacitor Discharge Current
2.5
2.5
VSYNC
SYNC Input Threshold
Rising Edge
●
0.8
2.0
fSYNC
SYNC Frequency Range
fSYNC ≤ 200kHz
●
fO
1.4fO
12VIN ≤ 8.2V
VRUN < 0.5V
●
●
●
●
Output Drivers
VGATE
Undervoltage Output Clamp
Standby Mode Output Clamp
Gate Output On Voltage
Gate Output Off Voltage
0.4
11
11.9
0.4
0.7
0.1
12
0.7
V
V
V
V
tGATER
Gate Output Rise Time
●
60
200
ns
tGATEF
Gate Output Fall Time
●
60
140
ns
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: Absolute Maximum Ratings are those values beyond which the life of
a device may be impaired.
Note 2: Supply current specification does not include external FET gate
charge currents. Actual supply currents will be higher and vary with
operating frequency, operating voltages and the type of external FETs used.
See Applications Information.
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LT1680
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TYPICAL PERFORMANCE CHARACTERISTICS
I12VIN Supply Current
vs Temperature
I12VIN Supply Current
vs 12VIN Supply Voltage
19
fO = 150kHz
TA = 25°C
17
16
15
14
13
35
CG = 10nF
30
25
CG = 4.7nF
20
CG = 3.3nF
CG = 1nF
12
11
–50 –25
0
25
50
75
100
GATE TRANSITION TIME (ns)
I12VIN SUPPLY CURRENT (mA )
18
I12VIN SUPPLY CURRENT (mA)
Gate Transition Time vs CGATE
150
40
12
13
14
11
12VIN SUPPLY VOLTAGE (V)
1680 G01
•
•
tf
tr
90
•
•
70
•
•
0
15
2500
5000
CGATE (pF)
7500
I12VIN Shutdown Current
vs Temperature
60
10000
1680 G03
1680 G02
5VREF Short-Circuit Current
vs Temperature
VREF Voltage
vs Temperature
80
1.252
75
1.251
55
50
45
40
35
30
– 50 –25
50
25
75
0
TEMPERATURE (°C)
100
VREF VOLTAGE (V)
5VREF = 0V
I12VIN SHUTDOWN CURRENT (µA)
5VREF SHORT-CIRCUIT CURRENT (mA)
110
30
10
TEMPERATURE (°C)
70
65
60
50
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
100
125
1.246
–50 –25
125
1680 G07
ERROR AMPLIFIER TRANSCONDUCTANCE (m )
ERROR AMPLIFIER VOLTAGE GAIN (kV/V)
100
4.0
3.5
3.0
2.5
2.0
1.5
1.0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1680 G06
Ω
4.99
50
25
75
0
TEMPERATURE (°C)
Error Amplifier Transconductance
vs Temperature
4.5
5.00
4
1.248
Error Amplifier Voltage Gain
vs Temperature
5.01
50
25
75
0
TEMPERATURE (°C)
1.249
1680 G05
5VREF Voltage
vs Temperature
4.98
–50 –25
1.250
1.247
55
1680 G04
5VREF VOLTAGE (V)
•
50
15
125
TA = 25°C
130
100
125
1680 G08
2.6
2.4
2.2
2.0
1.8
1.6
1.4
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1680 G09
LT1680
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TYPICAL PERFORMANCE CHARACTERISTICS
Error Amplifier Source Current
vs Temperature
SS Output Current
vs Temperature
1.26
9
RUN/SHDN RISING THRESHOLD (V)
SS OUTPUT CURRENT (µA)
325
300
275
250
8
7
225
200
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
6
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
1680 G10
1.23
1.22
1.21
1.20
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
125
UVLO Thresholds vs Temperature
10.00
160
FULL OPERATING
TEMPERATURE RANGE
150
13
100
1680 G12
Average Current Limit Threshold
Sense Voltage vs Common Mode
Voltage
14
9.75
9.50
140
RISING
UPPER LIMIT
11
10
9
8
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
130
V12VIN (V)
VSENSE (mV)
12
TYPICAL
120
LOWER LIMIT
110
8.25
0
1
2
3
4
5
8.00
–50 –25
60
800
RUN/SHDN INPUT CURRENT (nA )
1100
55
1000
IB(SINK) (µA)
50
45
40
600
35
500
25
50
75
100
125
TEMPERATURE (°C)
1680 G16
75
30
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
RUN/SHDN Input Current
vs Pin Voltage
VCMSENSE = 10V
700
50
1680 G15
60
VCMSENSE = 0V
800
25
1680 G14
Sense Amplifier Input Bias
Current (Sink) vs Temperature
900
0
TEMPERATURE (°C)
VSENSE(CM) (V)
1200
0
FALLING
8.75
90
Sense Amplifier Input Bias
Current (Source) vs Temperature
400
–50 –25
9.00
8.50
80
125
9.25
100
1680 G13
IB(SOURCE) (µA)
1.24
1680 G11
RUN/SHDN Threshold Hysteresis
vs Temperature
RUN/SHDN THRESHOLD HYSTERESIS (mV)
100
1.25
100
125
1680 G17
FULL OPERATING
TEMPERATURE
RANGE
700
600
500
400
UPPER
LIMIT
300
200
100
0
0
..................................................................
ERROR AMPLIFIER SOURCE CURRENT (µA)
350
RUN/SHDN Rising Threshold
vs Temperature
TYPICAL
LOWER
LIMIT
1.0 (1.25) 1.5
0.5
RUN/SHDN PIN VOLTAGE (V)
2.0
1680 G18
5
LT1680
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TYPICAL PERFORMANCE CHARACTERISTICS
RUN/SHDN Input Current
vs Pin Voltage
100
1.01
90
UPPER
LIMIT
450
TYPICAL
300
LOWER
LIMIT
150
IDISCHG = 2.75mA
80
70
60
50
IDISCHG = 2.1mA
40
30
20
FULL OPERATING
TEMPERATURE
RANGE
10
0
2
4
6
8
10
RUN/SHDN PIN VOLTAGE (V)
12
0
1
2
4
1680 G19
6
10
20
RCT (kΩ)
40 60 100
1680 G20
OPERATING FREQUENCY (NORMALIZED)
FULL OPERATING
TEMPERATURE
RANGE
MAXIMUM DUTY CYCLE (%)
RUN/SHDN INPUT CURRENT (µA)
600
0
Operating Frequency
(Normalized) vs Temperature
Maximum Duty Cycle vs RCT
1.00
0.99
0.98
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1680 G21
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PIN FUNCTIONS
SL/ADJ (Pin 1): Slope Compensation Adjustment. Allows
increased slope compensation for certain high duty cycle
applications. Resistive loading of this pin increases effective slope compensation. A resistor divider from the 5VREF
pin can tailor the onset of additional slope compensation
to specific regions in each switch cycle. Pin can be floated
or connected to 5VREF if no additional slope compensation
is required. (See Applications Information section for
slope compensation details.)
CT (Pin 2): Oscillator Timing Pin. Connect a capacitor
(CCT) to ground and a pull-up resistor (RCT) to the 5VREF
supply. Typical values are CCT = 1000pF and 10k ≤ RCT ≤
30k.
IAVG (Pin 3): Average Current Limit Integration. Frequency response characteristic is set using the 50kΩ
output impedance and external capacitor to ground.
Averaging roll-off is typically set 1 to 2 orders of magnitude below switching frequency. (Typical capacitor value
= 1000pF for fO = 100kHz.) Shorting this pin to SGND will
disable the average current limit function. In systems
where open-loop inductor current occurs, such as boost
supplies during output short-circuit condition and inrush
periods, an external small-signal protection diode should
be connected between IAVG and the VC pin (anode to IAVG
pin, cathode to VC pin). See Applications Information.
6
SS (Pin 4): Soft Start. Generates ramping threshold for
regulator current limit during start-up and after UVLO
events by sourcing about 10µA into an external capacitor.
VC (Pin 5): Error Amplifier Output. RC load creates dominant compensation in power supply regulation feedback
loop to provide optimum transient response. (See Applications Information section for compensation details.)
SGND (Pin 6): Small-Signal Ground. Connect to negative
terminal of COUT.
VFB (Pin 7): Error Amplifier Inverting Input. Used as
voltage feedback input node for regulator loop. Pin sources
about 0.5µA DC bias current to protect from an open
feedback path condition.
VREF (Pin 8): Bandgap Generated Voltage Reference
Decoupling. Connect a capacitor to signal ground. (Typical capacitor value ≈ 0.1µF.)
SENSE + (Pin 9): Current Sense Amplifier Inverting Input.
Connect to most positive (DC) terminal of current sense
resistor.
SENSE – (Pin 10): Current Sense Amplifier Noninverting
Input. Connect to most negative (DC) terminal of current
sense resistor.
LT1680
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PIN FUNCTIONS
RUN/SHDN (Pin 11): Precision Referenced Shutdown.
Can be used as logic level input for shutdown control or as
an analog monitor for input supply undervoltage protection, etc. IC is enabled when RUN/SHDN pin rising edge
exceeds 1.25V. 15mV of hysteresis helps assure stable
mode switching. All internal functions are disabled in
shutdown mode. If this function is not desired, connect
RUN/SHDN to 12VIN (typically through a 100k resistor).
See Applications Information.
PGND (Pin 12): Power Ground. References the output
switch and internal driver control circuits. Connect with
low impedance trace to VIN decoupling capacitor negative
(ground) terminal.
GATE (Pin 13): Driver Output. Connect to gate of external
power FET switch.
12VIN (Pin 14): 12V Power Supply Input. Bypass with at
least 1µF to PGND.
SYNC (Pin 15): Oscillator Synchronization Pin with
TTL-Level Compatible Input. Input drives internal rising
edge triggered one shot; SYNC signal on/off times should
be ≥1µs (10% to 90% duty cycle at 100kHz). Does not
contain internal pull-up. Connect to SGND if not used.
5VREF (Pin 16): 5V Reference Output. Allows connection
of external loads up to 10mA DC. Reference is not available
during shutdown. Typically bypassed with at least 1µF
capacitor to SGND.
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BLOCK DIAGRAM
SYNC
GATE
REFERENCE
5V
1.25V
5VREF
ONE SHOT
–
Q
S
VIN
RSENSE
UVLO
CIRCUIT
R
RCT
OSC
CT
CCT
+
SL/ADJ
+
×15
VC
–
+
VOUT
SENSE +
SENSE –
IC1
VOUT
CURRENT
SENSE
AMPLIFIER
VREF
–
IAVG
+
+
EA
VFB
–
–
2.5V
12VIN
SS
0.5µA
1.25V
RUN SHDN
12V
RUN/SHDN
10µA
SOFT START
–
PGND
CIRCUIT
ENABLE
+
SGND
1680 BD
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LT1680
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OPERATION
Basic Control Loop
The LT1680 uses a constant frequency, current mode
architecture. The timing of the IC is provided through an
internal oscillator circuit that can be synchronized to an
external clock and is programmable to operate at frequencies up to 200kHz. The oscillator creates a modified
sawtooth wave at its timing node (CT) with a slow charge,
rapid discharge characteristic.
During typical boost converter operation, the MOSFET
switch is enabled at the start of each oscillator cycle. The
switch stays enabled until the current through the switched
inductor, sensed via the voltage across a series sense
resistor (RSENSE), is sufficient to trip the current comparator (IC1) and reset the RS latch. When the switch is
disabled, the inductor current is redirected to the supply
output. If the current comparator threshold is not reached
throughout the entire oscillator charge period, the RS
latch is bypassed and the main switch is disabled during
the oscillator discharge time. This “minimum off time”
protects the switch, and is typically about 1µs.
The current comparator trip threshold is set on the VC pin,
which is the output of a transconductance amplifier, or
error amplifier (EA). The error amplifier integrates the
difference between a feedback voltage (on the VFB pin) and
an internal bandgap generated reference voltage of 1.25V,
forming a signal that represents required load current. If
the supplied current is insufficient for a given load, the
output will droop, thus reducing the feedback voltage. The
error amplifier responds by forcing current out of the VC
pin, increasing the current comparator threshold. Thus,
the circuit will servo until the provided current is equal to
the required load and the average output voltage is at the
value programmed by the feedback resistors.
Input Average Current Limit
The output of the sense amplifier is monitored by a single
pole integrator comprised of an external capacitor on the
IAVG pin and an output impedance of approximately 50kΩ.
If this averaged value signal exceeds a level corresponding
to 120mV across the external sense resistor, the current
comparator threshold is clamped and cannot continue to
rise in response to the error amplifier. Thus, if average
input current requirements exceed 120mV/RSENSE, the
8
supply will current limit and the output voltage will fall out
of regulation. The average current limit circuit monitors
the sense amplifier output without slope compensation or
ripple current contributions. Therefore, the average input
current limit threshold is unaffected by duty cycle.
Undervoltage Lockout
The LT1680 employs an undervoltage lockout circuit
(UVLO) that monitors the 12VIN supply rail. This circuit
disables the output drive capability of the LT1680 if the
12V supply drops below 9V. Unstable mode switching is
prevented through 350mV of UVLO threshold hysteresis.
Shutdown
The LT1680 can be put into low current shutdown by
pulling the RUN/SHDN pin low, disabling all circuit functions. The shutdown threshold is a bandgap referred
voltage of 1.25V typical. Use of a precision threshold on
the shutdown circuit enables use of this pin for undervoltage protection of the VIN supply and/or power supply
sequencing.
Soft Start
The LT1680 incorporates a soft start function that operates by slowly increasing current limit. This limit is
controlled by internally clamping the VC pin to a low
voltage that climbs with time as an external capacitor on
the SS pin is charged with about 10µA. This forces a
graceful climb of output current source capability, and
thus a graceful increase in output voltage until steadystate regulation is achieved. The soft start timing capacitor is clamped to ground during shutdown and during
undervoltage lockout, yielding a graceful output recovery
from either condition.
5V Internal Reference
Power for the oscillator timing elements and most other
internal LT1680 circuits is derived from an internal 5V
reference, accessible at the 5VREF pin. This supply pin
can be loaded with up to 10mA DC (20mA pulsed) for
convenient biasing of local elements such as control
logic, etc.
LT1680
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OPERATION
Slope Compensation
For duty cycles greater than 50%, slope compensation is
required to prevent current mode duty cycle instability in
the regulator control loop. The LT1680 employs internal
slope compensation that is adequate for most applica-
tions. However, if additional slope compensation is desired, it is available through the SL/ADJ pin. Excessive
slope compensation will cause reduction in maximum
load current capability and is generally not desirable.
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APPLICATIONS INFORMATION
RSENSE Selection for Input Current Limit
RSENSE generates a voltage that is proportional to the
inductor current for use by the LT1680 current sense
amplifier. The value of RSENSE is based on the required
input current. The average current limit function has a
typical threshold of 120mV/RSENSE, or:
RSENSE = 120mV/ILIMIT
Operation with VSENSE common mode voltage below 4.5V
may slightly degrade current limit accuracy. See Average
Current Limit Threshold Tolerance vs Common Mode
Voltage in the Typical Performance Characteristics section for more information.
Output Voltage Programming
Output voltage is programmed through a resistor feedback network to the VFB pin (Pin 7) on the LT1680. This pin
is the inverting input of the error amplifier, which is
internally referenced to 1.25V. The divider is ratioed to
provide 1.25V at the VFB pin when the output is at its
desired value. Output voltage is thus set following the
relation:
VOUT = 1.25V(1 + R2/R1)
when an external resistor divider is connected to the
output as shown in Figure 1.
VOUT
R2
LT1680
VFB
SGND
7
R1
6
1680 F01
If high value feedback resistors are used, the input bias
current of the VFB pin (1µA maximum) could cause a slight
increase in output voltage. A Thevenin resistance at the
VFB pin of < 5k is recommended.
Oscillator Components RCT and CCT
The LT1680 oscillator creates a modified sawtooth at its
timing node (CT) with a slow charge, rapid discharge
characteristic. The discharge time (tDISCH) corresponds to
the minimum off time of the PWM controller. This limits
maximum duty cycle (DCMAX) to:
DCMAX = 1 – (tDISCH)(fO)
This relation corresponds to the minimum value of the
timing resistor (RCT), which can be determined according
to the following relation (RCT vs DCMAX graph appears in
the Typical Performance Characteristics section):
RCT(MIN) ≈ [(0.8)(10 – 3)(1 – DCMAX)] – 1
Values for RCT > 15k yield maximum duty cycles above
90%. Given a timing resistor value, the value of the timing
capacitor (CCT) can then be determined for desired operating frequency (fO) using the relation:
(1/ fO ) – (100) 10– 9
CCT ≈
(RCT / 1.85) +  – 31.75
(2.5) 10  – (3.375 / RCT )
A plot of Operating Frequency vs RCT and CCT is shown in
Figure 2. Typical 100kHz operational values are CCT =
1000pF and RCT = 16.9k.
Figure 1. Programming LT1680 Output Voltage
9
LT1680
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and a soft start timing capacitor CSS, the start-up delay
time to full available average current will be:
200
CCT = 0.68nF
OSCILLATOR FREQUENCY (kHz)
180
CCT = 1nF
tSS = (1.8)(105)(CSS)
160
CCT = 1.5nF
140
CCT = 2.2nF
Shutdown Function—Input Undervoltage Detect
and Threshold Hysteresis
120
100
80
60
40
3
7 11 15 19 23 27 31 35 39 43 47
TIMING RESISTOR (kΩ)
1680 F02
Figure 2. Operating Frequency vs RCT, CCT
Average Current Limit
The average current limit function is implemented using
an external capacitor (CAVG) connected from IAVG to
SGND. This capacitor forms a single pole integrator with
the 50kΩ output impedance of the IAVG pin. The integrator
corner frequency is typically set 1 to 2 orders of magnitude
below the oscillator frequency and follows the relation:
f– 3dB = (3.2)(10 – 6)/CAVG
The average current limit function can be disabled by
shorting the IAVG pin directly to SGND. In some applications it is theoretically possible for the average current
limit circuit to overdrive the error amplifier output (VC pin)
beyond the operating range of the current sense comparator. These applications include those where open-loop
system operation occurs, such as boost regulators in
output short-circuit condition, or in systems with poor
signal ground integrity. The potential for this overdrive can
be eliminated by connecting an external clamp diode
between the IAVG and VC pins (anode to IAVG and cathode
to VC). Connection of this diode will have no adverse
effects in any system and is recommended. This clamp is
required for all boost converter topologies.
Soft Start Programming
The LT1680 current control pin (VC) limits inductor current to zero at voltages less than ≈0.7V through full
average current limit at VC ≈ 2.5V, yielding 1.8V over the
full regulation range of average load current. With the SS
pin at 0V, the VC pin is clamped to its zero inductor current
level. Given the typical soft start charge current of 10µA
10
The LT1680 RUN/SHDN pin uses a bandgap generated
reference threshold of about 1.25V. This precision threshold allows use of the RUN/SHDN pin for both logic-level
shutdown applications and analog monitoring applications such as power supply sequencing.
Because an LT1680 controlled converter is a power transfer device, a voltage that is lower than expected on the
input supply could require currents that exceed the sourcing capabilities of that supply, causing the system to lockup in an undervoltage state. Input supply start-up protection
can be achieved by enabling the RUN/SHDN pin using a
resistor divider from the input supply to ground. Setting
the divider output to 1.25V when the supply is almost fully
enabled prevents the LT1680 regulator from drawing large
currents until the input supply is able to supply the
required power.
If additional hysteresis is desired for the enable function,
an external feedback resistor can be used from the LT1680
regulator output. If connection to the regulator output is
not desired, the 5VREF internal supply pin can be used.
Figure 3 shows an input supply sequencing configuration
on a 24V input converter. This configuration yields an
enable condition of 90% VIN (~ 21.5V) with about 10%
threshold hysteresis.
The shutdown function can be disabled by connecting the
RUN/SHDN pin to the 12VIN rail. This pin is internally
clamped to 2.5V through a 20k series input resistance and
will therefore draw 0.5mA when tied directly to 12V. This
VIN
24V
160k
16
390k
LT1680
11
10k
5VREF
RUN/SHDN
1680 F03
Figure 3. Input Supply Sequencing Programming
LT1680
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additional current can be minimized by making the connection through an external resistor (100k is typically used).
MINIMUM SHUTDOWN CONTROL LIMIT (mV)
When shutting down the LT1680, the RUN/SHDN pin voltage must remain between the shutdown threshold (~1.13V)
and a minimum shutdown control limit voltage (see Figure 4) for a least 25µs. If a digital input or fast moving
clamp is used, this can be achieved by forcing a shutdown
control voltage above the minimum limit or by using a
simple integrator to increase the fall time of the input signal. A single pole integrator stage must have a
τ ≥ (7)(10 – 5).
800
DIGITAL
INPUT
R1
10k
RUN/SHDN
C1
10nF
LT1680
1680 F06
Figure 6. Digital Input Shutdown Integration Control
Figure 7 is an example of an integrator stage coupled with
a 24V input power supply sequencing circuit similar to that
shown in Figure 3. The integrator stage allows use of an
active shutdown clamp for implementation of both usercontrolled shutdown and input power supply sequencing
protection.
VIN
24V
R1
160k
R3
390k
5VREF
R4
10k
700
LT1680
RUN/SHDN
SHDN
R2
10k
C1
10nF
1680 F07
600
Figure 7. Input Supply Sequencing with
User-Controlled Shutdown
500
– 40
– 20
0
20
40
60
TEMPERATURE (°C)
80
1680 F04
Figure 4. Minimum Shutdown Control Limit vs Temperature
Figure 5 is an example of a digital control input clamp. A
logic high signal pulls the RUN/SHDN pin above its turnon threshold through the diode. When a shutdown (logic
low) signal is received, the RUN/SHDN pin is forced to
0.95V via the resistor divider until shutdown is fully established and the 5VREF voltage collapses.
Oscillator Synchronization
The LT1680 oscillator generates a modified sawtooth
waveform at the CT pin between low and high thresholds
of 0.8V (vl) and 2.5V (vh) respectively. The oscillator can
be synchronized by driving a TTL level pulse into the SYNC
pin. This pin connects to a one shot circuit that reduces the
oscillator high threshold to 2V for about 200ns. The SYNC
input signal should have minimum on/off times of ≥1µs.
SYNC
2.5V
5VREF
1N914
DIGITAL
INPUT
R1
43k
(vh)
2V
LT1680
VCT
RUN/SHDN
R2
10k
1680 F05
(vl)
0.8V
FREE RUN
SYNCHRONIZED
1680 F08
Figure 5. Digital Input Shutdown Level Control
Figure 6 is an example of a digital control integration stage
at the RUN/SHDN input. The integrator has a τ = (10)(103)
• (10)(10 –9) = (1.0)(10 – 4). This circuit technique, however,
delays initiation of controller shutdown about 125µs from
the reception of the shutdown signal (5V – 0V transition).
Figure 8. Free Run and Synchronized Oscillator
Waveforms (at CT Pin)
Inductor Selection
The inductor for an LT1680 converter is selected based on
output power, operating frequency and efficiency require-
11
LT1680
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ments. Generally, the selection of inductor value can be
reduced to desired maximum ripple current in the inductor
(∆I). For a boost converter, the minimum inductor value
for a given operating ripple current can be determined
using the following relation:
L MIN =
(
)
(∆I)(fO)(VOUT )
VIN VOUT – VIN
Given an inductor value (L), the peak inductor current is
the sum of the average inductor current (IAVG) and half the
inductor ripple current (∆I), or:
IPK = IAVG +
(
)
(2)(L)(fO)(VOUT )
VIN VOUT – VIN
The inductor core type is determined by peak current and
efficiency requirements. The inductor core must withstand this peak current without saturating, and the series
winding resistance and core losses should be kept as
small as is practical to maximize conversion efficiency.
The LT1680 peak current threshold is 40% greater than
the average limit threshold. Slope compensation effects
reduce this margin as duty cycle increases. This margin
must be maintained to prevent peak current limit from
corrupting the programmed value for average current
limit. Programming the peak ripple current to less than
15% of the desired average current limit value will assure
proper operation of the average current limit feature
through 90% duty cycle (see Slope Compensation).
Slope Compensation
Current mode switching regulators that operate with a
duty cycle greater than 50% and have continuous inductor
current can exhibit duty cycle instability. While a regulator
will not be damaged and may even continue to function
acceptably during this type of subharmonic oscillation, an
irritating high-pitched squeal is usually produced.
The criterion for current mode duty cycle instability is
met when the increasing slope of the inductor ripple
current is less than the decreasing slope, which is the
case at duty cycles greater than 50%. This condition is
illustrated in Figure 9a. The inductor ripple current starts
12
at I1, the beginning of each oscillator switch cycle.
Current increases at a rate S1 until the current reaches
the control trip level I2. The controller servo loop then
disables the switch and inductor current begins to decrease at a rate S2. If the current switch point (I2) is
perturbed slightly and increased by ∆I, the cycle time
ends such that the minimum current point is increased by
a factor of 1 + (S2/S1) to start the next cycle. On each
successive cycle, this error is multiplied by a factor of S2/
S1. Therefore, if S2/S1 is ≥ 1, the system is unstable.
Subharmonic oscillations can be eliminated by augmenting the increasing ripple current slope (S1) in the control
loop. This is accomplished by adding an artificial ramp on
the inductor current waveform internal to the IC (with a
slope SX) as shown in Figure 9b. If the sum of the slopes
S1 + SX is greater than S2, this condition for subharmonic
oscillation no longer exists.
∆I
T1
S1 + SX
I2
I1
0
S1
S2
S1
S2
OSCILLATOR
PERIOD
0
TIME
a
b
1680 F09
Figure 9. Inductor Current at DC > 50% and
Slope Compensation Adjusted Signal
For boost topologies, the required additional current waveform slope, or “Slope Compensation,” follows the relation:
SX ≥
(S1)(2DC – 1)
(1– DC)
For duty cycles less than 50% (DC < 0.5), SX is negative and
is not required. For duty cycles greater than 50%, SX takes
on values dependent on S1 and duty cycle. S1 is simply VIN/
L. This leads to a minimum inductance requirement for a
given VIN, duty cycle and slope compensation (SX) of:
 VIN 
  (2DC – 1)
S 
L MIN = X
1 – DC
The LT1680 contains an internal slope compensation
ramp that has an equivalent current referred value of:
LT1680
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 fO 
SX = 0.084

 RSENSE 
Amp/s
where fO is oscillator frequency and RSENSE is the external
current sense resistor. This yields a minimum inductance
requirement of:
(V )(R
)(2DC – 1)
L MIN ≥ IN SENSE
[(0.084)(fO)(1− DC )]
SX =
A down side of slope compensation is that, since the IC
servo loop senses an increase in perceived inductor current, the internal current limit functions are affected such
that the maximum current capability of a regulator is
reduced by the same amount as the effective current
referred slope compensation. The LT1680, however, uses
a current limit scheme that is independent of the slope
compensation effects (Average Current Limiting). This
provides operation at any duty cycle with no reduction in
current sourcing capability, provided ripple current peak
amplitude is less than 15% of the current limit value. For
example, if the converter is set up to average current limit
at 10A, as long as the peak inductor current is less than
11.5A, duty cycles up to 90% can be achieved without
compromising the average current limit value.
If an inductor smaller than the minimum required for
internal slope compensation (calculated above as LMIN) is
desired, additional slope compensation is necessary. The
LT1680 provides this capability through the SL/ADJ pin.
MAXIMUM PEAK RIPPLE CURRENT (IPK/IAVG)
This feature is implemented by referencing this pin via a
resistor divider from the 5VREF pin to ground. The additional slope compensation will be affected at the point in
the oscillator waveform (at pin CT) corresponding to the
voltage set by the resistor divider. Additional slope compensation can be calculated using the relation:
1.45
(2500)(fO)
(RTH )(RSENSE )
Amp/s
where RTH is the Thevenin resistance of the resistor
divider. Actual compensation will actually be somewhat
greater due to internal curvature correction circuitry that
imposes an exponential increase in the slope compensation waveform, further increasing the effective compensation slope up to 20% for a given setting.
Design example:
VIN = 20V
VOUT = 80V (DC = 0.75)
RSENSE = 0.01Ω
fO = 100kHz
L = 20µH
The minimum inductor usable with no additional slope
compensation is:
LMIN ≥
(20V)(0.01Ω)(1.5 – 1) = 47.6µH
(0.084)(100000)(1– 0.75)
Since L = 20µH is less than LMIN, additional slope
compensation is necessary. The total slope compensation required is:
1.40
 20V 

 (1.5 – 1)
 20µH 
SX ≥
= (2)(106 )
1 – 0.75
1.35
1.30
1.25
Amp/s
Subtracting the internally generated slope compensation
and solving for the required effective resistance at SL/ADJ
yields:
1.20
1.15
1.10
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
DUTY CYCLE
1680 F10
Figure 10. Maximum Peak Ripple Current (Normalized)
vs Duty Cycle for Average Current Limit
REQ ≤
(2500)(fO )
= 21.5k
6

(2) 10  (RSENSE ) – (0.084)(fO)
13
LT1680
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Setting the resistor divider reference voltage to 2V assures
that the additional compensation waveform will be enabled at a 75% duty cycle. As shown in Figure 11a, using
RSL1 = 45k and RSL2 = 30k sets the desired reference
voltage and has a RTH of 18k, which meets both design
requirements. Figure 11b shows the slope compensation
effective waveforms both with and without the SL/ADJ
external resistors.
16
RSL1
45k
RSL2
30k
5VREF
LT1680
1
SL/ADJ
1680 F11a
Figure 11a. External Slope Compensation Resistors
In a typical LT1680 boost converter, the switch current is
equal to the inductor current, but is chopped according to
duty cycle (DC). The conduction loss (PLOSS) for a given
FET RDS(ON) can be calculated using the relation:
PLOSS ≈ (DC)(RDS(ON))(IAVG2 + [∆I2/12])
where IAVG = average inductor current and ∆I = peak-topeak inductor ripple current.
The output diode is often a major source of power loss in
switching regulators and selection of adequately rated
diodes is important. In a boost converter, when the output
voltage is significantly higher than the input voltage, the
peak diode current becomes much higher than average
output currents and diode current ratings must be observed with caution. The peak diode current is:
ID(PEAK) = IAVG + ∆I/2
2.5V
and the average power dissipation (PD) in the diode is:
2V
PD = (IOUT)(Vf)
where Vf is the forward voltage of the diode at peak
current. The output diode must also be rated for maximum
reverse voltages exceeding VOUT.
0.8V
DC = 0.75
(0.084 + 0.139)(fO)
RSENSE
(0.084)(fO)
RSENSE
1680 F11b
Figure 11b. Slope Compensation Waveforms
Power MOSFET and Output Rectifying Diode Selection
LT1680 converter system parameters that dictate selection criteria for the switch MOSFET and output rectifying
diode include maximum load current (IOUT), inductor
average current (IAVG) and inductor ripple current (∆I),
and maximum input and output voltages.
The switch MOSFETs selected must have a maximum
operating VDSS exceeding the maximum output voltage
(VOUT). VGS rated operating maximums must exceed the
12VIN supply voltage. Once voltage requirements have
been determined, switch conduction resistance (RDS(ON))
can be determined based on allowable power dissipation.
14
CIN and COUT Supply Decoupling Capacitor Selection
The large currents typical of LT1680 applications require
special consideration for the regulator input and output
supply decoupling capacitors.
Under normal steady state boost operation, output current
provided by the converter is a square wave of duty cycle VIN/
VOUT, the average value being equal to the required DC load
current (IOUT). The continuity of the load current is maintained by the output bypass capacitors. To prevent excessive output voltage ripple and undue capacitor heating (and
associated catastrophic failure), low ESR output capacitors
sized for the maximum RMS current must be used. This
maximum capacitor RMS current follows the relation:
V

IRMS ≈ IOUT  OUT – 1
 VIN

1/ 2
Capacitor ripple current ratings are often based on only 2000
hours (3 months) lifetime; it is advisable to derate either the
ESR or temperature rating of capacitors for increased MTBF.
LT1680
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The input bypass capacitors generally have less ripple
current than the output bypass capacitors as the input
current in a boost converter is continuous. Input bypass
capacitor selection can be made using ripple current
ratings. Peak-to-peak ripple current is equal to the inductor ripple current (∆IL).
Efficiency Considerations and Heat Dissipation
High output power applications create an inherent concern regarding power dissipation in regulator components. Although high efficiencies are achieved using the
LT1680, the power dissipated in the regulator climbs to
relatively high values when the load draws large amounts
of power. Even at 90% efficiency, a 500W application has
conversion loss of 55W.
I2R dissipation in the MOSFET switch, sense resistor and
inductor series resistance can generate substantial conversion loss under high current conditions. Generally, the
dominant I2R loss is evidenced in the FET switch, which is
proportional to the steady-state duty cycle, or conduction
time of the switch. For example, in a 5V to 48V boost
converter, the duty cycle is:
DC = 1 – (VIN / VOUT)
DC = 1 – 5/48 ≈ 90%
The FET switch conducts inductor current for almost 90%
of the cycle time, and thus may require increased consideration for dissipating I2R power.
U
PACKAGE DESCRIPTION
Gate Drive Buffer
The LT1680 is designed to drive relatively large capacitive
loads. However, in certain applications, efficiency improvements can be realized by adding an external buffer
stage to drive the gate of the FET switch. When the switch
gate loads the driver output such that rise/fall times
exceed 100ns, buffers can sometimes result in efficiency
gains. Buffers can also reduce effects of back injection into
the gate driver output due to coupling of switch node
transitions through the switch FET CMILLER.
Optimizing Transient Response–
Compensation Component Values
The dominant compensation point for an LT1680 converter is the VC pin (Pin 5), or error amplifier output. This
pin connects to an external series RC network, RVC and
CVC. The infinite permutations of input/output filtering,
capacitor ESR, input voltage, load current, etc. make for an
empirical method of optimizing loop response for a specific set of conditions.
Loop response can be observed by injecting a step change
in load current. This can be achieved by using a switchable
load. With the load switching, the transient response of the
output voltage can be observed with an oscilloscope.
Iterating through RC combinations will yield optimized
response. Refer to Application Note 19 in the 1990 Linear
Applications Handbook, Volume 1 for more information.
Dimensions in inches (millimeters) unless otherwise noted.
N Package
16-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.130 ± 0.005
(3.302 ± 0.127)
0.300 – 0.325
(7.620 – 8.255)
0.009 – 0.015
(0.229 – 0.381)
(
+0.035
0.325 –0.015
+0.889
8.255
–0.381
)
0.770*
(19.558)
MAX
0.045 – 0.065
(1.143 – 1.651)
0.020
(0.508)
MIN
0.065
(1.651)
TYP
0.125
(3.175)
MIN
0.100 ± 0.010
(2.540 ± 0.254)
0.018 ± 0.003
(0.457 ± 0.076)
16
15
14
13
12
11
10
1
2
3
4
5
6
7
9
0.255 ± 0.015*
(6.477 ± 0.381)
8
N16 1197
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
15
LT1680
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TYPICAL APPLICATION
– 48V to 5V 30W Forward Converter
33Ω
+
24k
220µF*
35V
– 48V
INPUT
•
1.5nF
220µF*
35V
+
1µF
63V
L1
20µH
MBR2045CT
220µF*
35V
1k
33Ω
MBR2045CT
+
•
*SANYO CV-GX
**SANYO OS-CON
ALL RESISTORS 1.4W, 1% UNLESS
INDICATED OTHERWISE
5V
6A
OUT
•
220µF*
35V
•
24k
+
•
+
•
T1
INPUT
COM
300pF
1.5nF
0.033µF
50Ω
1W
330µF**
6.3V
OUTPUT
COM
0.015Ω
1W
IRF640
4.22k
10Ω
MBR0520LT1
24k
9
78.7k
BAV21
11
+
1M
•
0.1µF
220µF
35V
5VREF
4.75k
IAVG
2
SS
3
4
VFB
SGND PGND VREF
VC
5
6
12
8
0.22µF
0.1µF
16k
20k
2N3904
7
2.2nF
2N7000
SL/ADJ
LT1680
CT
16
1µF
1
SYNC
GATE
12VIN
7.5k
15
13
SENSE –
RUN/SHDN
14
L1
10
SENSE +
Q7
2N5401
1nF 0.1µF
20k
1k
L1: PHILIPS EFD20-3F3-E63-S
(CORE SET, AI = 63nH/T2)
OUTPUT 18T BIFILAR 22AWG
BIAS 54T BIFILAR 32AWG
T1: COILTRONICS VP5-1200, 1:1:1:1:1:1
(SIX WINDINGS EACH 77µH)
1.2k
1680 TA03
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PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
SW Package
16-Lead Plastic Small Outline (Wide 0.300)
(LTC DWG # 05-08-1620)
0.291 – 0.299**
(7.391 – 7.595)
0.010 – 0.029 × 45°
(0.254 – 0.737)
0.093 – 0.104
(2.362 – 2.642)
0.398 – 0.413*
(10.109 – 10.490)
0.037 – 0.045
(0.940 – 1.143)
16
15
14
13
12
11 10
9
0° – 8° TYP
0.009 – 0.013
(0.229 – 0.330)
NOTE 1
0.050
(1.270)
TYP
0.016 – 0.050
(0.406 – 1.270)
0.004 – 0.012
(0.102 – 0.305)
0.394 – 0.419
(10.007 – 10.643)
NOTE 1
0.014 – 0.019
(0.356 – 0.482)
TYP
NOTE:
1. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS.
THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1
2
3
4
5
6
7
8
S16 (WIDE) 0396
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1268
7.5A, 150kHz Switching Regulator
Integrated Switch Can Be Used in Isolated Flyback Mode
LT1270A
10A, 60kHz Switching Regulator
Integrated Switch Can Be Used in Isolated Flyback Mode
LT1339
High Power Synchronous DC/DC Controller
Operation to 60V, No Shoot-Through N-Channel Output Drivers
LT1370
500kHz, 6A Boost Switching Regulator
Integrated Switch, Regulates Positive or Negative Outputs
LT1371
500kHz, 3A Boost Switching Regulator
Integrated Switch, Regulates Positive or Negative Outputs
16
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900
FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com
1680f LT/TP 0298 4K • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1997
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