LINER LT3681EDE 36v, 2a, 2.8mhz step-down switching regulator with integrated power schottky diode Datasheet

LT3681
36V, 2A, 2.8MHz Step-Down
Switching Regulator with
Integrated Power Schottky Diode
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FEATURES
DESCRIPTIO
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The LT®3681 is an adjustable frequency (300kHz to 2.8MHz)
monolithic buck switching regulator that accepts input
voltages up to 34V (36V maximum). A high efficiency
0.18Ω switch is included on the die along with a boost
Schottky diode and the necessary oscillator, control, and
logic circuitry. An undedicated power Schottky diode is
integrated into the LT3681 to minimize the solution size.
Current mode topology is used for fast transient response
and good loop stability. Low ripple Burst Mode operation
maintains high efficiency at low output currents while
keeping output ripple below 15mV in a typical application.
In addition, the LT3681 can further enhance low output
current efficiency by drawing bias current from the output
when VOUT is above 3V. Shutdown reduces input supply
current to less than 1µA while a resistor and capacitor on
the RUN/SS pin provide a controlled output voltage ramp
(soft-start). A power good flag signals when VOUT reaches
90% of the programmed output voltage. The LT3681
is available in 14-Pin 4mm x 3mm DFN package with
exposed pads for low thermal resistance.
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Wide Input Voltage Range: 3.6V to 34V Operating,
36V Maximum
2A Maximum Output Current
Low Ripple Burst Mode® Operation
50µA IQ at 12VIN to 3.3VOUT
Output Ripple < 15mVP-P
Adjustable Switching Frequency: 300kHz to 2.8MHz
Low Shutdown Current: IQ < 1µA
Integrated Boost Diode
Integrated Power Schottky Diode
Power Good Flag
Saturating Switch Design: 0.18Ω On-Resistance
1.265V Feedback Reference Voltage
Output Voltage: 1.265V to 20V
Soft-Start Capability
Small 14-Pin Thermally Enhanced
(4mm x 3mm) DFN Package
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APPLICATIO S
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Automotive Battery Regulation
Power for Portable Products
Distributed Supply Regulation
Industrial Supplies
Wall Transformer Regulation
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation. All other
trademarks are the property of their respective owners.
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TYPICAL APPLICATIO
5V Step-Down Converter
BD
VIN
OFF ON
Efficiency
VOUT
5V
2A
RUN/SS
BOOST
LT3681
L
6.8mH
SW
DC
RT
DA
330pF
BIAS
PG
60.4k
60
0.1000
50
40
0.0100
30
VIN = 12V
VOUT = 3.3V
L = 4.7mH
F = 800 kHz
20
590k
GND
1.0000
70
10
FB
22mF
200k
0
0.0001
POWER LOSS (W)
VC
90
80
0.47mF
20k
4.7mF
10.0000
100
EFFICIENCY (%)
VIN
6.3V TO
34V
0.0010
0.0001
0.001
0.01
0.1
ILOAD (A)
1
10
3681 TA01b
L: NEC PLC-0745-6R8
3681 TA01
3681f
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LT3681
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ABSOLUTE
AXI U RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
TOP VIEW
VIN, RUN/SS Voltage .................................................36V
BOOST Pin Voltage ...................................................56V
BOOST Pin Above SW Pin.........................................30V
FB, RT, VC Voltage .......................................................5V
BIAS, PG, BD Voltage ................................................30V
Maximum Junction Temperature .......................... 125°C
DC above DA .............................................................40V
Operating Temperature Range (Note 2)
LT3681E............................................... –40°C to 85°C
Storage Temperature Range................... –65°C to 150°C
PG
1
BIAS
2
14 RUN/SS
FB
3
13 VIN
12 SW
GND
4
11 BOOST
VC
5
RT
6
GND
7
15
10 BD
16
9 DC
8 DA
DE14MA PACKAGE
14-LEAD (4mm ´ 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 43°C/W
EXPOSED PAD (PIN 15) IS GND, MUST BE SOLDERED TO PCB
EXPOSED PAD PIN 16 IS DC
ORDER PART NUMBER
DE PART MARKING
LT3681EDE
3681
Order Options Tape and Reel: Add #TR
Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF
Lead Free Part Marking: http://www.linear.com/leadfree/
Consult LTC Marketing for parts specified with wider operating temperature ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUNS/SS = 10V, VBOOST = 15V, VBIAS = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
●
Minimum Input Voltage
Quiescent Current from VIN
MIN
VRUN/SS = 0.2V
VBIAS = 3V, Not Switching
●
VBIAS = 0, Not Switching
Quiescent Current from BIAS
VRUN/SS = 0.2V
VBIAS = 3V, Not Switching
●
VBIAS = 0, Not Switching
Minimum Bias Voltage
Feedback Voltage
●
FB Pin Bias Current (Note 3)
VFB = 1.25V, VC = 0.4V
FB Voltage Line Regulation
4V < VIN < 34V
●
1.25
1.24
TYP
MAX
UNITS
3
3.6
V
0.01
0.5
μA
22
60
μA
75
120
μA
0.01
0.5
μA
50
120
μA
0
5
μA
2.7
3
V
1.265
1.265
1.29
1.3
V
V
30
100
nA
0.002
0.02
%/V
μMho
Error Amp GM
330
Error Amp Gain
800
VC Source Current
65
μA
VC Sink Current
85
μA
VC Pin to Switch Current Gain
3.5
A/V
VC Clamp Voltage
2
V
3681f
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LT3681
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VRUNS/SS = 10V VBOOST = 15V, VBIAS = 3.3V unless otherwise
noted. (Note 2)
PARAMETER
CONDITIONS
Power Schottky Diode Forward Voltage
IDA = 1A
IDA = 2A
Power Schottky Diode Leakage Current
VDC-DA = 40V
Switching Frequency
RT = 8.66k
RT = 29.4k
RT = 187k
MIN
MAX
0.50
0.56
2.5
1.25
250
●
Minimum Switch Off-Time
TYP
3.2
UNITS
V
V
100
μA
2.8
1.4
300
3.1
1.55
350
MHz
MHz
kHz
130
200
nS
3.8
4.4
A
Switch Current Limit
Duty Cycle = 5%
Switch VCESAT
ISW = 2A
360
Boost Schottky Reverse Leakage
VSW = 10V, VBIAS = 0V
0.02
2
µA
1.5
2.1
V
●
Minimum Boost Voltage (Note 4)
mV
BOOST Pin Current
ISW = 1A
18
35
mA
RUN/SS Pin Current
VRUN/SS = 2.5V
5
10
μA
RUN/SS Input Voltage High
2.5
V
RUN/SS Input Voltage Low
PG Threshold Offset from Feedback Voltage
0.2
VFB Rising
122
PG Leakage
VPG = 5V
0.1
PG Sink Current
VPG = 3V
PG Hysteresis
mV
5
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3681E is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
●
100
600
V
mV
1
μA
μA
Note 3: Bias current flows into the FB pin.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
3681f
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LT3681
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TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
Efficiency (VOUT = 3.3V)
Efficiency (VOUT = 5.0V)
90
VIN = 12V
80
EFFICIENCY (%)
50
40
40
20
20
0.001
0.01
0.1
LOAD CURRENT (A)
1
VIN = 24V
50
30
L: NEC PLC-0745-4R7
f: 800kHz
VIN = 12V
60
30
0
0.0001
80
70
60
70
65
0
0.0001
50
0.001
0.01
0.1
LOAD CURRENT (A)
1
0
10
100
90
Maximum Load Current
90
30
20
3.5
LOAD CURRENT (A)
SUPPLY CURRENT (µA)
40
INCREASED SUPPLY
CURRENT DUE TO CATCH
DIODE LEAKAGE AT
HIGH TEMPERATURE
80
70
60
50
3.0
2.5
2.0
VOUT = 3.3V
L = 4.7µH
f = 800 kHz
1.5
10
FRONT PAGE APPLICATION
0
0
5
10
20
25
15
INPUT VOLTAGE (V)
30
40
–40
35
1.0
–20
60
0
20
40
TEMPERATURE (°C)
3.5
3.5
3.0
2.5
2.0
VOUT = 5.0V
L = 4.7µH
f = 800 kHz
10
20
25
15
INPUT VOLTAGE (V)
30
3681 G07
30
Switch Current Limit
DUTY CYCLE = 10 %
4.0
3.0
2.5
2.0
3.0
2.5
DUTY CYCLE = 90 %
2.0
1.5
1.0
0.5
1.0
5
25
20
15
INPUT VOLTAGE (V)
4.5
1.5
1.0
10
3681 G06
SWITCH CURRENT LIMIT (A)
4.0
SWITCH CURRENT LIMIT(A)
LOAD CURRENT (A)
5
Switch Current Limit
Maximum Load Current
1.5
80
3681 G05
3681 G04
4.0
3
4.0
VIN = 12V
VOUT = 3.3V
80
50
1
2
2.5
1.5
SWITCHING FREQUENCY (MHz)
3681 G03
No Load Supply Current vs
Temperature
60
0.5
3681 G02
No Load Supply Current
70
VOUT = 3.3V
L = 10µH
LOAD = 1A
55
L: NEC PLC-0745-4R7
f: 800kHz
3681 G01
SUPPLY CURRENT (µA)
VIN = 24V
75
60
10
10
VIN = 12V
85
VIN = 7V
80
VIN = 24V
70
10
90
EFFICIENCY (%)
90
EFFICIENCY (%)
Efficiency vs Switching Frequency
100
100
0
20
60
40
DUTY CYCLE (%)
80
100
3681 G08
0
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
125
3681 G09
3681f
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LT3681
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TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
Switch Voltage Drop
400
300
200
1.290
80
1.285
FEEDBACK VOLTAGE (V)
500
90
70
BOOST PIN CURRENT (mA)
VOLTAGE DROP (mV)
600
60
50
40
30
100
0
500
0
–25
0
25
50
75
TEMPERATURE (°C)
Frequency Foldback
RT = 45.3kΩ
100
125
4681 G12
Minimum Switch On-Time
140
1200
SWITCHING FREQUENCY (kHz)
1.05
1.00
0.95
0.90
0.85
100
MINIMUM SWITCH ON TIME (ns)
RT = 45.3kΩ
1.10
FREQUENCY (MHz)
1.250
–50
500 1000 1500 2000 2500 3000 3500
SWITCH CURRENT (mA)
3681 G11
1.15
0
25
50
75
TEMPERATURE (°C)
1.265
1.255
1000 1500 2000 2500 3000 3500
SWITCH CURRENT (mA)
–25
1.270
10
Switching Frequency
0.80
–50
1.275
1.260
3681 G10
1.20
1.280
20
0
0
1000
800
600
400
200
0
200
120
100
80
60
40
20
0
–50
0
125
400 600 800 1000 1200 1400
FB PIN VOLTAGE (mV)
4681 G13
Soft Start
RUN/SS Pin Current
2.0
1.5
1.0
100
125
Boost Diode
1.6
1.4
10
BOOST DIODE Vf (V)
RUN/SS PIN CURRENT (µA)
3.5
2.5
25
0
50
75
TEMPERATURE (˚C)
3681 G15
12
3.0
–25
3681 G14
4.0
SWITCH CURRENT LIMIT (A)
Feedback Voltage
Boost Pin Current
700
8
6
4
1.2
1.0
0.8
0.6
0.4
2
0.5
0.2
0
0
0
0.5
1
2
2.5
1.5
RUN/SS PIN VOLTAGE (V)
3
3.5
3681 G16
0
5
20
30
15
25
10
RUN/SS PIN VOLTAGE (V)
35
3681 G17
0
0
1.0
0.5
1.5
BOOST DIODE CURRENT (A)
2.0
3681 G18
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LT3681
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TYPICAL PERFOR A CE CHARACTERISTICS TA = 25°C unless otherwise noted.
Error Amp Output Current
Minimum Input Voltage
Minimum Input Voltage
100
4.5
6.5
4.0
6.0
INPUT VOLTAGE (V)
VC PIN CURRENT (µA)
60
40
20
0
–20
–40
3.5
3.0
2.5
–60
–80
1.065
1.165
1.365
1 .265
FB PIN VOLTAGE (V)
INPUT VOLTAGE (V)
80
0.1
0.01
1
LOAD CURRENT (A)
1.200
2.00
1.180
THRESHOLD VOLTAGE (V)
2.50
1.00
SWITCHING THRESHOLD
Switching Waveforms;
Burst Mode
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 10mA
IL
0.5A/DIV
1.160
VSW
5V/DIV
1.140
VOUT
10mV/DIV
1.120
0.50
10
3681 G20
Power Good Threshold
CURRENT LIMIT CLAMP
0.1
0.01
1
LOAD CURRENT (A)
3681 G20
VC Voltages
1.50
VOUT = 5.0V
L = 4.7m
f = 800kHz
4.0
0.001
10
3681 G19
THRESHOLD VOLTAGE (V)
5.0
4.5
VOUT = 3.3V
L = 4.7m
f = 800kHz
2.0
0.001
1.465
5.5
PG RISING
0
–50
–25
0
50
25
75
TEMPERATURE (°C)
100
125
1.100
–50
–25
0
50
25
75
TEMPERATURE (°C)
100
125
2µs/DIV
3681 G24
3681 G23
3681 G22
Switching Waveforms; Transition
from Burst Mode to Full
Frequency
Power Schottky Diode Forward
Voltage vs Current
Switching Waveforms; Full
Frequency Continuous Operation
6000
IL
0.5A/DIV
VRUN/SS
5V/DIV
VRUN/SS
5V/DIV
VOUT
10mV/DIV
VOUT
10mV/DIV
3681 G25
4000
3000
2000
1000
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 1A
VIN = 12V; FRONT PAGE APPLICATION
ILOAD = 140mA
1µs/DIV
5000
CURRENT (mA)
IL
0.5A/DIV
1µs/DIV
0
3681 G26
0
0.1
0.2 0.3 0.4 0.5
FORWARD VOLTAGE (V)
0.6
0.7
3681 G29
3681f
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LT3681
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PI FU CTIO S
PG (Pin 1): The PG pin is the open collector output of an
internal comparator. PG remains low until the FB pin is
within 10% of the final regulation voltage. PG output is
valid when VIN is above 3.5V and RUN/SS is high.
BIAS (Pin 2): The BIAS pin supplies the current to the
LT3681’s internal regulator. Tie this pin to the lowest
available voltage source above 3V (typically VOUT). This
architecture increases efficiency especially when the input
voltage is much higher than the output.
FB (Pin 3): The LT3681 regulates the FB pin to 1.265V.
Connect the feedback resistor divider tap to this pin.
VC (Pin 5): The VC pin is the output of the internal error
amplifier. The voltage on this pin controls the peak switch
current. Tie an RC network from this pin to ground to
compensate the control loop.
RT (Pin 6): Oscillator Resistor Input. Connecting a resistor
to ground from this pin sets the switching frequency.
DA (Pin 8): This is the anode of the integrated power
Schottky diode. High frequency, large amplitude currents
flow through this pin, so tie it to ground through a low
impedance connection.
DC (Pin 9, Exposed Pad 16): These pins connect to the
cathode of the integrated power Schottky diode. High frequency, large amplitude currents flow through these pins,
so tie them to SW through a low impedance connection.
BD (Pin 10): This pin connects to the anode of the internal
boost Schottky diode.
BOOST (Pin 11): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch. Connect a capacitor between
this pin and SW.
SW (Pin 12): The SW pin is the output of the internal
power switch. Connect this pin to the inductor, DC and
boost capacitor.
VIN (Pin 13): The VIN pin supplies current to the LT3681’s
internal regulator and to the internal power switch. This
pin must be locally bypassed.
RUN/SS (Pin 14): The RUN/SS pin is used to put the
LT3681 in shutdown mode. Tie to ground to shut down
the LT3681. Tie to 2.3V or more for normal operation. If
the shutdown feature is not used, tie this pin to the VIN
pin. RUN/SS also provides a soft-start function; see the
Applications Information section.
GND (Pins 4, 7, Exposed Pad 15): All three of these
terminals internally connect to the LT3681 control IC’s
signal return, while exposed pad 15 performs the added
function of providing a low thermal resistance heat flow
path between the IC and the system heatsink. Tie all of
these terminals to a copper pour on the top layer of the
printed circuit board. Please refer to the Applications
Information section for more details.
3681f
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LT3681
W
BLOCK DIAGRA
VIN
13
VIN
C1
2
14
6
BIAS
–
+
INTERNAL 1.265V REF
RUN/SS
5
SLOPE COMP
BD
SWITCH
LATCH
BOOST
11
C3
R
RT
OSCILLATOR
300kHz–2.8MHz
Q
S
SW
RT
DISABLE
DC
SOFT-START
1
10
BurstMode
DETECT
DC
L1
VOUT
12
C2
9
16
PG
ERROR AMP
+
–
GND GND
4
7
+
–
1.12V
FB
GND
15
VC CLAMP
DA
VC
8
5
CC
RC
CF
3
R2
R1
3681 BD
OPERATION
The LT3681 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
sets an RS flip-flop, turning on the internal power switch.
An amplifier and comparator monitor the current flowing
between the VIN and SW pins, turning the switch off when
this current reaches a level determined by the voltage at
VC. An error amplifier measures the output voltage through
an external resistor divider tied to the FB pin and servos
the VC pin. If the error amplifier’s output increases, more
current is delivered to the output; if it decreases, less current is delivered. An active clamp on the VC pin provides
current limit. The VC pin is also clamped to the voltage on
the RUN/SS pin; soft-start is implemented by generating a
voltage ramp at the RUN/SS pin using an external resistor
and capacitor.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor is used to generate
a voltage at the BOOST pin that is higher than the input
supply. This allows the driver to fully saturate the internal
bipolar NPN power switch for efficient operation.
An internal regulator provides power to the control circuitry. The bias regulator normally draws power from the
VIN pin, but if the BIAS pin is connected to an external
voltage higher than 3V bias power will be drawn from the
external source (typically the regulated output voltage).
This improves efficiency. The RUN/SS pin is used to place
the LT3681 in shutdown, disconnecting the output and
reducing the input current to less than 1µA.
The LT3681 contains a power good comparator which trips
when the FB pin is at 91% of its regulated value. The PG
output is an open-collector transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3681 is
enabled and VIN is above 3.6V.
To further optimize efficiency, the LT3681 automatically
switches to Burst Mode operation in light load situations.
Between bursts, all circuitry associated with controlling
the output switch is shut down reducing the input supply
current to 55µA in a typical application.
The oscillator reduces the LT3681’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup
and overload.
The LT3681 integrates a high quality, 36V, 2A power
Schottky diode to reduce the overall solution size.
3681f
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LT3681
APPLICATIONS INFORMATION
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
⎞
⎛V
R1= R2 ⎜ OUT – 1⎟
⎝ 1.265 ⎠
Reference designators refer to the Block Diagram.
Setting the Switching Frequency
The LT3681 uses a constant frequency PWM architecture
that can be programmed to switch from 300kHz to 2.8MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Figure 1.
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.3
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
2.2
2.4
2.6
2.8
187
133
84.5
60.4
45.3
36.5
29.4
23.7
20.5
16.9
14.3
12.1
10.2
8.66
Figure 1. Switching Frequency vs. RT Value
Operating Frequency Tradeoffs
Selection of the operating frequency is a tradeoff between
efficiency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower efficiency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (fSW(MAX)) for a
given application can be calculated as follows:
fSW(MAX ) =
VD + VOUT
tON(MIN) ( VD + VIN – VSW )
where VIN is the typical input voltage, VOUT is the output voltage, VD is the power Schottky catch diode drop
(~0.55V), VSW is the internal switch drop (~0.5V at max
load). This equation shows that slower switching frequency
is necessary to safely accommodate high VIN/VOUT ratio.
Also, as shown in the next section, lower frequency allows a lower dropout voltage. The reason input voltage
range depends on the switching frequency is because the
LT3681 switch has finite minimum on and off times. The
switch can turn on for a minimum of ~150ns and turn off
for a minimum of ~150ns. This means that the minimum
and maximum duty cycles are:
DCMIN = fSW tON(MIN)
DCMAX = 1– fSW tOFF(MIN)
where fSW is the switching frequency, the tON(MIN) is the
minimum switch on time (~150ns), and the tOFF(MIN) is
the minimum switch off time (~150ns). These equations
show that duty cycle range increases when switching
frequency is decreased.
A good choice of switching frequency should allow adequate input voltage range (see next section) and keep
the inductor and capacitor values small.
Input Voltage Range
The maximum input voltage for LT3681 applications depends on switching frequency, the Absolute Maximum Ratings on VIN and BOOST pins, and on operating mode.
If the output is in start-up or short-circuit operating modes,
then VIN must be below 34V and below the result of the
following equation:
VIN(MAX ) =
VOUT + VD
–V +V
fSW tON(MIN) D SW
where VIN(MAX) is the maximum operating input voltage,
VOUT is the output voltage, VD is the catch diode drop
(~0.55V), VSW is the internal switch drop (~0.5V at max
load), fSW is the switching frequency (set by RT), and
tON(MIN) is the minimum switch on time (~150ns). Note that
a higher switching frequency will depress the maximum
operating input voltage. Conversely, a lower switching
3681f
9
LT3681
APPLICATIONS INFORMATION
frequency will be necessary to achieve safe operation at
high input voltages.
If the output is in regulation and no short-circuit or start-up
events are expected, then input voltage transients of up to
36V are acceptable regardless of the switching frequency.
In this mode, the LT3681 may enter pulse skipping operation where some switching pulses are skipped to maintain
output regulation. In this mode the output voltage ripple
and inductor current ripple will be higher than in normal
operation.
The minimum input voltage is determined by either the
LT3681’s minimum operating voltage of ~3.6V or by its
maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:
VIN(MIN) =
VOUT + VD
–V +V
1– fSW tOFF(MIN) D SW
where VIN(MIN) is the minimum input voltage, and tOFF(MIN)
is the minimum switch off time (150ns). Note that higher
switching frequency will increase the minimum input
voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.
Inductor Selection
For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current ΔIL increases with higher VIN or VOUT
and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting
the ripple current is:
ΔIL = 0.4(IOUT(MAX))
where IOUT(MAX) is the maximum output load current. To
guarantee sufficient output current, peak inductor current
must be lower than the LT3681’s switch current limit (ILIM).
The peak inductor current is:
at least 3.5A at low duty cycles and decreases linearly to
2.5A at DC = 0.8. The maximum output current is a function of the inductor ripple current:
IOUT(MAX) = ILIM – ΔIL/2
Be sure to pick an inductor ripple current that provides
sufficient maximum output current (IOUT(MAX)).
The largest inductor ripple current occurs at the highest
VIN. To guarantee that the ripple current stays below the
specified maximum, the inductor value should be chosen
according to the following equation:
⎛V +V ⎞⎛ V +V ⎞
L = ⎜ OUT D ⎟ ⎜ 1– OUT D ⎟
VIN(MAX ) ⎟⎠
⎝ f∆IL ⎠ ⎜⎝
where VD is the voltage drop of the integrated Schottky
diode (~0.55V), VIN(MAX) is the maximum input voltage,
VOUT is the output voltage, fSW is the switching frequency
(set by RT), and L is in the inductor value.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. For robust operation in fault conditions
(start-up or short circuit) and high input voltage (>30V),
the saturation current should be above 3.5A. To keep the
efficiency high, the series resistance (DCR) should be less
than 0.1Ω, and the core material should be intended for
high frequency applications. Table 1 lists several vendors
and suitable types.
Table 1. Inductor Vendors
VENDOR
URL
PART SERIES
TYPE
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com
SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
D75C
Shielded
D75F
Open
FDV0620
Shielded
CDRH74
Shielded
CDRH6D38
Shielded
CR75
Open
CDRH8D43
Shielded
PLC-0745
Shielded
Sumida
www.sumida.com
IL(PEAK) = IOUT(MAX) + ΔIL/2
where IL(PEAK) is the peak inductor current, IOUT(MAX) is
the maximum output load current, and ΔIL is the inductor
ripple current. The LT3681’s switch current limit (ILIM) is
NEC
www.nec.tokin.com
3681f
10
LT3681
APPLICATIONS INFORMATION
Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger
value inductor provides a slightly higher maximum load
current and will reduce the output voltage ripple. If your
load is lower than 2A, then you can decrease the value of
the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one
with a lower DCR resulting in higher efficiency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is okay
but further reduces maximum load current. For details of
maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50% (VOUT/VIN > 0.5), there
is a minimum inductance required to avoid subharmonic
oscillations. See Application Note 19.
Input Capacitor
Bypass the input of the LT3681 circuit with a ceramic capacitor of X7R or X5R type. Y5V types have poor performance
over temperature and applied voltage, and should not be
used. A 4.7µF to 10µF ceramic capacitor is adequate to
bypass the LT3681 and will easily handle the ripple current.
Note that larger input capacitance is required when a lower
switching frequency is used. If the input power source has
high impedance, or there is significant inductance due to
long wires or cables, additional bulk capacitance may be
necessary. This can be provided with a low performance
electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3681 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 4.7µF capacitor is capable of this task, but only if it is
placed close to the LT3681 and the catch diode (see the
PCB Layout section). A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT3681. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT3681 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT3681’s
voltage rating. This situation is easily avoided (see the Hot
Plugging Safely section).
For space sensitive applications, a 2.2µF ceramic capacitor can be used for local bypassing of the LT3681 input.
However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and
may couple noise into other circuitry. Also, the increased
voltage ripple will raise the minimum operating voltage
of the LT3681 to ~3.7V.
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it filters the square wave generated by the
LT3681 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3681’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
100
COUT =
VOUT fSW
where fSW is in MHz, and COUT is the recommended
output capacitance in µF. Use X5R or X7R types. This
choice will provide low output ripple and good transient
response. Transient performance can be improved with
a higher value capacitor if the compensation network is
also adjusted to maintain the loop bandwidth. A lower
value of output capacitor can be used to save space and
cost but transient performance will suffer. See the Frequency Compensation section to choose an appropriate
compensation network.
3681f
11
LT3681
APPLICATIONS INFORMATION
Table 2. Capacitor Vendors
VENDOR
PHONE
URL
PART SERIES
Panasonic
(714) 373-7366
www.panasonic.com
Ceramic,
Polymer,
COMMENTS
EEF Series
Tantalum
Kemet
(864) 963-6300
www.kemet.com
Ceramic,
Tantalum
Sanyo
(408) 749-9714
www.sanyovideo.com
T494, T495
Ceramic,
Polymer,
POSCAP
Tantalum
Murata
(408) 436-1300
AVX
www.murata.com
Ceramic
www.avxcorp.com
Ceramic,
Tantalum
Taiyo Yuden
(864) 963-6300
www.taiyo-yuden.com
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher voltage
rating, may be required. High performance tantalum or
electrolytic capacitors can be used for the output capacitor.
Low ESR is important, so choose one that is intended for
use in switching regulators. The ESR should be specified
by the supplier, and should be 0.05Ω or less. Such a
capacitor will be larger than a ceramic capacitor and will
have a larger capacitance, because the capacitor must be
large to achieve low ESR. Table 2 lists several capacitor
vendors.
TPS Series
Ceramic
frequencies, generating audible noise. Since the LT3681
operates at a lower current limit during Burst Mode operation, the noise is typically very quiet to a casual ear.
If this is unacceptable, use a high performance tantalum
or electrolytic capacitor at the output.
A final precaution regarding ceramic capacitors concerns
the maximum input voltage rating of the LT3681. A ceramic
input capacitor combined with trace or cable inductance
forms a high quality (under damped) tank circuit. If the
LT3681 circuit is plugged into a live supply, the input voltage can ring to twice its nominal value, possibly exceeding
the LT3681’s rating. This situation is easily avoided (see
the Hot Plugging Safely section).
Catch Diode
The integral power Schottky catch diode conducts current
only during switch off time. Average forward current in
normal operation can be calculated from:
ID(AVG) = IOUT (VIN – VOUT)/VIN
where IOUT is the output load current.
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT3681 due to their piezoelectric nature.
When in Burst Mode operation, the LT3681’s switching
frequency depends on the load current, and at very light
loads the LT3681 can excite the ceramic capacitor at audio
Frequency Compensation
The LT3681 uses current mode control to regulate the
output. This simplifies loop compensation. In particular, the
LT3681 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
VC pin, as shown in Figure 2. Generally a capacitor (CC)
and a resistor (RC) in series to ground are used. In addition, there may be a lower value capacitor in parallel.
This capacitor (CF) is not part of the loop compensation
but is used to filter noise at the switching frequency, and
is required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
3681f
12
LT3681
APPLICATIONS INFORMATION
Loop compensation determines the stability and transient
performance. Designing the compensation network is a
bit complicated and the best values depend on the application and in particular the type of output capacitor. A
practical approach is to start with one of the circuits in
this data sheet that is similar to your application and tune
the compensation network to optimize the performance.
Stability should then be checked across all operating
conditions, including load current, input voltage and
temperature. The LT1375 data sheet contains a more
thorough discussion of loop compensation and describes
how to test the stability using a transient load. Figure 2
shows an equivalent circuit for the LT3681 control loop.
The error amplifier is a transconductance amplifier with
finite output impedance. The power section, consisting of
the modulator, power switch and inductor, is modeled as
a transconductance amplifier generating an output current proportional to the voltage at the VC pin. Note that
the output capacitor integrates this current, and that the
capacitor on the VC pin (CC) integrates the error amplifier output current, resulting in two poles in the loop. In
most cases a zero is required and comes from either the
output capacitor ESR or from a resistor RC in series with
CC. This simple model works well as long as the value
of the inductor is not too high and the loop crossover
frequency is much lower than the switching frequency.
A phase lead capacitor (CPL) across the feedback divider
may improve the transient response. Figure 3 shows the
transient response when the load current is stepped from
500mA to 1500mA and back to 500mA.
VIN = 12V
IL
1A/DIV
VOUT
100mV/DIV
10ms/DIV
3681 F03
Figure 3. Transient Load Response of the LT3681 3.3V Application
as the Load Current is Stepped from 500mA to 1500mA.
Burst Mode Operation
To enhance efficiency at light loads, the LT3681 automatically switches to Burst Mode operation which keeps
the output capacitor charged to the proper voltage while
minimizing the input quiescent current. During Burst Mode
operation, the LT3681 delivers single cycle bursts of current
to the output capacitor followed by sleep periods where
the output power is delivered to the load by the output
capacitor. In addition, VIN and BIAS quiescent currents are
reduced to typically 20µA and 50µA respectively during
the sleep time. As the load current decreases towards a
no load condition, the percentage of time that the LT3681
operates in sleep mode increases and the average input
current is greatly reduced resulting in higher efficiency.
See Figure 4.
LT3681
CURRENT MODE
POWER STAGE
gm = 3.5mho
SW
ERROR
AMPLIFIER
OUTPUT
R1
IL
0.5A/DIV
FB
–
gm =
330µmho
ESR
1.265V
C1
+
+
3Meg
C1
VC
CF
RC
VIN = 12V; VOUT = 3.3V
ILOAD = 10mA
CPL
POLYMER
OR
TANTALUM
GND
CERAMIC
VSW
5V/DIV
VOUT
10mV/DIV
R2
CC
5ms/DIV
3681 F04
3681 F02
Figure 2. Model for Loop Response
Figure 4. Burst Mode Operation
3681f
13
LT3681
APPLICATIONS INFORMATION
BOOST and BIAS Pin Considerations
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.22µF capacitor will work well. Figure 5 shows three
ways to arrange the boost circuit. The BOOST pin must be
more than 2.3V above the SW pin for best efficiency. For
outputs of 2.8V and above, the standard circuit (Figure 5a)
is best. For outputs between 2.8V and 3V, use a 1µF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1µF boost
capacitor (see Figure 5b). For lower output voltages the
boost diode can be tied to the input (Figure 5c), or to
another supply greater than 2.8V. The circuit in Figure 5a
is more efficient because the BOOST pin current and BIAS
pin quiescent current comes from a lower voltage source.
You must also be sure that the maximum voltage ratings
of the BOOST and BIAS pins are not exceeded.
The minimum operating voltage of an LT3681 application
is limited by the minimum input voltage (3.6V) and by the
maximum duty cycle as outlined in a previous section. For
proper startup, the minimum input voltage is also limited
by the boost circuit. If the input voltage is ramped slowly,
or the LT3681 is turned on with its RUN/SS pin when the
output is already in regulation, then the boost capacitor
may not be fully charged. Because the boost capacitor is
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load generally goes to zero
once the circuit has started. If, however, the LT3681 is
started by the RUN/SS pin and the output is discharged, the
discharged output capacitance will often present enough
of a load to allow the circuit to start. Figure 6 gives plots
of the input voltage required for three different situations:
the worst case situation where RUN/SS is tied to VIN and
VIN is ramped up very slowly, the minimum input voltage
at which the circuit will regulate when start-up is controlled
by RUN/SS, and the minimum input voltage required to
maintain output regulation. For lower start-up voltage, the
boost diode can be tied to VIN; however, this restricts the
input range to one-half of the absolute maximum rating
of the BOOST pin.
At light loads, the inductor current becomes discontinuous
and the effective duty cycle at the BOOST pin (not the SW
pin) can be very high. This reduces the minimum input
voltage to approximately 300mV above VOUT. At higher load
currents, the inductor current is continuous and the duty
cycle is limited by the maximum duty cycle of the LT3681,
requiring a higher input voltage to maintain regulation.
VOUT
BD
BOOST
VIN
VIN
4.7mF
LT3681
GND
DC
DA
C3
SW
COUT
(5a) For VOUT > 2.8V
VOUT
D2
BD
BOOST
VIN
VIN
4.7mF
LT3681
GND
DC
DA
C3
SW
COUT
(5b) For 2.5V < VOUT < 2.8V
VOUT
BD
BOOST
VIN
4.7mF
VIN
LT3681
GND
DA
DC
C3
SW
COUT
(5c) For VOUT < 2.5V
3681 FO5
Figure 5. Three Circuits For Generating The Boost Voltage
3681f
14
LT3681
APPLICATIONS INFORMATION
6.0
INPUT VOLTAGE (V)
5.5
5.0
Synchronization
VOUT = 3.3V
TA = 25°C
L = 4.7m
f = 800 kHz
The internal oscillator of the LT3681 can be synchronized
to an external 275kHz to 475kHz clock by using a 5pF
to 20pF capacitor to connect the clock signal to the RT
pin. The resistor tying the RT pin to ground should be
chosen such that the LT3681 oscillates 20% lower than
the intended synchronization frequency (see Setting the
Switching Frequency section).
4.5
4.0
3.5
3.0
2.5
2.0
0.001
TO START (RUN/SS = VIN)
TO START (RUN/SS CONTROL)
TO RUN
0.1
0.01
1
LOAD CURRENT (A)
10
The LT3681 should not be synchronized until its output
is near regulation as indicated by the PG flag. This can be
done with the system microcontroller/microprocessor or
with a discrete circuit by using the PG output. If a sync
signal is applied while the PG is low, the LT3681 may
exhibit erratic operation.
8.0
7.5
7.0
INPUT VOLTAGE (V)
6.5
VOUT = 5.0V
TA = 25°C
L = 4.7m
f = 800 kHz
6.0
5.5
5.0
4.5
4.0
3.5
3.0
2.5
2.0
0.001
TO START (RUN/SS = VIN)
TO START (RUN/SS CONTROL)
TO RUN
0.1
0.01
1
LOAD CURRENT (A)
10
3681 F06
Figure 6. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
Soft-Start
The RUN/SS pin can be used to soft-start the LT3681,
reducing the maximum input current during start-up.
The RUN/SS pin is driven through an external RC filter to
create a voltage ramp at this pin. Figure 7 shows the startup and shut-down waveforms with the soft-start circuit.
By choosing a large RC time constant, the peak start-up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20µA when the RUN/SS
pin reaches 2.3V.
IL
1A/DIV
RUN
15k
RUN/SS
0.22µF
VRUN/SS
2V/DIV
GND
VOUT
2V/DIV
2ms/DIV
When applying a sync signal, positive clock transitions
reset LT3681’s internal clock and negative transitions
initiate a switch cycle. The amplitude of the sync signal
must be at least 2V. The sync signal duty cycle can range
from 5% up to a maximum value given by the following
equation:
⎛
VOUT + VD ⎞
– f • 600ns
DCSYNC(MAX ) = ⎜ 1 –
VIN – VSW + VD ⎟⎠ SW
⎝
where VOUT is the programmed output voltage, VD is the
diode forward drop, VIN is the typical input voltage, VSW
is the switch drop, and fSW is the desired switching frequency. For example, a 24V input to 5V output at 300kHz
can be synchronized to a square wave with a maximum
duty cycle of 60%. For some applications, such as 12VIN
to 5VOUT at 350kHz, the maximum allowable sync duty
cycle will be less than 50%. If a low duty cycle clock cannot
be obtained from the system, then a one-shot should be
used between the sync signal and the LT3681. The value
of the coupling capacitor which connects the clock signal
to the RT pin should be chosen based on the clock signal
amplitude. Good starting values for 3.3V and 5V clock
signals are 10pF and 5pF, respectively. These values should
be tested and adjusted for each individual application to
assure reliable operation.
3681 F07
Figure 7. To Soft-Start the LT3681, Add a Resistor
and Capacitor to the RUN/SS Pin
3681f
15
LT3681
APPLICATIONS INFORMATION
Caution should be used when synchronizing more than
50% above the initial switching frequency (as set by the
RT resistor) because at higher clock frequencies the
amplitude of the internal slope compensation used to
prevent subharmonic switching is reduced. This type of
subharmonic switching only occurs at input voltages less
than twice output voltage. Higher inductor values will tend
to reduce this problem.
Reversed Input Protection
In some systems, the output may be held high when the
input to the LT3681 is absent. This may occur in battery
charging applications or in battery backup systems where
a battery or some other supply is diode ORed with the
LT3681’s output. If the VIN pin is allowed to float and the
RUN/SS pin is held high (either by a logic signal or because
it is tied to VIN), then the LT3681’s internal circuitry will
pull its quiescent current through its SW pin. This is fine
if your system can tolerate a few mA in this state. If you
ground the RUN/SS pin, the SW pin current will drop to
essentially zero. However, if the VIN pin is grounded while
the output is held high, then parasitic diodes inside the
LT3681 can pull large currents from the output through
the SW pin and the VIN pin. Figure 8 shows a circuit that
will run only when the input voltage is present and that
protects against a shorted or reversed input.
D4
MBRS140
VIN
VIN
BD BOOST
LT3681
RUN/SS
VC
GND DA
SW
VOUT
DC
FB
BACKUP
3681 F08
Figure 8. Diode D4 Prevents a Shorted Input from Discharging a Backup Battery Tied
to the Output. It Also Protects the Circuit from a Reversed Input. The LT3681 Runs Only
When the Input is Present
3681f
16
LT3681
APPLICATIONS INFORMATION
PCB Layout
For proper operation and minimum EMI, care must be taken
during printed circuit board layout. Figure 9 shows the
recommended component placement with trace, ground
plane and via locations. Note that large, switched currents
flow in the LT3681’s VIN and SW pins, the integrated
Schottky diode the input capacitor (CIN) and the output
capacitor (COUT). The loop formed by these components
should be as small as possible. These components,
along with the inductor and output capacitor, should be
placed on the same side of the circuit board, and their
connections should be made on that layer. Place a local,
unbroken ground plane below these components. The SW
and BOOST nodes should be as small as possible. Finally,
keep the FB and VC nodes small so that the ground traces
will shield them from the SW and BOOST nodes. Each of
the Exposed Pads on the bottom of the package must be
soldered to copper pours so that the pad acts as a heat
sink. To keep thermal resistance low, extend the ground
plane as much as possible, and add thermal vias under
and near the LT3681 to additional ground planes within
the circuit board and on the bottom side. Keep in mind
that the thermal design must keep the junctions of the IC
and power diode below the specified absolute maximum
temperature of 125°C.
1
14
2
13
3
12
4
11
5
10
6
9
7
8
VIN
CIN
COUT
3681 F11
VIAS TO GND
VIAS TO VIN
VIAS TO VOUT
VIAS TO DC HEATSINK
Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation
3681f
17
LT3681
APPLICATIONS INFORMATION
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3681 circuits. However, these
capacitors can cause problems if the LT3681 is plugged
into a live supply (see Linear Technology Application
Note 88 for a complete discussion). The low loss ceramic
capacitor, combined with stray inductance in series with
the power source, forms an under damped tank circuit,
and the voltage at the VIN pin of the LT3681 can ring
to twice the nominal input voltage, possibly exceeding
the LT3681’s rating and damaging the part. If the input
supply is poorly controlled or the user will be plugging
the LT3681 into an energized supply, the input network
should be designed to prevent this overshoot. Figure 10
shows the waveforms that result when an LT3681 circuit
is connected to a 24V supply through six feet of 24-gauge
twisted pair. The first plot (10a) is the response with a
4.7µF ceramic capacitor at the input. The input voltage
rings as high as 50V and the input current peaks at 26A.
A good solution is shown in Figure 10b. A 0.7Ω resistor
is added in series with the input to eliminate the voltage
overshoot (it also reduces the peak inrush current). A
0.1µF capacitor improves high frequency filtering. For
high input voltages its impact on efficiency is minor,
reducing efficiency by 1.5 percent for a 5V output at
full load operating from 24V. Another effective method
of reducing the overshoot is to add a 22µF aluminum
electrolytic capacitor, as shown in Figure 10c.
High Temperature Considerations
125°C. When operating at high ambient temperatures, the
maximum load current should be derated as the ambient
temperature approaches 125°C.
Power dissipation within the LT3681 can be estimated
by calculating the total power loss from an efficiency
measurement. The die temperature is calculated by
multiplying the LT3681 power dissipation by the thermal
resistance from junction to ambient.
Also keep in mind that the leakage current of the integrated
power Schottky diode, like all Schottky diodes, goes up
with junction temperature. The curves in Figure 11 show
how the leakage current in the power Schottky diode
varies with temperature and reverse voltage. When the
power switch is closed, the power Schottky diode is in
parallel with the power converter’s output filter stage. As
a result, an increase in a diode’s leakage current results
in an effective increase in the load, and a corresponding
increase in input power.
10000
LEAKAGE CURRENT (µA)
Hot Plugging Safely
VR = 10V
VR = 25V
VR = 40V
1000
100
10
1
–50
0
50
100
TEMPERATURE (°C)
150
3681 F12
The PCB must provide heat sinking to keep the LT3681
cool. The Exposed Pads on the bottom of the package
must be soldered to copper pours, which in turn should be
tied to large copper layers below with thermal vias; these
layers will spread the heat dissipated by the LT3681. Place
additional vias to reduce thermal resistance further. With
these steps, the thermal resistance from die (or junction)
to ambient can be reduced to θJA = 35°C/W or less. With
100 LFPM airflow, this resistance can fall by another 25%.
Further increases in airflow will lead to lower thermal resistance. Because of the large output current capability of the
LT3681, it is possible to dissipate enough power to raise
the junction temperature beyond the absolute maximum of
Figure 11. Like all Schottky Diodes, the LT3681 Integrated Power
Diode Leakage Current Varies with Temperature and Applied
Reverse Voltage VR.
Other Linear Technology Publications
Application Notes 19, 35, 44 and 76 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.
3681f
18
LT3681
APPLICATIONS INFORMATION
CLOSING SWITCH
SIMULATES HOT PLUG
IIN
VIN
DANGER
VIN
20V/DIV
RINGING VIN MAY EXCEED
ABSOLUTE MAXIMUM RATING
LT3681
+
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
4.7mF
IIN
10A/DIV
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
20ms/DIV
(10a)
0.7W
+
0.1mF
LT3681
VIN
20V/DIV
4.7mF
IIN
10A/DIV
(10b)
LT3681
+
22mF
35V
AI.EI.
+
20ms/DIV
VIN
20V/DIV
4.7mF
IIN
10A/DIV
(10c)
20ms/DIV
3681 F10
Figure 10. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation when the LT3681 is Connected to a Live Supply
3681f
19
LT3681
TYPICAL APPLICATIONS
5V Step-Down Converter
VOUT
5V
2A
VIN
6.3V TO 34V
VIN
ON OFF
BD
RUN/SS
BOOST
0.47mF
VC
4.7mF
L
6.8mH
SW
LT3681
DC
RT
DA
20k
PG
BIAS
590k
60.4k
FB
GND
330pF
22mF
200k
f = 800kHz
3681 TA02
L: NEC PLC-0745-6R8
3.3V Step-Down Converter
VOUT
3.3V
2A
VIN
4.4V TO 34V
BD
VIN
ON OFF
RUN/SS
BOOST
0.47mF
VC
4.7mF
LT3681
L
4.7mH
SW
DC
RT
DA
20k
BIAS
PG
324k
60.4k
GND
330pF
f = 800kHz
FB
22mF
200k
3681 TA03
L: NEC PLC-0745-4R7
3681f
20
LT3681
TYPICAL APPLICATIONS
2.5V Step-Down Converter
VOUT
2.5V
2A
VIN
4V TO 34V
VIN
BD
RUN/SS
ON OFF
D2
BOOST
L
4.7mH
1mF
VC
4.7mF
SW
LT3681
DC
RT
22k
DA
BIAS
PG
196k
84.5k
FB
GND
680pF
47mF
200k
f = 600kHz
3681 TA04
D2: MBR0540
L: SUMIDA CDRH8D43-4R7
5V, 2MHz Step-Down Converter
VIN
8.6V TO 22V
TRANSIENT TO 36V
VIN
ON OFF
VOUT
5V
2A
BD
RUN/SS
BOOST
0.47mF
VC
2.2mF
LT3681
L
2.2mH
SW
DC
RT
6.8k
DA
BIAS
PG
590k
16.9k
GND
470pF
f = 2MHz
FB
10mF
200k
3681 TA05
L: TOKO FDV0620-2R2
3681f
21
LT3681
TYPICAL APPLICATIONS
12V Step-Down Converter
VIN
15V TO 34V
BD
VIN
RUN/SS
ON OFF
VOUT
12V
2A
BOOST
0.47mF
VC
10mF
L
10mH
SW
LT3681
DC
RT
DA
24k
BIAS
PG
845k
60.4k
FB
GND
470pF
22mF
100k
f = 800kHz
3681 TA06
L: SUMIDA CDRH8D43-100
1.8V Step-Down Converter
VOUT
1.8V
2A
VIN
3.6V TO 27V
BD
VIN
RUN/SS
ON OFF
BOOST
0.47µF
VC
10µF
LT3681
L
3.3µH
SW
DC
RT
DA
15k
BIAS
PG
84.5k
105k
GND
330pF
f = 500kHz
FB
47µF
200k
3681 TA09
L: SUMIDA CDRH8D28-3R3
3681f
22
LT3681
PACKAGE DESCRIPTION
DE14MA Package
14-Lead Plastic DFN, Multichip (4mm × 3mm)
(Reference LTC DWG # 05-08-1731 Rev 0)
1.78 ±0.05
0.70 ±0.05
0.10 TYP
0.51 TYP
3.50 ±0.05
1.65 ± 0.05
2.10 ±0.05
1.07
±0.05
1.65 ± 0.05
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
3.00 REF
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ±0.10
(2 SIDES)
R = 0.05
TYP
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
8
1.78 ±0.10
14
1.07
±0.10
1.65 ± 0.10
0.10 TYP
0.51 TYP
1.65 ± 0.10
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
0.75 ±0.05
0.40 ± 0.10
7
1
0.25 ± 0.05
0.50 BSC
3.00 REF
0.00 – 0.05
PIN 1 NOTCH
R = 0.20 OR
0.25 × 45°
CHAMFER
(DE14MA) DFN 1106 REV Ø
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED IS NOT A JEDEC PACKAGE OUTLINE
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
3681f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
23
LT3681
U
TYPICAL APPLICATIO
1.265V Step-Down Converter
VOUT
1.265V
2A
VIN
3.6V TO 27V
BD
VIN
RUN/SS
ON OFF
BOOST
0.47mF
VC
4.7mF
L
3.3mH
SW
LT3681
DC
RT
DA
13k
BIAS
PG
105k
GND
FB
47mF
330pF
f = 500kHz
3681 TA10
L: NEC PLC-0745-3R3
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1766
60V, 1.2A (IOUT), 200kHz, High Efficiency Step-Down DC/DC
Converter
VIN = 5.5V to 60V, VOUT = 1.20V, IQ = 2.5mA, ISD 25μA
TSSOP16E Package
LT1767
25V, 1.2A (IOUT), 1.2MHz, High Efficiency Step-Down DC/DC
Converter
VIN = 3.0V to 25V, VOUT = 1.20V, IQ = 1mA, ISD < 6μA
MS8E Package
LT1933
500mA (IOUT), 500kHz Step-Down Switching Regulator in SOT-23
VIN = 3.6V to 36V, VOUT = 1.2V, IQ = 1.6mA, ISD < 1μA
ThinSOT Package
LT1936
36V, 1.4A (IOUT), 500kHz High Efficiency Step-Down DC/DC Converter VIN = 3.6V to 36V, VOUT = 1.2V, IQ = 1.9mA, ISD < 1μA
MS8E Package
LT1940
Dual 25V, 1.4A (IOUT), 1.1MHz, High Efficiency Step-Down DC/DC
Converter
VIN = 3.6V to 25V, VOUT = 1.20V, IQ = 3.8mA, ISD < 30μA
TSSOP16E Package
LT1976/LT1977
60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC
Converter with Burst Mode
VIN = 3.3V to 60V, VOUT = 1.20V, IQ = 100μA, ISD < 1μA
TSSOP16E Package
LT3434/LT3435
60V, 2.4A (IOUT), 200/500kHz, High Efficiency Step-Down DC/DC
Converter with Burst Mode
VIN = 3.3V to 60V, VOUT = 1.20V, IQ = 100μA, ISD < 1μA
TSSOP16E Package
LT3437
60V, 400mA (IOUT), MicroPower Step-Down DC/DC Converter with
Burst Mode
VIN = 3.3V to 60V, VOUT = 1.25V, IQ = 100μA, ISD < 1μA
(3mm × 3mm) DFN-10 TSSOP16E Package
LT3480
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN = 3.6V to 38V, VOUT = 0.78V, IQ = 70μA, ISD < 1μA
(3mm × 3mm) DFN-10 MSOP10E Package
LT3481
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High
Efficiency Step-Down DC/DC Converter with Burst Mode Operation
VIN = 3.6V to 34V, VOUT = 1.265V, IQ = 50μA, ISD < 1μA
(3mm × 3mm) DFN-10 MSOP10E Package
LT3493
36V, 1.4A (IOUT), 750kHz High Efficiency Step-Down DC/DC Converter VIN = 3.6V to 36V, VOUT = 0.8V, IQ = 1.9mA, ISD < 1μA
(2mm × 3mm)DFN-6 Package
LT3505
36V with Transient Protection to 40V, 1.4A (IOUT), 3MHz, High
Efficiency Step-Down DC/DC Converter
VIN = 3.6V to 34V, VOUT = 0.78V, IQ = 2mA, ISD < 2μA
(3mm × 3mm) DFN-8 MSOP8E Package
LT3508
36V with Transient Protection to 40V, Dual 1.4A (IOUT), 3MHz, High
Efficiency Step-Down DC/DC Converter
VIN = 3.7V to 37V, VOUT = 0.8V, IQ = 4.6mA, ISD < 1μA
(4mm × 4mm) QFN-24 TSSOP16E Package
LT3684
34V with Transient Protection to 36V, 2A (IOUT), 2.8MHz, High
Efficiency Step-Down DC/DC Converter
VIN = 3.6V to 34V, VOUT = 01.26V, IQ = 850mA, ISD < 1μA
(3mm × 3mm) DFN-10 MSOP10E Package
LT3685
36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High
Efficiency Step-Down DC/DC Converter
VIN = 3.6V to 38V, VOUT = 0.78V, IQ = 70mA, ISD < 1μA
(3mm × 3mm) DFN-10 MSOP10E Package
3681f
24 Linear Technology Corporation
LT 0407 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2006
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