LINER LT3758 Polyphase synchronous boost controller Datasheet

LTC3787
PolyPhase Synchronous
Boost Controller
FEATURES
DESCRIPTION
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The LTC®3787 is a high performance PolyPhase® single
output synchronous boost converter controller that drives
two N-channel power MOSFET stages out-of-phase.
Multiphase operation reduces input and output capacitor
requirements and allows the use of smaller inductors than
the single-phase equivalent. Synchronous rectification increases efficiency, reduces power losses and eases thermal
requirements, enabling high power boost applications.
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2-Phase Operation Reduces Required Input and
Output Capacitance and Power Supply Induced Noise
Synchronous Operation for Highest Efficiency and
Reduced Heat Dissipation
Wide VIN Range: 4.5V to 38V (40V Abs Max) and
Operates Down to 2.5V After Start-Up
Output Voltage Up to 60V
±1% 1.200V Reference Voltage
RSENSE or Inductor DCR Current Sensing
100% Duty Cycle Capability for Synchronous MOSFET
Low Quiescent Current: 135μA
Phase-Lockable Frequency (75kHz to 850kHz)
Programmable Fixed Frequency (50kHz to 900kHz)
Power Good Output Voltage Monitor
Low Shutdown Current, IQ < 8μA
Internal LDO Powers Gate Drive from VBIAS or EXTVCC
Thermally Enhanced Low Profile 28-Pin 4mm × 5mm
QFN Package and Narrow SSOP Package
The SS pin ramps the output voltage during start-up. The
PLLIN/MODE pin selects Burst Mode® operation, pulseskipping mode or forced continuous mode at light loads.
APPLICATIONS
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A 4.5V to 38V input supply range encompasses a wide
range of system architectures and battery chemistries.
When biased from the output of the boost converter or
another auxiliary supply, the LTC3787 can operate from
an input supply as low as 2.5V after start-up. The operating frequency can be set for a 50kHz to 900kHz range or
synchronized to an external clock using the internal PLL.
PolyPhase operation allows the LTC3787 to be configured
for 2-, 3-, 4-, 6- and 12-phase operation.
Industrial
Automotive
Medical
Military
L, LT, LTC, LTM, Linear Technology, the Linear logo, Burst Mode, OPTI-LOOP and PolyPhase
are registered trademarks and No RSENSE and ThinSOT are trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners. Protected by
U. S. Patents, including 5408150, 5481178, 5705919, 5929620, 6144194, 6177787, 6580258.
TYPICAL APPLICATION
Efficiency and Power Loss
vs Output Current
12V to 24V/10A 2-Phase Synchronous Boost Converter
VIN 4.5V TO 24V START-UP VOLTAGE
OPERATES THROUGH TRANSIENTS DOWN TO 2.5V
100
VIN
10000
90
BOOST1
TG2
BOOST2
0.1μF
SW1
80
47μF
3.3μH
0.1μF
VOUT
24V AT 10A
SW2
LTC3787
12.1k
100
50
40
10
VIN = 12V
1
VOUT = 24V
Burst Mode OPERATION
FIGURE 10 CIRCUIT
0
0.1
0.1
1
10
0.00001 0.0001 0.001 0.01
OUTPUT CURRENT (A)
20
10
15nF
8.66k
60
30
BG1
BG2
SENSE1+
–
SENSE1
SENSE2+
VFB
SENSE2–
FREQ
PGND
PLLIN/MODE
ITH SS SGND
232k
1000
70
POWER LOSS (mW)
TG1 VBIAS INTVCC
3.3μH
4mΩ
EFFICIENCY (%)
4.7μF
4.7μF
4mΩ
220μF
3787 TA01b
BURST EFFICIENCY
BURST LOSS
100pF
0.1μF
3787 TA01a
3787fc
1
LTC3787
ABSOLUTE MAXIMUM RATINGS
(Notes 1, 3)
VBIAS ........................................................ –0.3V to 40V
BOOST1 and BOOST2 ................................ –0.3V to 76V
SW1 and SW2............................................ –0.3V to 70V
RUN ............................................................. –0.3V to 8V
Maximum Current Sourced into Pin
From Source >8V ..............................................100μA
PGOOD, PLLIN/MODE ................................. –0.3V to 6V
INTVCC, (BOOST1 - SW1), (BOOST2 - SW2) ...–0.3V to 6V
EXTVCC ........................................................ –0.3V to 6V
SENSE1+, SENSE1–, SENSE2+, SENSE2– ... –0.3V to 40V
(SENSE1+ - SENSE1–), (SENSE2+ - SENSE2–) ...–0.3V to 0.3V
ILIM, SS, ITH, FREQ, PHASMD, VFB ..... –0.3V to INTVCC
Operating Junction Temperature
Range (Note 2)........................................–55°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
PIN CONFIGURATION
TOP VIEW
26 TG1
SGND
8
RUN
9
SS 10
23 VBIAS
CLKOUT 3
21 BG1
22 PGND
PLLIN/MODE 4
20 VBIAS
21 EXTVCC
19 PGND
29
GND
SGND 5
18 EXTVCC
17 INTVCC
RUN 6
20 INTVCC
16 BG2
SS 7
19 BG2
18 BOOST2
SENSE2+ 12
17 TG2
VFB 13
16 SW2
ITH 14
15 NC
15 BOOST2
9 10 11 12 13 14
SENSE2+
SENSE2– 11
SENSE2– 8
TG2
7
24 BG1
SW2
PLLIN/MODE
6
22 BOOST1
PHASMD 2
NC
CLKOUT
5
25 BOOST1
ITH
PHASMD
4
28 27 26 25 24 23
FREQ 1
VFB
FREQ
TG1
27 SW1
3
SW1
2
SENSE1–
PGOOD
SENSE1+
ILIM
28 PGOOD
SENSE1+
1
SENSE1–
TOP VIEW
ILIM
UFD PACKAGE
28-LEAD (4mm s 5mm) PLASTIC QFN
GN PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 43°C/W
EXPOSED PAD (PIN 29) IS GND, MUST BE CONNECTED TO GND
TJMAX = 125°C, θJA = 90°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3787EUFD#PBF
LTC3787EUFD#TRPBF
3787
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3787IUFD#PBF
LTC3787IUFD#TRPBF
3787
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 125°C
LTC3787HUFD#PBF
LTC3787HUFD#TRPBF
3787
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 150°C
LTC3787MPUFD#PBF
LTC3787MPUFD#TRPBF
3787
28-Lead (4mm × 5mm) Plastic QFN
–55°C to 150°C
LTC3787EGN#PBF
LTC3787EGN#TRPBF
LTC3787GN
28-Lead Plastic SSOP
–40°C to 125°C
LTC3787IGN#PBF
LTC3787IGN#TRPBF
LTC3787GN
28-Lead Plastic SSOP
–40°C to 125°C
LTC3787HGN#PBF
LTC3787HGN#TRPBF
LTC3787GN
28-Lead Plastic SSOP
–40°C to 150°C
LTC3787MPGN#PBF
LTC3787MPGN#TRPBF
LTC3787GN
28-Lead Plastic SSOP
–55°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
3787fc
2
LTC3787
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C, VBIAS = 12V, unless otherwise noted (Note 2).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
VBIAS
Chip Bias Voltage Operating Range
4.5
VFB
Regulated Feedback Voltage
ITH = 1.2V (Note 4)
IFB
Feedback Current
(Note 4)
VREFLNREG
Reference Line Voltage Regulation
VBIAS = 6V to 38V
VLOADREG
Output Voltage Load Regulation
(Note 4)
Measured in Servo Loop;
ΔITH Voltage = 1.2V to 0.7V
Measured in Servo Loop;
ΔITH Voltage = 1.2V to 2V
l
1.188
38
1.200
1.212
V
±5
±50
nA
0.002
0.02
%/V
l
0.01
0.1
%
l
–0.01
–0.1
%
gm
Error Amplifier Transconductance
ITH = 1.2V
IQ
Input DC Supply Current
Pulse-Skipping or Forced Continuous Mode
Sleep Mode
Shutdown
(Note 5)
RUN = 5V; VFB = 1.25V (No Load)
RUN = 5V; VFB = 1.25V (No Load)
RUN = 0V
INTVCC Undervoltage Lockout Thresholds
VINTVCC Ramping Up
VINTVCC Ramping Down
l
l
4.1
3.8
4.3
3.6
VRUN Rising
l
1.18
1.28
1.38
UVLO
VRUN
RUN Pin ON Threshold
VRUNHYS
RUN Pin Hysteresis
IRUNHYS
RUN Pin Hysteresis Current
IRUN
V
2
1.2
135
8
mmho
300
20
mA
μA
μA
V
V
V
100
mV
VRUN > 1.28V
4.5
μA
RUN Pin Current
VRUN < 1.28V
0.5
μA
ISS
Soft-Start Charge Current
VSS = GND
VSENSE1,2(MAX)
Maximum Current Sense Threshold
VFB = 1.1V, ILIM = INTVCC
VFB = 1.1V, ILIM = Float
VFB = 1.1V, ILIM = GND
VSENSE(MATCH)
Matching Between VSENSE1(MAX) and
VSENSE2(MAX)
VFB = 1.1V, ILIM = INTVCC
VFB = 1.1V, ILIM = Float
VFB = 1.1V, ILIM = GND
VSENSE(CM)
SENSE Pins Common Mode Range (BOOST
Converter Input Supply Voltage VIN)
ISENSE1,2+
SENSE+ Pin Current
VFB = 1.1V, ILIM = Float
ISENSE1,2–
SENSE– Pin Current
VFB = 1.1V, ILIM = Float
tr(TG1,2)
Top Gate Rise Time
CLOAD = 3300pF (Note 6)
20
ns
tf(TG1,2)
Top Gate Fall Time
CLOAD = 3300pF (Note 6)
20
ns
tr(BG1,2)
Bottom Gate Rise Time
CLOAD = 3300pF (Note 6)
20
ns
tr(BG1,2)
Bottom Gate Fall Time
CLOAD = 3300pF (Note 6)
20
ns
RUP(TG1,2)
Top Gate Pull-Up Resistance
1.2
Ω
RDN(TG1,2)
Top Gate Pull-Down Resistance
1.2
Ω
RUP(TG1,2)
Bottom Gate Pull-Up Resistance
1.2
Ω
RDN(TG1,2)
Bottom Gate Pull-Down Resistance
1.2
Ω
tD(TG/BG)
Top Gate Off to Bottom Gate On Switch-On
Delay Time
CLOAD = 3300pF (Each Driver)
70
ns
tD(BG/TG)
Bottom Gate Off to Top Gate On Switch-On
Delay Time
CLOAD = 3300pF (Each Driver)
70
ns
DFBG1,2(MAX)
Maximum BG Duty Factor
96
%
tON(MIN)
Minimum BG On-Time
7
10
13
μA
l
l
l
90
68
42
100
75
50
110
82
56
mV
mV
mV
l
l
l
–12
–10
–9
0
0
0
12
10
9
mV
mV
mV
38
V
300
μA
±1
μA
2.5
(Note 7)
200
110
ns
3787fc
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LTC3787
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the specified operating
junction temperature range, otherwise specifications are at TA = 25°C, VBIAS = 12V, unless otherwise noted (Note 2).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
5.2
5.4
5.6
V
0.5
2
%
5.2
5.4
5.6
V
0.5
2
%
4.5
4.8
5
V
INTVCC Linear Regulator
VINTVCC(VIN)
Internal VCC Voltage
6V < VBIAS < 38V, VEXTVCC = 0
VLDO INT
INTVCC Load Regulation
ICC = 0mA to 50mA
VINTVCC(EXT)
Internal VCC Voltage
VEXTVCC = 6V
VLDO EXT
INTVCC Load Regulation
ICC = 0mA to 40mA, VEXTVCC = 6V
VEXTVCC
EXTVCC Switchover Voltage
EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
l
250
mV
Oscillator and Phase-Locked Loop
fPROG
Programmable Frequency
RFREQ = 25k
RFREQ = 60k
RFREQ = 100k
335
105
400
760
465
kHz
kHz
kHz
fLOW
Lowest Fixed Frequency
VFREQ = 0V
320
350
380
kHz
fHIGH
Highest Fixed Frequency
VFREQ = INTVCC
488
535
585
kHz
fSYNC
Synchronizable Frequency
PLLIN/MODE = External Clock
850
kHz
VPGL
PGOOD Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD Leakage Current
VPGOOD = 5V
VPGOOD
PGOOD Trip Level
VFB with Respect to Set Regulated Voltage
l
75
PGOOD Output
tPGOOD(DELAY)
PGOOD Delay
0.2
0.4
V
±1
μA
VFB Ramping Negative
Hysteresis
–12
–10
2.5
–8
%
%
VFB Ramping Positive
Hysteresis
8
10
2.5
12
%
%
PGOOD Going High to Low
25
μs
VSW1,2 = 12V; VBOOST1,2 – VSW1,2 = 4.5V;
FREQ = 0V, Forced Continuous or
Pulse-Skipping Mode
55
μA
BOOST1 and BOOST2 Charge Pump
IBOOST1,2
BOOST Charge Pump Available Output
Current
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC3787 is tested under pulsed load conditions such that
TJ ≈ TA. The LTC3787E is guaranteed to meet specifications from
0°C to 85°C junction temperature. Specifications over the –40°C to
125°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC3787I is guaranteed over the –40°C to 125°C operating junction
temperature range, the LTC3787H is guaranteed over the –40°C to 150°C
operating temperature range and the LTC3787MP is tested and guaranteed
over the full –55°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes; operating lifetime
is derated for junction temperatures greater than 125°C. Note that the
maximum ambient temperature consistent with these specifications is
determined by specific operating conditions in conjunction with board
layout, the rated package thermal impedance and other environmental
factors. The junction temperature (TJ, in °C) is calculated from the ambient
temperature (TA, in °C) and power dissipation (PD, in Watts) according to
the formula: TJ = TA + (PD • θJA), where θJA = 43°C/W for the QFN package
and θJA = 90°C/W for the SSOP package.
Note 3: This IC includes overtemperature protection that is intended to
protect the device during momentary overload conditions. The maximum
rated junction temperature will be exceeded when this protection is active.
Continuous operation above the specified absolute maximum operating
junction temperature may impair device reliability or permanently damage
the device.
Note 4: The LTC3787 is tested in a feedback loop that servos VFB to the
output of the error amplifier while maintaining ITH at the midpoint of the
current limit range.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: see Minimum On-Time Considerations in the Applications
Information section.
3787fc
4
LTC3787
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency and Power Loss
vs Output Current
Efficiency and Power Loss
vs Output Current
100
10000
100
1000
80
90
EFFICIENCY (%)
50
10
40
30
20
1
VIN = 12V
VOUT = 24V
FIGURE 10 CIRCUIT
10
0
0.01
60
40
0.1
3787 G01
VIN = 12V
1
VOUT = 24V
10
Burst Mode OPERATION
FIGURE 10 CIRCUIT
0
0.1
0.1
1
10
0.00001 0.0001 0.001 0.01
OUTPUT CURRENT (A)
3787 G02
BURST EFFICIENCY
BURST LOSS
Load Step
Forced Continuous Mode
Efficiency vs Load Current
100
VIN = 12V
ILOAD = 2A
FIGURE 10 CIRCUIT
99
98
10
30
BURST LOSS
PULSE-SKIPPING
LOSS
FORCED CONTINUOUS
MODE LOSS
BURST EFFICIENCY
PULSE-SKIPPING
EFFICIENCY
FORCED CONTINUOUS
MODE EFFICIENCY
100
50
20
10
0.1
1
OUTPUT CURRENT (A)
1000
70
POWER LOSS (mW)
100
60
POWER LOSS (mW)
70
EFFICIENCY (%)
90
80
EFFICIENCY (%)
10000
LOAD STEP
5A/DIV
97
96
VOUT = 12V
VOUT = 24V
INDUCTOR
CURRENT
5A/DIV
95
94
93
VOUT
500mV/DIV
92
91
90
0
5
15
10
INPUT VOLTAGE (V)
20
200μs/DIV
VIN = 12V
VOUT = 24V
LOAD STEP FROM 100mA TO 5A
FIGURE 10 CIRCUIT
25
3787 G03
3787 G04
Load Step
Pulse-Skipping Mode
Load Step
Burst Mode Operation
LOAD STEP
5A/DIV
LOAD STEP
5A/DIV
INDUCTOR
CURRENT
5A/DIV
INDUCTOR
CURRENT
5A/DIV
VOUT
500mV/DIV
VOUT
500mV/DIV
200μs/DIV
VIN = 12V
VOUT = 24V
LOAD STEP FROM 100mA TO 5A
FIGURE 10 CIRCUIT
3787 G05
200μs/DIV
VIN = 12V
VOUT = 24V
LOAD STEP FROM 100mA TO 5A
FIGURE 10 CIRCUIT
3787 G06
3787fc
5
LTC3787
TYPICAL PERFORMANCE CHARACTERISTICS
Inductor Current at Light Load
Soft Start-Up
FORCED
CONTINUOUS MODE
VOUT
5V/DIV
Burst Mode
OPERATION
5A/DIV
PULSE-SKIPPING
MODE
0V
5μs/DIV
VIN = 12V
VOUT = 24V
ILOAD = 200μA
FIGURE 10 CIRCUIT
3787 G07
20ms/DIV
VIN = 12V
VOUT = 24V
FIGURE 10 CIRCUIT
Regulated Feedback Voltage
vs Temperature
Soft-Start Pull-Up Current
vs Temperature
11.0
1.212
1.209
SOFT-START CURRENT (μA)
REGULATED FEEDBACK VOLTAGE (V)
3787 G08
1.206
1.203
1.200
1.197
1.194
10.5
10.0
9.5
1.191
1.188
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G09
9.0
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G10
Shutdown Current vs Temperature
11.0
10.5
Shutdown Current vs Input Voltage
20
VIN = 12V
9.5
9.0
8.5
8.0
7.5
7.0
6.5
SHUTDOWN CURRENT (μA)
SHUTDOWN CURRENT (μA)
10.0
15
10
5
6.0
5.5
5.0
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G11
0
0
5
10
15 20 25 30
INPUT VOLTAGE (V)
35
40
3787 G12
3787fc
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LTC3787
TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown (RUN) Threshold
vs Temperature
Quiescent Current vs Temperature
1.40
180
1.35
RUN PIN VOLTAGE (V)
QUIESCENT CURRENT (μA)
VIN = 12V
VFB = 1.25V
170 RUN = GND
160
150
140
130
RUN RISING
1.30
1.25
1.20
RUN FALLING
1.15
120
1.10
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
110
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G14
3787 G13
Undervoltage Lockout Threshold
vs Temperature
INTVCC Line Regulation
4.4
5.5
4.3
5.4
INTVCC RISING
5.3
4.1
INTVCC VOLTAGE (V)
INTVCC VOLTAGE (V)
4.2
4.0
3.9
INTVCC FALLING
3.8
3.7
5.2
5.1
5.0
4.9
4.8
3.6
4.7
3.5
4.6
3.4
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G15
4.5
0
EXTVCC = 0V
5.30
5.25
5.20
EXTVCC = 6V
5.15
35
40
3787 G16
5.8
EXTVCC AND INTVCC VOLTAGE (V)
INTVCC VOLTAGE (V)
5.40
5.35
15 20 25 30
INPUT VOLTAGE (V)
6.0
VIN = 12V
5.45
10
EXTVCC Switchover and INTVCC
Voltages vs Temperature
INTVCC vs INTVCC Load Current
5.50
5
5.10
5.6
5.4
INTVCC
5.2
5.0
4.8
EXTVCC RISING
4.6
4.4
EXTVCC FALLING
4.2
5.05
5.00
0
20 40 60 80 100 120 140 160 180 200
INTVCC LOAD CURRENT (mA)
3787 G17
4.0
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G18
3787fc
7
LTC3787
TYPICAL PERFORMANCE CHARACTERISTICS
Oscillator Frequency
vs Temperature
Oscillator Frequency
vs Input Voltage
360
600
FREQ = INTVCC
OSCILLATOR FREQUENCY (kHz)
550
FREQUENCY (kHz)
FREQ = GND
358
500
450
400
FREQ = GND
350
356
354
352
350
348
346
344
342
340
300
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
5
10
20
25
30
15
INPUT VOLTAGE (V)
35
3787 G19
SENSE Pin Input Current
vs Temperature
120
100
PULSE-SKIPPING MODE
SENSE CURRENT (μA)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
Maximum Current Sense
Threshold vs ITH Voltage
80
Burst Mode
OPERATION
60
40
20
ILIM = GND
ILIM = FLOAT
ILIM = INTVCC
0
–20
FORCED CONTINUOUS MODE
–40
–60
0
0.2
0.4
0.6 0.8 1.0
ITH VOLTAGE (V)
1.2
1.4
260
VSENSE = 12V
240
ILIM = FLOAT
220
SENSE+ PIN
200
180
160
140
120
100
80
60
40
20
SENSE – PIN
0
–60 –35 –10 15 40 65 90 115 140
TEMPERATURE (°C)
3787 G22
3787 G21
SENSE Pin Input Current
vs VSENSE Voltage
VSENSE = 12V
SENSE+ PIN
ILIM = INTVCC
ILIM = FLOAT
SENSE CURRENT (μA)
SENSE CURRENT (μA)
SENSE Pin Input Current
vs ITH Voltage
260
240
220
200
180
160
140
120
100
80
60
40
20
0
ILIM = GND
SENSE – PIN
0
0.5
ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
2
1.5
1
ITH VOLTAGE (V)
40
3787 G20
2.5
3
3787 G23
260
240
220
200
180
160
140
120
100
80
60
40
20
0
SENSE+ PIN
ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
SENSE – PIN
2.5
ILIM = INTVCC
ILIM = FLOAT
ILIM = GND
7.5 12.5 17.5 22.5 27.5 32.5 37.5
VSENSE COMMON MODE VOLTAGE (V)
3787 G24
3787fc
8
LTC3787
TYPICAL PERFORMANCE CHARACTERISTICS
Charge Pump Charging Current
vs Operating Frequency
120
Charge Pump Charging Current
vs Switch Voltage
80
ILIM = INTVCC
100
ILIM = FLOAT
80
60
ILIM = GND
40
20
0
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
80
T = –60°C
70
T = –45°C
60
50
T = 25°C
40
T = 130°C
30
T = 155°C
20
10
0
50
150 250 350 450 550 650 750
OPERATING FREQUENCY (kHz)
3787 G25
PIN FUNCTIONS
CHARGE PUMP CHARGING CURRENT (μA)
CHARGE PUMP CHARGING CURRENT (μA)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
Maximum Current Sense
Threshold vs Duty Cycle
3787 G26
FREQ = GND
70
FREQ = INTVCC
60
50
40
30
20
10
0
5
10
15
25
30
20
SWITCH VOLTAGE (V)
35
40
3787 G27
(QFN/SSOP)
FREQ (Pin 1/Pin 4): Frequency Control Pin for the Internal
VCO. Connecting the pin to GND forces the VCO to a fixed
low frequency of 350kHz. Connecting the pin to INTVCC
forces the VCO to a fixed high frequency of 535kHz. The
frequency can be programmed from 50kHz to 900kHz
by connecting a resistor from the FREQ pin to GND. The
resistor and an internal 20μA source current create a voltage used by the internal oscillator to set the frequency.
Alternatively, this pin can be driven with a DC voltage to
vary the frequency of the internal oscillator.
PHASMD (Pin 2/Pin 5): This pin can be floated, tied to
SGND, or tied to INTVCC to program the phase relationship
between the rising edges of BG1 and BG2, as well as the
phase relationship between BG1 and CLKOUT.
CLKOUT (Pin 3/Pin 6): A Digital Output Used for Daisychaining Multiple LTC3787 ICs in Multiphase Systems. The
PHASMD pin voltage controls the relationship between BG1
and CLKOUT. This pin swings between SGND and INTVCC.
PLLIN/MODE (Pin 4/Pin 7): External Synchronization Input
to Phase Detector and Forced Continuous Mode Input.
When an external clock is applied to this pin, it will force
the controller into forced continuous mode of operation
and the phase-locked loop will force the rising BG1 signal
to be synchronized with the rising edge of the external
clock. When not synchronizing to an external clock, this
input determines how the LTC3787 operates at light loads.
Pulling this pin to ground selects Burst Mode operation.
An internal 100k resistor to ground also invokes Burst
Mode operation when the pin is floated. Tying this pin
to INTVCC forces continuous inductor current operation.
Tying this pin to a voltage greater than 1.2V and less than
INTVCC – 1.3V selects pulse-skipping operation. This can
be done by adding a 100k resistor between the PLLIN/
MODE pin and INTVCC.
SGND (Pin 5/Pin 8): Signal Ground. All small-signal
components and compensation components should
connect to this ground, which in turn connects to PGND
at a single point.
RUN (Pin 6/Pin 9): Run Control Input. Forcing this pin
below 1.28V shuts down the controller. Forcing this pin
below 0.7V shuts down the entire LTC3787, reducing
quiescent current to approximately 8μA. An external
resistor divider connected to VIN can set the threshold
for converter operation. Once running, a 4.5μA current is
sourced from the RUN pin allowing the user to program
hysteresis using the resistor values.
3787fc
9
LTC3787
PIN FUNCTIONS
(QFN/SSOP)
SS (Pin 7/Pin 10): Output Soft-Start Input. A capacitor to
ground at this pin sets the ramp rate of the output voltage
during start-up.
PGND (Pin 19/Pin 22): Driver Power Ground. Connects
to the sources of bottom (main) N-channel MOSFETs and
the (–) terminal(s) of CIN and COUT.
SENSE2– , SENSE1– (Pin 8, Pin 28/Pin 11, Pin 3): Negative Current Sense Comparator Input. The (–) input to the
current comparator is normally connected to the negative
terminal of a current sense resistor connected in series
with the inductor.
BG2, BG1 (Pin 16, Pin 21/Pin 19, Pin 24): Bottom Gate.
Connect to the gate of the main N-channel MOSFET.
SENSE2+, SENSE1+ (Pin 9, Pin 27/Pin 12, Pin 2): Positive Current Sense Comparator Input. The (+) input to the
current comparator is normally connected to the positive
terminal of a current sense resistor. The current sense resistor is normally placed at the input of the boost controller in
series with the inductor. This pin also supplies power to the
current comparator. The common mode voltage range on
SENSE+ and SENSE– pins is 2.5V to 38V (40V abs max).
VFB (Pin 10/Pin 13): Error Amplifier Feedback Input. This
pin receives the remotely sensed feedback voltage from
an external resistive divider connected across the output.
ITH (Pin 11/Pin 14): Current Control Threshold and Error
Amplifier Compensation Point. The voltage on this pin sets
the current trip threshold.
NC (Pin 12/Pin 15): No Connect.
SW2, SW1 (Pin 13, Pin 24/Pin 16, Pin 27): Switch Node.
Connect to the source of the synchronous N-channel
MOSFET, the drain of the main N-channel MOSFET and
the inductor.
TG2, TG1 (Pin 14, Pin 23/Pin 17, Pin 26): Top Gate. Connect to the gate of the synchronous N-channel MOSFET.
BOOST2, BOOST1 (Pin 15, Pin 22/Pin 18, Pin 25): Floating power supply for the synchronous N-channel MOSFET.
Bypass to SW with a capacitor and supply with a Schottky
diode connected to INTVCC.
INTVCC (Pin 17/Pin 20): Output of Internal 5.4V LDO.
Power supply for control circuits and gate drivers. Decouple this pin to GND with a minimum 4.7μF low ESR
ceramic capacitor.
EXTVCC (Pin 18/Pin 21): External Power Input. When this
pin is between 4.8V and 6V, an internal switch bypasses
the internal regulator and supply power to INTVCC directly
from EXTVCC. Do not float this pin. It can be connected to
ground when not used.
VBIAS (Pin 20/Pin 23): Main Supply Pin. It is normally
tied to the input supply VIN or to the output of the boost
converter. A bypass capacitor should be tied between this
pin and the signal ground pin. The operating voltage range
on this pin is 4.5V to 38V (40V abs max).
PGOOD (Pin 25/Pin 28): Power Good Indicator. Open-drain
logic output that is pulled to ground when the output voltage is more than ±10 % away from the regulated output
voltage. To avoid false trips the output voltage must be
outside the range for 25μs before this output is activated.
ILIM (Pin 26/Pin 1): Current Comparator Sense Voltage
Range Input. This pin is used to set the peak current
sense voltage in the current comparator. Connect this pin
to SGND, open, and INTVCC to set the peak current sense
voltage to 50mV, 75mV and 100mV, respectively.
GND (Exposed Pad Pin 29) UFD Package: Ground. Must
be soldered to the PCB for rated thermal performance.
3787fc
10
LTC3787
BLOCK DIAGRAM
PHASMD
INTVCC
CLKOUT
DUPLICATE FOR SECOND CONTROLLER CHANNEL
S
BOOST
DB
TG
CB
Q
R
SHDN
SWITCHING
LOGIC
AND
CHARGE
PUMP
20μA
FREQ
CLK2
VCO
0.425V
CLK1
+
COUT
INTVCC
BG
SLEEP
PGND
–
PFD
–
ICMP
+
– +
+
IREV
–
+ –
VOUT
SW
L
SENSE –
2mV
2.8V
0.7V
PLLIN/
MODE
SENSE+
SLOPE COMP
SYNC
DET
VIN
CIN
+
100k
SENS LO
–
2.5V
ILIM
VFB
CURRENT
LIMIT
+
EA –
–
VBIAS
SHDN
EXTVCC
EN
5.4V
LDO
EN
–
0.5μA/
4.5μA
+
10μA
3.8V
1.32V
11V
–
+
–
1.32V
ITH
PGOOD
CC
CC2
RC
–
+
4.8V
1.2V
SS
+
OV
5.4V
LDO
RSENSE
VFB
INTVCC
SGND
SHDN
SENS
LO
RUN
+
SS
1.08V
–
CSS
3787 BD
OPERATION
Main Control Loop
The LTC3787 uses a constant-frequency, current mode
step-up architecture with the two controller channels
operating out of phase. During normal operation, each
external bottom MOSFET is turned on when the clock for
that channel sets the RS latch, and is turned off when the
main current comparator, ICMP, resets the RS latch. The
peak inductor current at which ICMP trips and resets the
latch is controlled by the voltage on the ITH pin, which is
the output of the error amplifier EA. The error amplifier
compares the output voltage feedback signal at the VFB
pin (which is generated with an external resistor divider
connected across the output voltage, VOUT , to ground), to
the internal 1.200V reference voltage. In a boost converter,
the required inductor current is determined by the load
current, VIN and VOUT . When the load current increases,
it causes a slight decrease in VFB relative to the reference,
which causes the EA to increase the ITH voltage until the
average inductor current in each channel matches the new
requirement based on the new load current.
After the bottom MOSFET is turned off each cycle, the
top MOSFET is turned on until either the inductor current
starts to reverse, as indicated by the current comparator,
IR, or the beginning of the next clock cycle.
3787fc
11
LTC3787
OPERATION
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin.
When the EXTVCC pin is tied to a voltage less than 4.8V,
the VBIAS LDO (low dropout linear regulator) supplies
5.4V from VBIAS to INTVCC. If EXTVCC is taken above
4.8V, the VBIAS LDO is turned off and an EXTVCC LDO is
turned on. Once enabled, the EXTVCC LDO supplies 5.4V
from EXTVCC to INTVCC. Using the EXTVCC pin allows the
INTVCC power to be derived from an external source, thus
removing the power dissipation of the VBIAS LDO.
Shutdown and Start-Up (RUN and SS Pins)
The two internal controllers of the LTC3787 can be shut
down using the RUN pin. Pulling this pin below 1.28V
shuts down the main control loops for both phases.
Pulling this pin below 0.7V disables both controllers and
most internal circuits, including the INTVCC LDOs. In this
state, the LTC3787 draws only 8μA of quiescent current.
NOTE: Do not apply a heavy load for an extended time
while the chip is in shutdown. The top MOSFETs will be
turned off during shutdown and the output load may cause
excessive dissipation in the body diodes.
The RUN pin may be externally pulled up or driven directly
by logic. When driving the RUN pin with a low impedance
source, do not exceed the absolute maximum rating of
8V. The RUN pin has an internal 11V voltage clamp that
allows the RUN pin to be connected through a resistor to
a higher voltage (for example, VIN), as long as the maximum current into the RUN pin does not exceed 100μA.
An external resistor divider connected to VIN can set the
threshold for converter operation. Once running, a 4.5μA
current is sourced from the RUN pin allowing the user to
program hysteresis using the resistor values.
The start-up of the controller’s output voltage VOUT is
controlled by the voltage on the SS pin. When the voltage
on the SS pin is less than the 1.2V internal reference, the
LTC3787 regulates the VFB voltage to the SS pin voltage
instead of the 1.2V reference. This allows the SS pin to
be used to program a soft-start by connecting an external
capacitor from the SS pin to SGND. An internal 10μA
pull-up current charges this capacitor creating a voltage
ramp on the SS pin. As the SS voltage rises linearly from
0V to 1.2V (and beyond up to INTVCC), the output voltage
rises smoothly to its final value.
Light Load Current Operation—Burst Mode Operation,
Pulse-Skipping or Continuous Conduction
(PLLIN/MODE Pin)
The LTC3787 can be enabled to enter high efficiency
Burst Mode operation, constant-frequency, pulse-skipping
mode or forced continuous conduction mode at low
load currents. To select Burst Mode operation, tie the
PLLIN/MODE pin to ground (e.g., SGND). To select
forced continuous operation, tie the PLLIN/MODE pin to
INTVCC. To select pulse-skipping mode, tie the PLLIN/
MODE pin to a DC voltage greater than 1.2V and less
than INTVCC – 1.3V.
When the controller is enabled for Burst Mode operation, the minimum peak current in the inductor is set to
approximately 30% of the maximum sense voltage even
though the voltage on the ITH pin indicates a lower value.
If the average inductor current is higher than the required
current, the error amplifier EA will decrease the voltage
on the ITH pin. When the ITH voltage drops below 0.425V,
the internal sleep signal goes high (enabling sleep mode)
and both external MOSFETs are turned off.
In sleep mode much of the internal circuitry is turned off
and the LTC3787 draws only 135μA of quiescent current.
In sleep mode the load current is supplied by the output
capacitor. As the output voltage decreases, the EA’s output
begins to rise. When the output voltage drops enough, the
sleep signal goes low and the controller resumes normal
operation by turning on the bottom external MOSFET on
the next cycle of the internal oscillator.
When the controller is enabled for Burst Mode operation,
the inductor current is not allowed to reverse. The reverse
current comparator (IR) turns off the top external MOSFET
just before the inductor current reaches zero, preventing
it from reversing and going negative. Thus, the controller
operates in discontinuous current operation.
3787fc
12
LTC3787
OPERATION
In forced continuous operation or when clocked by an
external clock source to use the phase-locked loop (see
the Frequency Selection and Phase-Locked Loop section),
the inductor current is allowed to reverse at light loads or
under large transient conditions. The peak inductor current is determined by the voltage on the ITH pin, just as
in normal operation. In this mode, the efficiency at light
loads is lower than in Burst Mode operation. However,
continuous operation has the advantages of lower output
voltage ripple and less interference to audio circuitry, as
it maintains constant-frequency operation independent
of load current.
When the PLLIN/MODE pin is connected for pulse-skipping
mode, the LTC3787 operates in PWM pulse-skipping mode
at light loads. In this mode, constant-frequency operation
is maintained down to approximately 1% of designed
maximum output current. At very light loads, the current
comparator ICMP may remain tripped for several cycles
and force the external bottom MOSFET to stay off for
the same number of cycles (i.e., skipping pulses). The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audio noise and
reduced RF interference as compared to Burst Mode
operation. It provides higher low current efficiency than
forced continuous mode, but not nearly as high as Burst
Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ and PLLIN/MODE Pins)
The selection of switching frequency is a trade-off between
efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage.
The switching frequency of the LTC3787’s controllers can
be selected using the FREQ pin.
If the PLLIN/MODE pin is not being driven by an external
clock source, the FREQ pin can be tied to SGND, tied to
INTVCC, or programmed through an external resistor. Tying
FREQ to SGND selects 350kHz while tying FREQ to INTVCC
selects 535kHz. Placing a resistor between FREQ and SGND
allows the frequency to be programmed between 50kHz
and 900kHz, as shown in Figure 6.
A phase-locked loop (PLL) is available on the LTC3787
to synchronize the internal oscillator to an external clock
source that is connected to the PLLIN/MODE pin. The
LTC3787’s phase detector adjusts the voltage (through
an internal lowpass filter) of the VCO input to align the
turn-on of the first controller’s external bottom MOSFET
to the rising edge of the synchronizing signal. Thus, the
turn-on of the second controller’s external bottom MOSFET
is 180 or 240 degrees out-of-phase to the rising edge of
the external clock source.
The VCO input voltage is prebiased to the operating frequency set by the FREQ pin before the external clock is
applied. If prebiased near the external clock frequency,
the PLL loop only needs to make slight changes to the
VCO input in order to synchronize the rising edge of the
external clock’s to the rising edge of BG1. The ability to
prebias the loop filter allows the PLL to lock-in rapidly
without deviating far from the desired frequency.
The typical capture range of the LTC3787’s PLL is from
approximately 55kHz to 1MHz, and is guaranteed to lock
to an external clock source whose frequency is between
75kHz and 850kHz.
The typical input clock thresholds on the PLLIN/MODE
pin are 1.6V (rising) and 1.2V (falling).
PolyPhase Applications (CLKOUT and PHASMD Pins)
The LTC3787 features two pins, CLKOUT and PHASMD,
that allow other controller ICs to be daisychained with
the LTC3787 in PolyPhase applications. The clock output
signal on the CLKOUT pin can be used to synchronize
additional power stages in a multiphase power supply
solution feeding a single, high current output or multiple
separate outputs. The PHASMD pin is used to adjust the
phase of the CLKOUT signal as well as the relative phases
between the two internal controllers, as summarized in
Table 1. The phases are calculated relative to the zero
degrees phase being defined as the rising edge of the
bottom gate driver output of controller 1 (BG1). Depending on the phase selection, a PolyPhase application with
3787fc
13
LTC3787
OPERATION
multiple LTC3787s can be configured for 2-, 3-, 4- , 6- and
12-phase operation.
Power Good
CLKOUT is disabled when the controller is in shutdown
or in sleep mode.
The PGOOD pin is connected to an open drain of an
internal N-channel MOSFET. The MOSFET turns on and
pulls the PGOOD pin low when the VFB pin voltage is not
within ±10% of the 1.2V reference voltage. The PGOOD
pin is also pulled low when the corresponding RUN pin
is low (shut down). When the VFB pin voltage is within
the ±10% requirement, the MOSFET is turned off and the
pin is allowed to be pulled up by an external resistor to a
source of up to 6V (abs max).
Operation When VIN > Regulated VOUT
Operation at Low SENSE Pin Common Mode Voltage
When VIN rises above the regulated VOUT voltage, the boost
controller can behave differently depending on the mode,
inductor current and VIN voltage. In forced continuous
mode, the loop works to keep the top MOSFET on continuously once VIN rises above VOUT. The internal charge
pump delivers current to the boost capacitor to maintain
a sufficiently high TG voltage. The amount of current the
charge pump can deliver is characterized by two curves
in the Typical Performance Characteristics section.
The current comparator in the LTC3787 is powered directly
from the SENSE+ pin. This enables the common mode
voltage of the SENSE+ and SENSE– pins to operate at as
low as 2.5V, which is below the UVLO threshold. The figure
on the first page shows a typical application in which the
controller’s VBIAS is powered from VOUT while the VIN
supply can go as low as 2.5V. If the voltage on SENSE+
drops below 2.5V, the SS pin will be held low. When the
SENSE voltage returns to the normal operating range, the
SS pin will be released, initiating a new soft-start cycle.
Table 1.
VPHASMD
CONTROLLER 2 PHASE (°C)
CLKOUT PHASE (°C)
GND
180
60
Floating
180
90
INTVCC
240
120
In pulse-skipping mode, if VIN is between 100% and
110% of the regulated VOUT voltage, TG turns on if the
inductor current rises above a certain threshold and turns
off if the inductor current falls below this threshold. This
threshold current is set to approximately 6%, 4% or
3% of the maximum ILIM current when the ILIM pin is
grounded, floating or tied to INTVCC, respectively. If the
controller is programmed to Burst Mode operation under
this same VIN window, then TG remains off regardless of
the inductor current.
If VIN rises above 110% of the regulated VOUT voltage in
any mode, the controller turns on TG regardless of the
inductor current. In Burst Mode operation, however, the
internal charge pump turns off if the chip is asleep. With
the charge pump off, there would be nothing to prevent
the boost capacitor from discharging, resulting in an
insufficient TG voltage needed to keep the top MOSFET
completely on. To prevent excessive power dissipation
across the body diode of the top MOSFET in this situation,
the chip can be switched over to forced continuous mode
to enable the charge pump or a Schottky diode can also
be placed in parallel to the top MOSFET.
BOOST Supply Refresh and Internal Charge Pump
Each top MOSFET driver is biased from the floating
bootstrap capacitor, CB, which normally recharges during
each cycle through an external diode when the bottom
MOSFET turns on. There are two considerations for keeping the BOOST supply at the required bias level. During
start-up, if the bottom MOSFET is not turned on within
100μs after UVLO goes low, the bottom MOSFET will be
forced to turn on for ~400ns. This forced refresh generates enough BOOST-SW voltage to allow the top MOSFET
ready to be fully enhanced instead of waiting for the initial
few cycles to charge up. There is also an internal charge
pump that keeps the required bias on BOOST. The charge
pump always operates in both forced continuous mode
and pulse-skipping mode. In Burst Mode operation, the
charge pump is turned off during sleep and enabled when
the chip wakes up. The internal charge pump can normally
supply a charging current of 55μA.
3787fc
14
LTC3787
APPLICATIONS INFORMATION
The Typical Application on the first page is a basic LTC3787
application circuit. LTC3787 can be configured to use either
inductor DCR (DC resistance) sensing or a discrete sense
resistor (RSENSE) for current sensing. The choice between
the two current sensing schemes is largely a design tradeoff between cost, power consumption and accuracy. DCR
sensing is becoming popular because it does not require
current sensing resistors and is more power-efficient,
especially in high current applications. However, current
sensing resistors provide the most accurate current limits
for the controller. Other external component selection is
driven by the load requirement, and begins with the selection of RSENSE (if RSENSE is used) and inductor value.
Next, the power MOSFETs are selected. Finally, input and
output capacitors are selected. Note that the two controller channels of the LTC3787 should be designed with the
same components.
The SENSE+ pin also provides power to the current comparator. It draws ~200μA during normal operation. There
is a small base current of less than 1μA that flows into
the SENSE– pin. The high impedance SENSE– input to the
current comparators allows accurate DCR sensing.
Filter components mutual to the sense lines should be
placed close to the LTC3787, and the sense lines should
run close together to a Kelvin connection underneath the
current sense element (shown in Figure 1). Sensing current elsewhere can effectively add parasitic inductance
and capacitance to the current sense element, degrading
the information at the sense terminals and making the
programmed current limit unpredictable. If DCR sensing
is used (Figure 2b), sense resistor R1 should be placed
close to the switching node, to prevent noise from coupling
into sensitive small-signal nodes.
TO SENSE FILTER,
NEXT TO THE CONTROLLER
SENSE+ and SENSE– Pins
The SENSE+ and SENSE– pins are the inputs to the current comparators. The common mode input voltage range
of the current comparators is 2.5V to 38V. The current
sense resistor is normally placed at the input of the boost
controller in series with the inductor.
VBIAS
VIN
INDUCTOR OR RSENSE
3787 F01
Figure 1. Sense Lines Placement with
Inductor or Sense Resistor
VBIAS
VIN
VIN
SENSE+
SENSE+
C1
(OPTIONAL)
R2
DCR
SENSE–
SENSE–
INTVCC
INTVCC
R1
LTC3787
LTC3787
BOOST
BOOST
TG
TG
VOUT
SW
INDUCTOR
L
VOUT
SW
BG
BG
SGND
SGND
3787 F02b
3787 F02a
PLACE C1 NEAR SENSE PINS
(2a) Using a Resistor to Sense Current
(R1||R2) • C1 =
L
DCR
RSENSE(EQ) = DCR •
R2
R1 + R2
(2b) Using the Inductor DCR to Sense Current
Figure 2. Two Different Methods of Sensing Current
3787fc
15
LTC3787
APPLICATIONS INFORMATION
Sense Resistor Current Sensing
Inductor DCR Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 2a. RSENSE is chosen based on the required
output current.
For applications requiring the highest possible efficiency
at high load currents, the LTC3787 is capable of sensing
the voltage drop across the inductor DCR, as shown in
Figure 2b. The DCR of the inductor can be less than 1mΩ
for high current inductors. In a high current application
requiring such an inductor, conduction loss through a
sense resistor could reduce the efficiency by a few percent
compared to DCR sensing.
The current comparator has a maximum threshold
VSENSE(MAX). When the ILIM pin is grounded, floating or
tied to INTVCC, the maximum threshold is set to 50mV,
75mV or 100mV, respectively. The current comparator
threshold sets the peak of the inductor current, yielding
a maximum average inductor current, IMAX, equal to the
peak value less half the peak-to-peak ripple current, ΔIL.
To calculate the sense resistor value, use the equation:
RSENSE =
VSENSE(MAX)
ΔI
IMAX + L
2
The actual value of IMAX for each channel depends on the
required output current IOUT(MAX) and can be calculated
using:
⎛ IOUT(MAX) ⎞ ⎛ VOUT ⎞
IMAX = ⎜
⎟ • ⎜⎝ V ⎟⎠
2
⎝
⎠
IN
When using the controller in low VIN and very high voltage
output applications, the maximum inductor current and
correspondingly the maximum output current level will
be reduced due to the internal compensation required to
meet stability criterion for boost regulators operating at
greater than 50% duty factor. A curve is provided in the
Typical Performance Characteristics section to estimate
this reduction in peak inductor current level depending
upon the operating duty factor.
If the external R1||R2 • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature. Consult
the manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the inductor value calculation section, the target sense resistor
value is:
VSENSE(MAX)
ΔI
IMAX + L
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the maximum current sense threshold
(VSENSE(MAX)).
RSENSE(EQUIV) =
Next, determine the DCR of the inductor. Where provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of resistance, which is approximately 0.4%/°C.
A conservative value for the maximum inductor temperature
(TL(MAX)) is 100°C.
3787fc
16
LTC3787
APPLICATIONS INFORMATION
To scale the maximum inductor DCR to the desired sense
resistor value, use the divider ratio:
RD =
RSENSE(EQUIV)
DCRMAX at TL(MAX)
C1 is usually selected to be in the range of 0.1μF to 0.47μF.
This forces R1|| R2 to around 2k, reducing error that might
have been caused by the SENSE– pin’s ±1μA current.
The equivalent resistance R1|| R2 is scaled to the room
temperature inductance and maximum DCR:
R1||R2 =
L
(DCR at 20°C) • C1
The sense resistor values are:
R1=
R1• RD
R1||R2
; R2 =
RD
1− RD
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at VIN = 1/2VOUT :
(VOUT − VIN ) • VIN
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
PLOSS _ R1 =
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the
use of smaller inductor and capacitor values. Why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
of MOSFET gate charge and switching losses. Also, at
higher frequency the duty cycle of body diode conduction
is higher, which results in lower efficiency. In addition to
this basic trade-off, the effect of inductor value on ripple
current and low current operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current ΔIL decreases with higher
inductance or frequency and increases with higher VIN:
ΔIL =
VIN
f •L
⎛
VIN ⎞
⎜⎝ 1− V ⎟⎠
OUT
Accepting larger values of ΔIL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ΔIL = 0.3(IMAX). The maximum
ΔIL occurs at VIN = 1/2VOUT.
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
25% of the current limit determined by RSENSE. Lower
inductor values (higher ΔIL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease. Once the value of L is known, an
inductor with low DCR and low core losses should be
selected.
3787fc
17
LTC3787
APPLICATIONS INFORMATION
Power MOSFET Selection
Two external power MOSFETs must be selected for each
controller in the LTC3787: one N-channel MOSFET for the
bottom (main) switch, and one N-channel MOSFET for the
top (synchronous) switch.
The peak-to-peak gate drive levels are set by the INTVCC
voltage. This voltage is typically 5.4V during start-up
(see EXTVCC pin connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
Pay close attention to the BVDSS specification for the
MOSFETs as well; many of the logic level MOSFETs are
limited to 30V or less.
Selection criteria for the power MOSFETs include the
on-resistance RDS(ON), Miller capacitance CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturer’s data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result
is then multiplied by the ratio of the application applied
VDS to the gate charge curve specified VDS. When the IC
is operating in continuous mode, the duty cycles for the
top and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT − VIN
VOUT
Synchronous Switch Duty Cycle =
VIN
VOUT
If the maximum output current is IOUT(MAX) and each channel takes one half of the total output current, the MOSFET
power dissipations in each channel at maximum output
current are given by:
2
⎛ IOUT(MAX) ⎞
(V − V )V
PMAIN = OUT 2IN OUT • ⎜
⎟ • (1+ δ )
2
⎝
⎠
V IN
• RDS(ON) + k • V 3OUT •
IOUT(MAX)
2 • VIN
• CMILLER • f
V
PSYNC = IN
VOUT
2
⎛ IOUT(MAX) ⎞
•⎜
⎟ • (1+ δ ) • RDS(ON)
2
⎝
⎠
where δ is the temperature dependency of RDS(ON) (approximately 1Ω) is the effective driver resistance at the
MOSFET’s Miller threshold voltage. The constant k, which
accounts for the loss caused by reverse recovery current,
is inversely proportional to the gate drive current and has
an empirical value of 1.7.
Both MOSFETs have I2R losses while the bottom N-channel
equation includes an additional term for transition losses,
which are highest at low input voltages. For high VIN the
high current efficiency generally improves with larger
MOSFETs, while for low VIN the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the bottom switch duty factor is low or during overvoltage when the synchronous switch is on close
to 100% of the period.
The term (1+ δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
3787fc
18
LTC3787
APPLICATIONS INFORMATION
The input ripple current in a boost converter is relatively
low (compared with the output ripple current), because this
current is continuous. The input capacitor CIN voltage rating
should comfortably exceed the maximum input voltage.
Although ceramic capacitors can be relatively tolerant of
overvoltage conditions, aluminum electrolytic capacitors
are not. Be sure to characterize the input voltage for any
possible overvoltage transients that could apply excess
stress to the input capacitors.
The value of CIN is a function of the source impedance, and
in general, the higher the source impedance, the higher the
required input capacitance. The required amount of input
capacitance is also greatly affected by the duty cycle. High
output current applications that also experience high duty
cycles can place great demands on the input supply, both
in terms of DC current and ripple current.
In a boost converter, the output has a discontinuous current,
so COUT must be capable of reducing the output voltage
ripple. The effects of ESR (equivalent series resistance) and
the bulk capacitance must be considered when choosing
the right capacitor for a given output ripple voltage. The
steady ripple voltage due to charging and discharging
the bulk capacitance in a single phase boost converter
is given by:
VRIPPLE =
IOUT(MAX) • (VOUT − VIN(MIN) )
COUT • VOUT • f
V
where COUT is the output filter capacitor.
The steady ripple due to the voltage drop across the ESR
is given by:
ΔVESR = IL(MAX) • ESR
The LTC3787 is configured as a 2-phase single output
converter where the outputs of the two channels are
connected together and both channels have the same
duty cycle. With 2-phase operation, the two channels
are operated 180 degrees out-of-phase. This effectively
interleaves the output capacitor current pulses, greatly
reducing the output capacitor ripple current. As a result,
the ESR requirement of the capacitor can be relaxed.
Because the ripple current in the output capacitor is a
square wave, the ripple current requirements for the output
capacitor depend on the duty cycle, the number of phases
and the maximum output current. Figure 3 illustrates the
normalized output capacitor ripple current as a function of
duty cycle in a 2-phase configuration. To choose a ripple
current rating for the output capacitor, first establish the
duty cycle range based on the output voltage and range
of input voltage. Referring to Figure 3, choose the worstcase high normalized ripple current as a percentage of the
maximum load current.
Multiple capacitors placed in parallel may be needed to
meet the ESR and RMS current handling requirements.
Dry tantalum, special polymer, aluminum electrolytic and
ceramic capacitors are all available in surface mount
packages. Ceramic capacitors have excellent low ESR
characteristics but can have a high voltage coefficient.
Capacitors are now available with low ESR and high ripple
current ratings (e.g., OS-CON and POSCAP).
IORIPPLE /IOUT
CIN and COUT Selection
3.25
3.00
2.75
2.50
2.25
2.00
1.75
1.50
1.25
1.00
0.75
0.50
0.25
0
0.1
1-PHASE
2-PHASE
0.2
0.3 0.4 0.5 0.6 0.7 0.8
DUTY CYCLE OR (1-VIN /VOUT)
0.9
3787 F03
Figure 3. Normalized Output Capacitor Ripple
Current (RMS) for a Boost Converter
PolyPhase Operation
For output loads that demand high current, multiple
LTC3787s can be cascaded to run out-of-phase to provide
more output current and at the same time to reduce input
and output voltage ripple. The PLLIN/MODE pin allows the
LTC3787 to synchronize to the CLKOUT signal of another
LTC3787. The CLKOUT signal can be connected to the
PLLIN/MODE pin of the following LTC3787 stage to line
up both the frequency and the phase of the entire system.
3787fc
19
LTC3787
APPLICATIONS INFORMATION
Tying the PHASMD pin to INTVCC, SGND or floating
generates a phase difference (between PLLIN/MODE
and CLKOUT) of 240°, 60° or 90°, respectively, and a
phase difference (between CH1 and CH2) of 120°, 180°
0,240
VOUT
or 180°. Figure 4 shows the connections necessary for
3-, 4-, 6- or 12-phase operation. A total of 12 phases can
be cascaded to run simultaneously out-of-phase with
respect to each other.
120, CHANNEL 2 NOT USED
+120
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PHASMD
LTC3787
PHASMD
LTC3787
SS
RUN
VFB
SS
RUN
VFB
ITH
ITH
INTVCC
(4a) 3-Phase Operation
0,180
VOUT
90,270
+90
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PHASMD
LTC3787
PHASMD
LTC3787
SS
RUN
VFB
SS
RUN
VFB
ITH
ITH
(4b) 4-Phase Operation
0,180
VOUT
60,240
+60
120,300
+60
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PHASMD
LTC3787
PHASMD
LTC3787
PHASMD
LTC3787
SS
RUN
VFB
SS
RUN
VFB
ITH
SS
RUN
VFB
ITH
ITH
(4c) 6-Phase Operation
0,180
VOUT
60,240
+60
120,300
+60
+90
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PHASMD
LTC3787
PHASMD
LTC3787
PHASMD
LTC3787
SS
RUN
VFB
VFB
ITH
210,30
SS
RUN
VFB
ITH
270,90
+60
SS
RUN
ITH
330,150
+60
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PLLIN/MODE CLKOUT
PHASMD
LTC3787
PHASMD
LTC3787
PHASMD
LTC3787
SS
RUN
VFB
ITH
SS
RUN
VFB
ITH
SS
RUN
VFB
ITH
3787 F04
(4d) 12-Phase Operation
Figure 4. PolyPhase Operation
3787fc
20
LTC3787
APPLICATIONS INFORMATION
Setting Output Voltage
INTVCC Regulators
The LTC3787 output voltage is set by an external feedback
resistor divider carefully placed across the output, as shown
in Figure 5. The regulated output voltage is determined by:
The LTC3787 features two separate internal P-channel
low dropout linear regulators (LDO) that supply power at
the INTVCC pin from either the VBIAS supply pin or the
EXTVCC pin depending on the connection of the EXTVCC
pin. INTVCC powers the gate drivers and much of the
LTC3787’s internal circuitry. The VBIAS LDO and the
EXTVCC LDO regulate INTVCC to 5.4V. Each of these can
supply at least 50mA and must be bypassed to ground with
a minimum of 4.7μF ceramic capacitor. Good bypassing
is needed to supply the high transient currents required
by the MOSFET gate drivers and to prevent interaction
between the channels.
⎛ R ⎞
VOUT = 1.2V ⎜ 1+ B ⎟
⎝ RA ⎠
Great care should be taken to route the VFB line away
from noise sources, such as the inductor or the SW line.
Also keep the VFB node as small as possible to avoid
noise pickup.
VOUT
RB
LTC3787
VFB
RA
3787 F05
Figure 5. Setting Output Voltage
Soft-Start (SS Pin)
The start-up of VOUT is controlled by the voltage on the
SS pin. When the voltage on the SS pin is less than the
internal 1.2V reference, the LTC3787 regulates the VFB
pin voltage to the voltage on the SS pin instead of 1.2V.
Soft-start is enabled by simply connecting a capacitor from
the SS pin to ground, as shown in Figure 6. An internal
10μA current source charges the capacitor, providing a
linear ramping voltage at the SS pin. The LTC3787 will
regulate the VFB pin (and hence, VOUT) according to the
voltage on the SS pin, allowing VOUT to rise smoothly
from VIN to its final regulated value. The total soft-start
time will be approximately:
t SS = CSS •
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maximum junction temperature rating for the LTC3787 to be
exceeded. The INTVCC current, which is dominated by the
gate charge current, may be supplied by either the VBIAS
LDO or the EXTVCC LDO. When the voltage on the EXTVCC
pin is less than 4.8V, the VBIAS LDO is enabled. In this
case, power dissipation for the IC is highest and is equal
to VBIAS • IINTVCC. The gate charge current is dependent
on operating frequency, as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 3 of the
Electrical Characteristics. For example, at 70°C ambient
temperature, the LTC3787 INTVCC current is limited to less
than 32mA in the QFN package from a 40V VBIAS supply
when not using the EXTVCC supply:
TJ = 70°C + (32mA)(40V)(43°C/W) = 125°C
In an SSOP package, the INTVCC current is limited to
less than 15mA from a 40V supply when not using the
EXTVCC supply:
TJ = 70°C + (15mA)(40V)(90°C/W) = 125°C
1.2V
10µA
LTC3787
SS
CSS
SGND
3787 F06
Figure 6. Using the SS Pin to Program Soft-Start
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked while
operating in continuous conduction mode (PLLIN/MODE
= INTVCC) at maximum VIN.
When the voltage applied to EXTVCC rises above 4.8V, the
VIN LDO is turned off and the EXTVCC LDO is enabled. The
EXTVCC LDO remains on as long as the voltage applied to
3787fc
21
LTC3787
APPLICATIONS INFORMATION
EXTVCC remains above 4.55V. The EXTVCC LDO attempts
to regulate the INTVCC voltage to 5.4V, so while EXTVCC
is less than 5.4V, the LDO is in dropout and the INTVCC
voltage is approximately equal to EXTVCC. When EXTVCC
is greater than 5.4V, up to an absolute maximum of 6V,
INTVCC is regulated to 5.4V.
Significant thermal gains can be realized by powering
INTVCC from an external supply. Tying the EXTVCC pin
to a 5V supply reduces the junction temperature in the
previous example from 125°C to 79°C in a QFN package:
TJ = 70°C + (32mA)(5V)(43°C/W) = 77°C
and from 125°C to 74°C in an SSOP package:
TJ = 70°C + (15mA)(5V)(90°C/W) = 77°C
If more current is required through the EXTVCC LDO than
is specified, an external Schottky diode can be added between the EXTVCC and INTVCC pins. Make sure that in all
cases EXTVCC ≤ VBIAS (even at start-up and shutdown).
The following list summarizes possible connections for
EXTVCC:
EXTVCC Grounded. This will cause INTVCC to be powered
from the internal 5.4V regulator resulting in an efficiency
penalty at high input voltages.
EXTVCC Connected to an External Supply. If an external
supply is available in the 5V to 6V range, it may be used
to provide power. Ensure that EXTVCC is always lower
than VBIAS.
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors CB connected to the BOOST
pins supply the gate drive voltages for the topside
MOSFETs. Capacitor CB in the Block Diagram is charged
though external diode DB from INTVCC when the SW pin
is low. When one of the topside MOSFETs is to be turned
on, the driver places the CB voltage across the gate and
source of the desired MOSFET. This enhances the MOSFET
and turns on the topside switch. The switch node voltage, SW, rises to VOUT and the BOOST pin follows. With
the topside MOSFET on, the boost voltage is above the
output voltage: VBOOST = VOUT + VINTVCC. The value of
the boost capacitor CB needs to be 100 times that of the
total input capacitance of the topside MOSFET(s). The
reverse breakdown of the external Schottky diode must
be greater than VOUT(MAX).
The external diode DB can be a Schottky diode or silicon
diode, but in either case it should have low leakage and fast
recovery. Pay close attention to the reverse leakage at high
temperatures where it generally increases substantially.
Each of the topside MOSFET drivers includes an internal
charge pump that delivers current to the bootstrap capacitor from the BOOST pin. This charge current maintains
the bias voltage required to keep the top MOSFET on
continuously during dropout/overvoltage conditions. The
Schottky/silicon diodes selected for the topside drivers
should have a reverse leakage less than the available output
current the charge pump can supply. Curves displaying
the available charge pump current under different operating conditions can be found in the Typical Performance
Characteristics section.
A leaky diode DB in the boost converter can not only
prevent the top MOSFET from fully turning on but it can
also completely discharge the bootstrap capacitor CB and
create a current path from the input voltage to the BOOST
pin to INTVCC. This can cause INTVCC to rise if the diode
leakage exceeds the current consumption on INTVCC.
This is particularly a concern in Burst Mode operation
where the load on INTVCC can be very small. The external
Schottky or silicon diode should be carefully chosen such
that INTVCC never gets charged up much higher than its
normal regulation voltage.
Fault Conditions: Overtemperature Protection
At higher temperatures, or in cases where the internal
power dissipation causes excessive self heating on-chip
(such as an INTVCC short to ground), the overtemperature
shutdown circuitry will shut down the LTC3787. When the
junction temperature exceeds approximately 170°C, the
overtemperature circuitry disables the INTVCC LDO, causing
the INTVCC supply to collapse and effectively shut down
3787fc
22
LTC3787
APPLICATIONS INFORMATION
the entire LTC3787 chip. Once the junction temperature
drops back to approximately 155°C, the INTVCC LDO turns
back on. Long term overstress (TJ > 125°C) should be
avoided as it can degrade the performance or shorten
the life of the part.
Since the shutdown may occur at full load, beware that
the load current will result in high power dissipation in
the body diodes of the top MOSFETs. In this case, PGOOD
output may be used to turn the system load off.
Phase-Locked Loop and Frequency Synchronization
The LTC3787 has an internal phase-locked loop (PLL)
comprised of a phase frequency detector, a lowpass filter
and a voltage-controlled oscillator (VCO). This allows the
turn-on of the bottom MOSFET of channel 1 to be locked
to the rising edge of an external clock signal applied to
the PLLIN/MODE pin. The turn-on of channel 2’s bottom MOSFET is thus 180 degrees out-of-phase with the
external clock. The phase detector is an edge-sensitive
digital type that provides zero degrees phase shift between
the external and internal oscillators. This type of phase
detector does not exhibit false lock to harmonics of the
external clock.
If the external clock frequency is greater than the internal
oscillator’s frequency, fOSC, then current is sourced continuously from the phase detector output, pulling up the VCO
input. When the external clock frequency is less than fOSC,
current is sunk continuously, pulling down the VCO input.
If the external and internal frequencies are the same but
exhibit a phase difference, the current sources turn on for
an amount of time corresponding to the phase difference.
The voltage at the VCO input is adjusted until the phase
and frequency of the internal and external oscillators are
identical. At the stable operating point, the phase detector
output is high impedance and the internal filter capacitor,
CLP , holds the voltage at the VCO input.
Typically, the external clock (on the PLLIN/MODE pin) input
high threshold is 1.6V, while the input low threshold is 1.2V.
Note that the LTC3787 can only be synchronized to an
external clock whose frequency is within range of the
LTC3787’s internal VCO, which is nominally 55kHz to
1MHz. This is guaranteed to be between 75kHz and 850kHz.
Rapid phase locking can be achieved by using the FREQ pin
to set a free-running frequency near the desired synchronization frequency. The VCO’s input voltage is prebiased
at a frequency corresponding to the frequency set by the
FREQ pin. Once prebiased, the PLL only needs to adjust
the frequency slightly to achieve phase lock and synchronization. Although it is not required that the free-running
frequency be near external clock frequency, doing so will
prevent the operating frequency from passing through a
large range of frequencies as the PLL locks.
1000
900
FREQUENCY (kHz)
800
700
600
500
400
300
200
100
0
15 25 35 45 55 65 75 85 95 105 115 125
FREQ PIN RESISTOR (k)
3787 F07
Figure 7. Relationship Between Oscillator
Frequency and Resistor Value at the FREQ Pin
3787fc
23
LTC3787
APPLICATIONS INFORMATION
Table 2 summarizes the different states in which the FREQ
pin can be used.
INTVCC regulator current, 3) I2R losses, 4) bottom MOSFET transition losses, 5) body diode conduction losses.
Table 2.
1. The VBIAS current is the DC supply current given in the
Electrical Characteristics table, which excludes MOSFET
driver and control currents. VBIAS current typically
results in a small (<0.1%) loss.
FREQ PIN
PLLIN/MODE PIN
FREQUENCY
0V
DC Voltage
350kHz
INTVCC
DC Voltage
535kHz
Resistor
DC Voltage
50kHz to 900kHz
Any of the Above
External Clock
Phase Locked to
External Clock
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time duration
that the LTC3787 is capable of turning on the bottom
MOSFET. It is determined by internal timing delays and
the gate charge required to turn on the top MOSFET. Low
duty cycle applications may approach this minimum ontime limit.
In forced continuous mode, if the duty cycle falls below
what can be accommodated by the minimum on-time,
the controller will begin to skip cycles but the output will
continue to be regulated. More cycles will be skipped when
VIN increases. Once VIN rises above VOUT, the loop keeps
the top MOSFET continuously on. The minimum on-time
for the LTC3787 is approximately 110ns.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the greatest improvement. Percent efficiency
can be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc., are the individual losses as a percentage of input power.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge, dQ, moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
3. DC I2R losses. These arise from the resistances of the
MOSFETs, sensing resistor, inductor and PC board traces
and cause the efficiency to drop at high output currents.
4. Transition losses apply only to the bottom MOSFET(s),
and become significant only when operating at low
input voltages. Transition losses can be estimated from:
V OUT3 IOUT(MAX)
Transition Loss = (1.7)
•
•CRSS • f
VIN
2
5. Body diode conduction losses are more significant at
higher switching frequency. During the dead time, the loss
in the top MOSFETs is IOUT • VDS, where VDS is around
0.7V. At higher switching frequency, the dead time becomes a good percentage of switching cycle and causes
the efficiency to drop.
Other hidden losses, such as copper trace and internal
battery resistances, can account for an additional efficiency
degradation in portable systems. It is very important to
include these system-level losses during the design phase.
Although all dissipative elements in the circuit produce
losses, five main sources usually account for most of
the losses in LTC3787 circuits: 1) IC VBIAS current, 2)
3787fc
24
LTC3787
APPLICATIONS INFORMATION
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by an
amount equal to ΔILOAD(ESR), where ESR is the effective
series resistance of COUT . ΔILOAD also begins to charge or
discharge COUT generating the feedback error signal that
forces the regulator to adapt to the current change and
return VOUT to its steady-state value. During this recovery
time VOUT can be monitored for excessive overshoot or
ringing, which would indicate a stability problem. OPTILOOP compensation allows the transient response to be
optimized over a wide range of output capacitance and
ESR values. The availability of the ITH pin not only allows
optimization of control loop behavior, but it also provides
a DC coupled and AC filtered closed loop response test
point. The DC step, rise time and settling at this test
point truly reflects the closed loop response. Assuming a
predominantly second order system, phase margin and/
or damping factor can be estimated using the percentage
of overshoot seen at this pin. The bandwidth can also be
estimated by examining the rise time at the pin. The ITH
external components shown in the Figure 10 circuit will
provide an adequate starting point for most applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
to optimize transient response once the final PC layout
is complete and the particular output capacitor type and
value have been determined. The output capacitors must
be selected because the various types and values determine
the loop gain and phase. An output current pulse of 20%
to 80% of full-load current having a rise time of 1μs to
10μs will produce output voltage and ITH pin waveforms
that will give a sense of the overall loop stability without
breaking the feedback loop.
Placing a power MOSFET and load resistor directly across
the output capacitor and driving the gate with an appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is
in the feedback loop and is the filtered and compensated
control loop response.
The gain of the loop will be increased by increasing RC
and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that CC
is decreased, the zero frequency will be kept the same,
thereby keeping the phase shift the same in the most
critical frequency range of the feedback loop. The output
voltage settling behavior is related to the stability of the
closed-loop system and will demonstrate the actual overall
supply performance.
A second, more severe transient is caused by switching
in loads with large (>1μF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT , causing a rapid drop in VOUT . No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited to
approximately 25 • CLOAD. Thus, a 10μF capacitor would
require a 250μs rise time, limiting the charging current
to about 200mA.
Design Example
As a design example, assume VIN = 12V (nominal),
VIN = 22V (max), VOUT = 24V, IOUT(MAX) = 8A, VSENSE(MAX) =
75mV, and f = 350kHz.
The components are designed based on single channel
operation. The inductance value is chosen first based on
a 30% ripple current assumption. Tie the PLLIN/MODE
pin to GND, generating 350kHz operation. The minimum
inductance for 30% ripple current is:
VIN ⎛
VIN ⎞
1−
f • L ⎜⎝ VOUT ⎟⎠
The largest ripple happens when VIN = 1/2VOUT = 12V,
where the average maximum inductor current for each
channel is:
ΔIL =
⎛ IOUT(MAX) ⎞ ⎛ VOUT ⎞
IMAX = ⎜
⎟ • ⎜⎝ V ⎟⎠ = 8A
2
⎝
⎠
IN
3787fc
25
LTC3787
APPLICATIONS INFORMATION
A 6.8μH inductor will produce a 31% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 9.25A.
The RSENSE resistor value can be calculated by using the
maximum current sense voltage specification with some
accommodation for tolerances:
RSENSE ≤
75mV
= 0.008Ω
9.25A
Choosing 1% resistors: RA = 5k and RB = 95.3k yields an
output voltage of 24.072V.
The power dissipation on the top side MOSFET in each channel can be easily estimated. Choosing a Vishay Si7848BDP
MOSFET results in: RDS(ON) = 0.012Ω, CMILLER = 150pF.
At maximum input voltage with T (estimated) = 50°C:
PMAIN =
(24V – 12V) 24V
2
(12V)
• (4A)2
• ⎡⎣1+ (0.005)(50°C – 25°C)⎤⎦ • 0.008Ω
4A
(150pF)(350kHz) = 0.7W
12V
COUT is chosen to filter the square current in the output.
The maximum output current peak is:
+ (1.7)(24V)3
⎛ 31% ⎞
IOUT(PEAK) = 8 • ⎜ 1+
= 9.3A
⎝
2 ⎟⎠
A low ESR (5mΩ) capacitor is suggested. This capacitor
will limit output voltage ripple to 46.5mV (assuming ESR
dominate ripple).
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 8. Figure 9 illustrates the current
waveforms present in the various branches of the 2-phase
synchronous regulators operating in the continuous mode.
Check the following in your layout:
1. Put the bottom N-channel MOSFETs MBOT1 and MBOT2
and the top N-channel MOSFETs MTOP1 and MTOP2
in one compact area with COUT .
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return of
CINTVCC must return to the combined COUT (–) terminals.
The path formed by the bottom N-channel MOSFET
and the capacitor should have short leads and PC trace
lengths. The output capacitor (–) terminals should be
connected as close as possible to the source terminals
of the bottom MOSFETs.
3. Does the LTC3787 VFB pin’s resistive divider connect to
the (+) terminal of COUT? The resistive divider must be
connected between the (+) terminal of COUT and signal
ground and placed close to the VFB pin. The feedback
resistor connections should not be along the high current input feeds from the input capacitor(s).
4. Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the sense resistor.
5. Is the INTVCC decoupling capacitor connected close
to the IC, between the INTVCC and the power ground
pins? This capacitor carries the MOSFET drivers’ current peaks. An additional 1μF ceramic capacitor placed
immediately next to the INTVCC and PGND pins can help
improve noise performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2) and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from
the opposites channel’s voltage and current sensing
feedback pins. All of these nodes have very large and
fast moving signals and, therefore, should be kept on
the output side of the LTC3787 and occupy a minimal
PC trace area.
7. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on
the same side of the PC board as the input and output
capacitors with tie-ins for the bottom of the INTVCC
decoupling capacitor, the bottom of the voltage feedback
resistive divider and the SGND pin of the IC.
3787fc
26
LTC3787
APPLICATIONS INFORMATION
SENSE1–
SENSE1+
LTC3787
fIN
PHSMD
CLKOUT
FREQ
PLLIN/MODE
ILIM
PGOOD
SW1
TG1
VPULL-UP
L1
RSENSE1
CB1
BOOST1
+
M1
M2
BG1
VBIAS
+
GND
PGND
EXTVCC
INTVCC
SGND
RUN
VFB
ITH
BG2
CB2
M3
VIN
+
M4
VOUT
BOOST2
SS
TG2
SW2
SENSE2+
SENSE2–
L2
RSENSE2
3787 F08
Figure 8. Recommended Printed Circuit Layout Diagram
RSENSE1
L1
SW1
VOUT
VIN
RIN
CIN
COUT
RSENSE2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
RL
L2
SW2
3787 F09
Figure 9. Branch Current Waveforms
3787fc
27
LTC3787
APPLICATIONS INFORMATION
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output voltage. Check for proper performance over the operating
voltage and current range expected in the application.
The frequency of operation should be maintained over the
input voltage range down to dropout and until the output
load drops below the low current operation threshold—
typically 10% of the maximum designed current level in
Burst Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator
bandwidth optimization is not required. Only after each
controller is checked for its individual performance should
both controllers be turned on at the same time. A particularly difficult region of operation is when one controller
channel is nearing its current comparator trip point while
the other channel is turning on its bottom MOSFET. This
occurs around the 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
Reduce VIN from its nominal level to verify operation with
high duty cycle. Check the operation of the undervoltage
lockout circuit by further lowering VIN while monitoring
the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltages and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling.
An embarrassing problem which can be missed in an otherwise properly working switching regulator, results when
the current sensing leads are hooked up backwards. The
output voltage under this improper hook-up will still be
maintained, but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
3787fc
28
LTC3787
TYPICAL APPLICATIONS
SENSE1–
PGOOD
SENSE1+
ILIM
TG1
PHASMD
SW1
CLKOUT
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
INTVCC
MTOP1
SS
L1
3.3μH
RSENSE1
4mΩ
COUTB1
220μF
MBOT1
CINT
4.7μF
CIN
22μF
s2
D2
ITH
+
D1
PGND
RITH, 8.66k
COUTA1
22μF
s4
CB1, 0.1μF
VBIAS
INTVCC
FREQ
CSS, 0.1μF
CITH, 15nF
100k
BG2
VIN
5V TO 24V
VOUT
24V, 10A*
MBOT2
CB2, 0.1μF
CITHA, 220pF
L2
3.3μH
BOOST2
RSENSE2
4mΩ
SW2
RA, 12.1k
RB
232k
VFB
SENSE2+
SENSE2–
TG2
MTOP2
COUTA2
22μF
s4
+
COUTA1
6.8μF
×4
+
COUTB2
220μF
3787 F10
CIN, COUTA1, COUTA2: TDK C4532X5R1E226M
COUTB1, COUTB2: SANYO, 50CE220LX
L1, L2: PULSE PA1494.362NL
MBOT1, MBOT2, MTOP1, MTOP2: RENESAS HAT2169H
D1, D2: BAS140W
*WHEN VIN < 8V, MAXIMUM LOAD CURRENT AVAILABLE IS REDUCED.
Figure 10. High Efficiency 2-Phase 24V Boost Converter
SENSE1–
PGOOD
SENSE1+
TG1
ILIM
PHASMD
SW1
CLKOUT
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
INTVCC
MTOP1
SS
L1
3.3μH
RSENSE1
4mΩ
MBOT1
D1
CINT
4.7μF
CIN
6.8μF
×4
PGND
D2
RITH, 8.66k
ITH
COUTB1
220μF
CB1, 0.1μF
VBIAS
INTVCC
FREQ
CSS, 0.1μF
CITH, 15nF
100k
BG2
VIN
5V TO 28V
VOUT
28V, 8A
MBOT2
CB2, 0.1μF
CITHA, 220pF
L2
3.3μH
BOOST2
RSENSE2
4mΩ
SW2
RA, 12.1k
RB
271k
VFB
SENSE2+
SENSE2–
TG2
CIN, COUTA1, COUTA2: TDK C4532X7RIH685K
COUTB1, COUTB2: SANYO, 50CE220LX
L1, L2: PULSE PA1494.362NL
MBOT1, MBOT2, MTOP1, MTOP2: RENESAS HAT2169H
D1, D2: BAS140W
MTOP2
COUTA2
6.8μF
×4
+
COUTB2
220μF
3787 F11
Figure 11. High Efficiency 2-Phase 28V Boost Converter
3787fc
29
LTC3787
TYPICAL APPLICATIONS
SENSE1–
SENSE1+
ILIM
100k
PGOOD
TG1
PHASMD
SW1
CLKOUT
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
MTOP1
SS
L1
10.2μH
RSENSE1
5mΩ
COUTB1
220μF
MBOT1
CINT
4.7μF
CIN
6.8μF
×4
D2
ITH
+
D1
PGND
RITH, 3.57k
COUTA1
6.8μF
×4
CB1, 0.1μF
VBIAS
INTVCC
FREQ
CSS, 0.1μF
CITH, 15nF
INTVCC
BG2
VIN
5V TO 36V
VOUT
36V, 6A
MBOT2
CB2, 0.1μF
CITHA, 220pF
L2
10.2μH
BOOST2
RSENSE2
5mΩ
SW2
RA, 12.1k
RB
348k
VFB
SENSE2+
SENSE2–
TG2
MTOP2
COUTA2
6.8μF
×4
+
COUTA1
6.8μF
×4
+
COUTB2
220μF
3787 F12
CIN, COUTA1, COUTA2: TDK C4532X7RIH685K
COUTB1, COUTB2: SANYO, 50CE220LX
L1, L2: PULSE PA2050.103NL
MBOT1, MBOT2, MTOP1, MTOP2: RENESAS RJICO652DPB
D1, D2: BAS170W
Figure 12. High Efficiency 2-Phase 36V Boost Converter
SENSE1–
SENSE1+
ILIM
CSS, 0.1μF
PHASMD
SW1
CLKOUT
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
VBIAS
FREQ
INTVCC
SS
CITH, 10nF
100k
PGOOD
TG1
INTVCC
MTOP1
L1
16μH
RSENSE1
8mΩ
CB1, 0.1μF
MBOT1
D1
CINT
4.7μF
CIN
6.8μF
×4
PGND
D2
RITH, 23.7k
ITH
COUTB1
220μF
BG2
VIN
5V TO 38V
VOUT
48V, 4A
MBOT2
CB2, 0.1μF
CITHA, 220pF
L2
16μH
BOOST2
RSENSE2
8mΩ
SW2
RA, 12.1k
RB
475k
VFB
SENSE2+
SENSE2–
TG2
MTOP2
COUTA2
6.8μF
×4
+
COUTB2
220μF
3787 F13
CIN, COUTA1, COUTA2: TDK C4532X7RIH685K
COUTB1, COUTB2: SANYO, 63CE220K
L1, L2: PULSE PA2050.163NL
MBOT1, MBOT2, MTOP1, MTOP2: RENESAS RJK0652DPB
D1, D2: BAS170W
Figure 13. High Efficiency 2-Phase 48V Boost Converter
3787fc
30
LTC3787
TYPICAL APPLICATIONS
RS1, 53.6, 1%
RS2
26.1k
1%
C1
0.1μF
C3
0.1μF
INTVCC
RFREQ, 41.2k
SENSE1–
SENSE1+
ILIM
PHASMD
SW1
CLKOUT
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
FREQ
CSS, 0.1μF
SS
CITH, 15nF
100k
PGOOD
TG1
INTVCC
MTOP1
D3
L1
10.2μH
COUTA1
6.8μF
s4
+
COUTB1
220μF
CB1, 0.1μF
MBOT1
VIN
5V TO 24V
D1
VBIAS
INTVCC
CINT
4.7μF
PGND
CINA
22μF
s4
+
COUTA2
6.8μF
s4
+
VOUT
24V, 8A
CINB
220μF
D2
RITH, 8.87k, 1%
ITH
BG2
MBOT2
CB2, 0.1μF
CITHA, 220pF
L2
10.2μH
BOOST2
RA
12.1k, 1%
SW2
RS
232k
1%
RS4
26.1k
1%
VFB
TG2
MTOP2
D4
C4
0.1μF
SENSE2+
C2
0.1μF
SENSE2–
COUTB2
220μF
RS3, 53.6k, 1%
3787 F14
COUTA1, COUTA2: C4532x7R1H685K
COUTB1, COUTB2: SANYO 63CE220KX
CINA: TDK C4532X5R1E226M
CINB: SANYO 50CE220AX
L1, L2: PULSE PA2050.103NL
MBOT1, MBOT2, MTOP1, MTOP2: RENESAS RJK0305
D1, D2: BAS140W
D3, D4: DIODES INC. B340B
Figure 14. High Efficiency 2-Phase 24V Boost Converter with Inductor DCR Current Sensing
3787fc
31
LTC3787
TYPICAL APPLICATIONS
SENSE1+
SENSE1–
100k
PGOOD
TG1
INTVCC
ILIM
SW1
PHASMD
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
FREQ
CSS, 0.1μF
SS
CITH, 15nF
INTVCC
MTOP1
L1
3.3μH
RSENSE1
4mΩ
L2
3.3μH
RSENSE2
4mΩ
COUTA1
22μF
s4
+
COUTA2
22μF
s4
+
CINA
22μF
s4
+
COUTA3
22μF
s4
+
COUTA4
22μF
s4
+
COUTB1
220μF
CB1, 0.1μF
MBOT1
D1
VBIAS
INTVCC
CINT1
4.7μF
PGND
D2
RITH, 8.66k
ITH
BG2
MBOT2
CB2, 0.1μF
CITHA, 220pF
BOOST2
SW2
RA, 12.1k
VFB
RB
232k
CLKOUT
TG2
MTOP2
SENSE2–
SENSE2+
SENSE1+
SENSE1–
100k
PGOOD
TG1
ILIM
MTOP3
L3
3.3μH
RSENSE3
4mΩ
L4
3.3μH
RSENSE4
4mΩ
SW1
PHASMD
PLLIN/MODE BOOST1
SGND
BG1
EXTVCC
LTC3787
RUN
FREQ
INTVCC
COUTB2
220μF
VIN
5V to 24V
CINB
220μF
VOUT
24V, 20A*
COUTB3
220μF
CB3, 0.1μF
MBOT3
D3
VBIAS
INTVCC
CINT2
4.7μF
PGND
D4
SS
BG2
MBOT4
CB4, 0.1μF
ITH
BOOST2
SW2
VFB
CLKOUT
TG2
MTOP4
SENSE2–
SENSE2+
3787 F15
COUTB4
220μF
CINA, COUTA1, COUTA2, COUTA3, COUTA4: TDK C4532X5R1E226M
CINB, COUTB1, COUTB2, COUTB3, COUTB4: SANYO, 50CE220LX
L1, L2, L3, L4: PULSE PA1494.362NL
MBOT1, MBOT2, MBOT3, MBOT4, MTOP1, MTOP2, MTOP3, MTOP4: RENESAS HAT2169H
D1, D2, D3, D4: BAS140W
*WHEN VIN < 8V, MAXIMUM LOAD CURRENT AVAILABLE IS REDUCED.
Figure 15. 4-Phase Single Output Boost Converter
3787fc
32
LTC3787
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641 Rev B)
.386 – .393*
(9.804 – 9.982)
.045 ±.005
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.033
(0.838)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ±.0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 ±.004
w 45°
(0.38 ±0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
.0532 – .0688
(1.35 – 1.75)
9 10 11 12 13 14
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN28 REV B 0212
3. DRAWING NOT TO SCALE
4. PIN 1 CAN BE BEVEL EDGE OR A DIMPLE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
3787fc
33
LTC3787
PACKAGE DESCRIPTION
Please refer to http://www.linear.com/designtools/packaging/ for the most recent package drawings.
UFD Package
28-Lead Plastic QFN (4mm × 5mm)
(Reference LTC DWG # 05-08-1712 Rev B)
0.70 ±0.05
4.50 ± 0.05
3.10 ± 0.05
2.50 REF
2.65 ± 0.05
3.65 ± 0.05
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
3.50 REF
4.10 ± 0.05
5.50 ± 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ± 0.10
(2 SIDES)
0.75 ± 0.05
R = 0.05
TYP
PIN 1 NOTCH
R = 0.20 OR 0.35
× 45° CHAMFER
2.50 REF
R = 0.115
TYP
27
28
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
5.00 ± 0.10
(2 SIDES)
3.50 REF
3.65 ± 0.10
2.65 ± 0.10
(UFD28) QFN 0506 REV B
0.200 REF
0.00 – 0.05
0.25 ± 0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WXXX-X).
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
3787fc
34
LTC3787
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
12/10
Updated PGND, BG2, BG1, INTVCC and EXTVCC Pin numbers
Updated Block Diagram
Updated Figures 11, 12, 13
Updated Related Parts
B
C
9/11
4/12
Updated graphs on TA01b, G02, G09, G10, G11, G13, G14, G15, G18, G19, G22, and G26.
10
11
29, 30
36
1, 5, 6, 7, 8, 9
Updated the Storage Temperature Range
2
Updated Topside MOSFET Driver Supply (CB, DB) section
22
Updated Related Parts List
36
Added H and MP grades
2, 4
3787fc
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC3787
TYPICAL APPLICATION
IIN
CIN
12V
I1
BG1
TG1
PHASMD
0°
I1
BOOST: 24V, 5A
LTC3787
I2
I2
BG2
TG2
CLKOUT
+90°
180°
BOOST: 24V, 5A
I3
24V, 20A
I3
90,270
CLKOUT
PHASMD
BG1
TG1
I4
90°
BOOST: 24V, 5A
COUT
ICOUT
LTC3787
I4
BG2
TG2
I*IN
270°
BOOST: 24V, 5A
I*COUT
REFER TO FIGURE 15 FOR APPLICATION CIRCUITS
* RIPPLE CURRENT CANCELLATION INCREASES THE RIPPLE
FREQUENCY AND REDUCES THE RMS INPUT/OUTPUT RIPPLE
CURRENT, THUS SAVING INPUT/OUTPUT CAPACITORS
3787 F16
Figure 16. PolyPhase Application
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC3788/LTC3788-1
Multiphase, Dual Output Synchronous Step-Up
Controller
4.5V (Down to 2.5V After Start-Up) ≤ VIN ≤ 38V, VOUT Up to 60V, 50kHz to
900kHz Fixed Operating Frequency, 5mm × 5mm QFN-32, SSOP-28
LTC3786
Low IQ Synchronous Step-Up Controller
4.5V (Down to 2.5V After Start-Up) ≤ VIN ≤ 38V, VOUT Up to 60V, 50kHz to
900kHz Fixed Operating Frequency, 3mm × 3mm QFN-32, MSOP-16E
LTC3862/LTC3862-1
Multiphase, Dual Channel Single Output Current
Mode Step-Up DC/DC Controller
4V ≤ VIN ≤ 36V, 5V or 10V Gate Drive, 75kHz to 500kHz Fixed Operating
Frequency, SSOP-24, TSSOP-24, 5mm × 5mm QFN-24
LT3757/LT3758
Boost, Flyback, SEPIC and Inverting Controller
2.9V ≤ VIN ≤ 40V/100V, 100kHz to 1MHz Fixed Operating Frequency,
3mm × 3mm DFN-10 and MSOP-10E
LTC1871/LTC1871-1/
LTC1871-7
Wide Input Range, No RSENSE Low Quiescent Current
Flyback, Boost and SEPIC Controller
2.5V ≤ VIN ≤ 36V, 50kHz to 1MHz Fixed Operating Frequency, IQ = 250μA ,
MSOP-10
LTC3859
Low IQ, Triple Output Buck/Buck/Boost Synchronous
DC/DC Controller
All Outputs Remain in Regulation Through Cold Crank, 4.5V (Down to
2.5V After Start-Up) ≤ VIN ≤ 38V, VOUT(BUCKS) Up to 24V, VOUT(BOOST)
Up to 60V, IQ = 55μA
LTC3789
High Efficiency Synchronous 4-Switch Buck-Boost
DC/DC Controller
4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 38V, 4mm × 5mm QFN-28 and SSOP-2
3787fc
36 Linear Technology Corporation
LT 0412 REV C • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2010
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