LINER LTC3868IUFD-1 Low iq, dual 2-phase synchronous step-down controller Datasheet

LTC3868-1
Low IQ, Dual
2-Phase Synchronous
Step-Down Controller
Description
Features
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The LTC®3868-1 is a high performance dual step-down
switching regulator controller that drives all N-channel
synchronous power MOSFET stages. A constant frequency
current mode architecture allows a phase-lockable frequency of up to 850kHz. Power loss and noise due to the
input capacitor ESR are minimized by operating the two
controller outputs out of phase.
Low Operating IQ: 170µA (One Channel On)
Wide Output Voltage Range: 0.8V ≤ VOUT ≤ 14V
Wide VIN Range: 4V to 24V
RSENSE or DCR Current Sensing
Out-of-Phase Controllers Reduce Required Input
Capacitance and Power Supply Induced Noise
OPTI-LOOP® Compensation Minimizes COUT
Phase-Lockable Frequency (75kHz to 850kHz)
Programmable Fixed Frequency (50kHz to 900kHz)
Selectable Continuous, Pulse-Skipping or
Burst Mode® Operation at Light Loads
Very Low Dropout Operation: 99% Duty Cycle
Adjustable Output Voltage Soft-Start
Power Good Output Voltage Monitor
Output Overvoltage Protection
Output Latchoff Protection During Short Circuit
Low Shutdown IQ: 8µA
Internal LDO Powers Gate Drive from VIN or EXTVCC
No Current Foldback During Start-Up
Small 4mm × 5mm QFN and Narrow SSOP Packages
The 170μA no-load quiescent current extends operating
life in battery powered systems. OPTI-LOOP compensation allows the transient response to be optimized over
a wide range of output capacitance and ESR values. The
LTC3868-1 features a precision 0.8V reference and a power
good output indicator. A wide 4V to 24V input supply range
encompasses a wide range of intermediate bus voltages
and battery chemistries.
Independent soft-start pins for each controller ramp the
output voltages during start-up. Current foldback limits
MOSFET heat dissipation during short-circuit conditions.
The output short-circuit latchoff feature further protects
the circuit in short-circuit conditions.
Applications
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For a leadless 32-pin QFN package with the additional features of adjustable current limit, clock out, phase modulation and two PGOOD outputs, see the LTC3868 data sheet.
Notebook and Palmtop Computers
Portable Instruments
Battery Operated Digital Devices
Distributed DC Power Systems
L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, µModule, Linear Technology and the Linear logo
are registered trademarks and No RSENSE and UltraFast are trademarks of Linear Technology
Corporation. All other trademarks are the property of their respective owners. Protected by U.S.
Patents, including 5481178, 5705919, 5929620, 6100678, 6144194, 6177787, 6304066, 6580258.
Typical Application
High Efficiency Dual 8.5V/3.3V Step-Down Converter
22µF
50V
4.7µF
0.1µF
3.3µH
VIN
INTVCC
BOOST1
BOOST2
LTC3868-1
0.1µF
80
7.2µH
BG2
PGND
SENSE2+
0.01Ω
0.007Ω
62.5k
150µF
680pF
20k
15k
SENSE2–
VFB2
ITH2
SENSE1–
VFB1
ITH1
SS1
0.1µF
SGND
SS2
0.1µF
193k
680pF
15k
10000
90
SW2
SENSE1+
VOUT1
3.3V
5A
100
20k
VOUT2
8.5V
3.5A
150µF
1000
70
EFFICIENCY
60
50
100
POWER LOSS
40
10
30
20
VIN = 12V
VOUT = 3.3V
FIGURE 12 CIRCUIT
10
0
0.0001
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
10
POWER LOSS (mW)
SW1
BG1
TG2
EFFICIENCY (%)
TG1
Efficiency and Power Loss
vs Load Current
VIN
9V TO 24V
1
0.1
38681 TA01b
38681 TA01
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LTC3868-1
Absolute Maximum Ratings (Note 1)
Input Supply Voltage (VIN).......................... –0.3V to 28V
Topside Driver Voltages
BOOST1, BOOST2 ................................. –0.3V to 34V
Switch Voltage (SW1, SW2) ......................... –5V to 28V
(BOOST1-SW1), (BOOST2-SW2) . ............... –0.3V to 6V
RUN1, RUN2 . .............................................. –0.3V to 8V
Maximum Current Sourced into Pin from
Source >8V .......................................................100µA
SENSE1+, SENSE2+, SENSE1–
SENSE2– Voltages....................................... –0.3V to 16V
PLLIN/MODE, FREQ Voltages ............... –0.3V to INTVCC
EXTVCC . ..................................................... –0.3V to 14V
ITH1, ITH2, VFB1, VFB2 Voltages...................... –0.3V to 6V
PGOOD1 Voltage . ........................................ –0.3V to 6V
SS1, SS2, INTVCC Voltages .......................... –0.3V to 6V
Operating Temperature Range (Note 2).... –40°C to 85°C
Junction Temperature (Note 3).............................. 125°C
Storage Temperature Range.................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec)
SSOP................................................................. 300°C
Pin Configuration
TOP VIEW
28 27 26 25 24 23
3
26 TG1
SENSE1–
4
25 SW1
FREQ
5
24 BOOST1
PLLIN/MODE
6
23 BG1
SGND
7
22 VIN
RUN1
8
21 PGND
RUN2
9
20 EXTVCC
SENSE2–
10
19 INTVCC
SENSE2+
11
18 BG2
TG1
27 PGOOD1
SENSE1+
SS1
VFB1
ITH1
28 SS1
2
VFB1
ITH1
1
SW1
PGOOD1
TOP VIEW
SENSE1+ 1
22 BOOST1
SENSE1– 2
21 BG1
FREQ 3
20 VIN
PLLIN/MODE 4
19 PGND
29
SGND
SGND 5
18 EXTVCC
RUN1 6
17 INTVCC
RUN2 7
16 BG2
SENSE2– 8
15 BOOST2
SW2
TG2
SS2
ITH2
VFB2
SENSE2+
9 10 11 12 13 14
UFD PACKAGE
28-LEAD (4mm s 5mm) PLASTIC QFN
VFB2 12
17 BOOST2
ITH2 13
16 SW2
SS2 14
15 TG2
GN PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 43°C/W
EXPOSED PAD (PIN 29) IS SGND, MUST BE SOLDERED TO PCB
TJMAX = 125°C, θJA = 90°C/W
order information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3868EUFD-1#PBF
LTC3868EUFD-1#TRPBF
38681
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 85°C
LTC3868IUFD-1#PBF
LTC3868IUFD-1#TRPBF
38681
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 85°C
LTC3868EGN-1#PBF
LTC3868EGN-1#TRPBF
LTC3868GN-1
28-Lead Plastic SSOP
–40°C to 85°C
LTC3868IGN-1#PBF
LTC3868IGN-1#TRPBF
LTC3868GN-1
28-Lead Plastic SSOP
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping
container.Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
38681fb
LTC3868-1
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted.
SYMBOL
PARAMETER
VIN
Input Supply Operating Voltage Range
CONDITIONS
VFB1,2
Regulated Feedback Voltage
(Note 4) ITH1,2 Voltage = 1.2V
IFB1,2
Feedback Current
(Note 4)
VREFLNREG
Reference Voltage Line Regulation
(Note 4) VIN = 4.5V to 24V
VLOADREG
Output Voltage Load Regulation
(Note4)
Measured in Servo Loop,
∆ITH Voltage = 1.2V to 0.7V
(Note4)
Measured in Servo Loop,
∆ITH Voltage = 1.2V to 2V
MIN
TYP
4
24
UNITS
V
0.8
0.812
V
±5
±50
nA
0.002
0.02
%/V
l
0.01
0.1
%
l
–0.01
–0.1
%
l
0.788
MAX
gm1,2
Transconductance Amplifier gm
(Note 4) ITH1,2 = 1.2V, Sink/Source = 5µA
IQ
Input DC Supply Current
(Note 5)
Pulse-Skipping or Forced Continuous
Mode
(One Channel On)
RUN1 = 5V and RUN2 = 0V or
RUN1 = 0V and RUN2 = 5V,
VFB1 = 0.83V (No Load)
Pulse-Skipping or Forced Continuous
Mode
(Both Channels On)
RUN1,2 = 5V, VFB1,2 = 0.83V (No Load)
Sleep Mode (One Channel On)
RUN1 = 5V and RUN2 = 0V or
RUN1 = 0V and RUN2 = 5V,
VFB1 = 0.83V (No Load)
170
250
µA
Sleep Mode (Both Channels On)
RUN1,2 = 5V, VFB1,2 = 0.83V (No Load)
300
450
µA
8
25
µA
3.6
4
3.8
4.2
4
V
V
7
10
13
%
±1
µA
±1
700
µA
µA
µA
Shutdown
RUN1,2 = 0V
UVLO
Undervoltage Lockout
INTVCC Ramping Up
INTVCC Ramping Down
VOVL
Feedback Overvoltage Protection
Measured at VFB1,2, Relative to Regulated
VFB1,2
SENSE+ Pin Current
Each Channel
SENSE – Pins Current
Each Channel
VOUT1,2 < INTVCC – 0.5V
VOUT1,2 > INTVCC + 0.5V
ISENSE+
ISENSE
–
2
l
l
mmho
1.3
mA
2
mA
540
DFMAX
Maximum Duty Factor
In Dropout, FREQ = 0V
98
99.4
ISS1,2
Soft-Start Charge Current
VSS1,2 = 0V
0.7
1
1.4
VRUN1,2 On
RUN Pin On Threshold
VRUN1, VRUN2 Rising
1.21
1.26
1.31
l
VRUN1,2 Hyst RUN Pin Hysteresis
VSS1,2 LA
SS Pin Latchoff Arming Threshold
VSS1,2 LT
IDSC1,2 LT
%
50
V
mV
VSS1, VSS2 Rising from 1V
1.9
2
2.1
SS Pin Latchoff Threshold
VSS1, VSS2 Rising from 2V
1.3
1.5
1.7
V
SS Discharge Current
Short-Circuit Condition VFB1,2 = 0.5V
VSS1,2 = 4.5V
7
10
13
µA
VFB1,2 = 0.7V, VSENSE1–,2– = 3.3V
43
50
57
mV
VSENSE(MAX) Maximum Current Sense Threshold
V
Gate Driver
TG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.5
1.5
Ω
Ω
BG1,2
Pull-Up On-Resistance
Pull-Down On-Resistance
2.4
1.1
Ω
Ω
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LTC3868-1
Electrical Characteristics
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 12V, VRUN1,2 = 5V, EXTVCC = 0V unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
TG1,2 tr
TG1,2 t f
TG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
25
16
ns
ns
BG1,2 tr
BG1,2 t f
BG Transition Time:
Rise Time
Fall Time
(Note 6)
CLOAD = 3300pF
CLOAD = 3300pF
28
13
ns
ns
TG/BG t1D
Top Gate Off to Bottom Gate On Delay
Synchronous Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
BG/TG t1D
Bottom Gate Off to Top Gate On Delay
Top Switch-On Delay Time
CLOAD = 3300pF Each Driver
30
ns
tON(MIN)
Minimum On-Time
(Note 7)
95
ns
INTVCC Linear Regulator
VINTVCCVIN
Internal VCC Voltage
6V < VIN < 24V, VEXTVCC = 0V
VLDOVIN
INTVCC Load Regulation
ICC = 0mA to 50mA, VEXTVCC = 0V
VINTVCCEXT
Internal VCC Voltage
6V < VEXTVCC < 13V
4.85
4.85
VLDOEXT
INTVCC Load Regulation
ICC = 0mA to 50mA, VEXTVCC = 8.5V
VEXTVCC
EXTVCC Switchover Voltage
EXTVCC Ramping Positive
VLDOHYS
EXTVCC Hysteresis
4.5
5.1
5.35
V
0.7
1.1
%
5.1
5.35
V
0.6
1.1
%
4.7
4.9
V
250
mV
105
kHz
Oscillator and Phase-Locked Loop
f 25kΩ
Programmable Frequency
RFREQ = 25k, PLLIN/MODE = DC Voltage
f 65kΩ
Programmable Frequency
RFREQ = 65k, PLLIN/MODE = DC Voltage
f105kΩ
Programmable Frequency
RFREQ = 105k, PLLIN/MODE = DC Voltage
fLOW
Low Fixed Frequency
VFREQ = 0V, PLLIN/MODE = DC Voltage
fHIGH
High Fixed Frequency
VFREQ = INTVCC, PLLIN/MODE = DC Voltage
fSYNC
Synchronizable Frequency
PLLIN/MODE = External Clock
375
440
505
835
l
kHz
kHz
320
350
380
kHz
485
535
585
kHz
850
kHz
0.4
V
±1
µA
75
PGOOD1 Output
VPGL
PGOOD1 Voltage Low
IPGOOD = 2mA
IPGOOD
PGOOD1 Leakage Current
VPGOOD = 5V
VPG
PGOOD1 Trip Level
VFB with Respect to Set Regulated Voltage
VFB Ramping Negative
Hysteresis
–13
–10
2.5
–7
%
%
VFB with Respect to Set Regulated Voltage
VFB Ramping Positive
Hysteresis
7
10
2.5
13
%
%
tPG
Delay for Reporting a Fault (PGOOD Low)
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Ratings for extended periods may affect device reliability and
lifetime.
Note 2: The LTC3868E-1 is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC3868I-1 is guaranteed over the
full –40°C to 85°C operating temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJA)
0.2
25
µs
where θJA = 43°C for the QFN package and θJA = 90°C for the SSOP
package.
Note 4: The LTC3868-1 is tested in a feedback loop that servos VITH1,2 to a
specified voltage and measures the resultant VFB1,2.
Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications information.
Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.
Note 7: The minimum on-time condition is specified for an inductor
peak-to-peak ripple current ≥ 40% of IMAX (See Minimum On-Time
Considerations in the Applications Information section).
38681fb
LTC3868-1
Typical Performance Characteristics
Efficiency and Power Loss
vs Output Current
Efficiency vs Load Current
100
90
60
100
50
10
40
30
Burst Mode
OPERATION
PULSESKIPPING
FCM
20
10
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
EFFICIENCY (%)
70
VIN = 5V
80
1000
POWER LOSS (mW)
EFFICIENCY (%)
FIGURE 12 CIRCUIT
90 VIN = 12V
VOUT = 3.3V
80
0
0.0001
100
10000
70
VIN = 12V
60
50
40
30
20
1
10
10
0.1
0
0.0001
VOUT = 3.3V
FIGURE 12 CIRCUIT
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
38681 G02
38681 G01
98
FIGURE 12 CIRCUIT
VOUT = 3.3V
IOUT = 4A
96
EFFICIENCY (%)
94
Load Step
(Forced Continuous Mode)
Load Step (Burst Mode Operation)
Efficiency vs Input Voltage
VOUT
100mV/DIV
ACCOUPLED
92
10
VOUT
100mV/DIV
ACCOUPLED
90
88
IL
2A/DIV
86
IL
2A/DIV
84
82
80
0
5
20
10
15
INPUT VOLTAGE (V)
25
28
VOUT = 3.3V
20µs/DIV
FIGURE 12 CIRCUIT
38681 G04
38681 G05
20µs/DIV
VOUT = 3.3V
FIGURE 12 CIRCUIT
38681 G03
Inductor Current at Light Load
Load Step (Pulse-Skipping Mode)
VOUT
100mV/DIV
ACCOUPLED
IL
2A/DIV
Soft Start-Up
FORCED
CONTINUOUS
MODE
VOUT2
2V/DIV
Burst Mode
OPERATION
2A/DIV
VOUT1
2V/DIV
PULSESKIPPING
MODE
VOUT = 3.3V
20µs/DIV
FIGURE 12 CIRCUIT
38681 G06
VOUT = 3.3V
2µs/DIV
ILOAD = 200µA
FIGURE 12 CIRCUIT
38681 G07
20ms/DIV
FIGURE 12 CIRCUIT
38681 G08
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LTC3868-1
Typical Performance Characteristics
Total Input Supply Current
vs Input Voltage
300µA LOAD
250
200
NO LOAD
150
100
50
0
10
5
15
20
INPUT VOLTAGE (V)
25
5.4
5.2
INTVCC
5.0
EXTVCC RISING
4.8
EXTVCC FALLING
4.6
4.4
4.0
–45 –20
28
80
55
30
TEMPERATURE (°C)
5
38681 G10
Maximum Current Sense Voltage
vs ITH Voltage
–50
–20
–150
–200
–250
–300
–350
–400
–450
–500
5% DUTY CYCLE
0.2
0.4 0.6 0.8 1.0
ITH PIN VOLTAGE
1.2
1.4
–550
–600
0
10
5
VSENSE COMMON MODE VOLTAGE (V)
80
50
40
30
20
10
0
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
FEEDBACK VOLTAGE (V)
38681 G16
10
15
20
INPUT VOLTAGE (V)
240
230 PLLIN/MODE = 0
V = 12V
220 VIN = 3.3V
OUT
210 ONE CHANNEL ON
200
190
180
170
160
150
140
130
120
110
80
55
5
–45 –20
30
TEMPERATURE (°C)
25 28
38681 G12
80
60
40
20
0
15
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
38681 G15
Shutdown Current vs Temperature
Quiescent Current vs Temperature
QUIESCENT CURRENT (µA)
MAXIMUM CURRENT SENSE VOLTAGE (mV)
Foldback Current Limit
90
60
5
38681 G14
38681 G13
70
0
Maximum Current Sense
Threshold vs Duty Cycle
–100
0
0
5.0
130
0
20
–40
105
MAXIMUM CURRENT SENSE VOLTAGE (mV)
40
5.1
SENSE– Pins Input Bias Current
PULSE-SKIPPING
FORCED CONTINUOUS
Burst Mode OPERATION
(FALLING)
Burst Mode OPERATION
(RISING)
60
5.1
38681 G11
SENSE– CURRENT (µA)
CURRENT SENSE THRESHOLD (mV)
80
5.2
4.2
10
SHUTDOWN CURRENT (µA)
SUPPLY CURRENT (µA)
300
5.2
INTVCC VOLTAGE (V)
FIGURE 12 CIRCUIT
VOUT = 3.3V
ONE CHANNEL ON
350
INTVCC Line Regulation
5.6
EXTVCC AND INTVCC VOLTAGE (V)
400
EXTVCC Switchover and INTVCC
Voltages vs Temperature
9
8
7
6
5
105
130
38681 G17
4
–45 –20
55
30
80
5
TEMPERATURE (°C)
105
130
38681 G18
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LTC3868-1
Typical Performance Characteristics
Soft-Start Pull-Up Current
vs Temperature
1.40
1.20
808
REGULATED FEEDBACK VOLTAGE (mV)
1.35
1.15
1.30
1.10
RUN PIN VOLTAGE (V)
SS PULL-UP CURRENT (µA)
Regulated Feedback Voltage
vs Temperature
Shutdown (RUN) Threshold
vs Temperature
1.05
1.00
0.95
0.90
1.25
1.20
1.15
1.10
1.05
1.00
0.85
0.95
0.80
–45 –20
80
55
30
TEMPERATURE (°C)
5
105
0.90
–45
130
55
30
5
80
TEMPERATURE (°C)
–20
105
800
798
796
794
792
–45 –20
130
130
14
800
12
700
600
10
8
6
4
0
105
130
38681 G21
FREQUENCY (kHz)
INPUT CURRENT (µA)
105
80
55
30
TEMPERATURE (°C)
5
Oscillator Frequency
vs Temperature
FREQ = INTVCC
500
FREQ = GND
400
300
200
2
VOUT = 28V
80
55
5
30
TEMPERATURE (°C)
802
Shutdown Current
vs Input Voltage
VOUT = 3.3V
100
5
10
25
20
15
INPUT VOLTAGE (V)
38681 G22
28
0
–45 –20
80
55
30
TEMPERATURE (°C)
5
38681 G23
105
130
38681 G24
Undervoltage Lockout Threshold
vs Temperature
Oscillator Frequency
vs Input Voltage
4.4
356
4.3
354
4.2
INTVCC VOLTAGE (V)
OSCILLATOR FREQUENCY (kHz)
SENSE– CURRENT (µA)
SENSE– Pin Input Current
vs Temperature
–20
804
38681 G20
38681 G19
50
0
–50
–100
–150
–200
–250
–300
–350
–400
–450
–500
–550
–600
–45
806
352
350
348
4.0
3.9
3.8
3.7
3.6
346
344
4.1
3.5
5
10
20
15
INPUT VOLTAGE (V)
25
28
38681 G28
3.4
–45
–20
55
30
5
80
TEMPERATURE (°C)
105
130
38681 G25
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LTC3868-1
Typical Performance Characteristics
Latchoff Thresholds
vs Temperature
INTVCC vs Load Current
5.20
2.3
VIN = 12V
2.2
2.1
INTVCC VOLTAGE (V)
INTVCC VOLTAGE (V)
5.15
5.10
EXTVCC = 0V
5.05
EXTVCC = 8V
5.00
ARMING THRESHOLD
2.0
1.9
1.8
1.7
1.6
LATCH-OFF THRESHOLD
1.5
1.4
1.3
4.95
0
20 40 60 80 100 120 140 160 180 200
LOAD CURRENT (mA)
38681 G26
Pin Functions
1.2
–45
–20
55
30
5
80
TEMPERATURE (°C)
105
130
38681 G27
(QFN/SSOP)
SENSE1–, SENSE2– (Pin 2, Pin 4/Pin 8, Pin 10): The (–)
Input to the Differential Current Comparators. When greater
than INTVCC – 0.5V, the SENSE– pin supplies current to
the current comparator.
FREQ (Pin 3/Pin 5): The Frequency Control Pin for the
Internal VCO. Connecting this pin to GND forces the VCO
to a fixed low frequency of 350kHz. Connecting this pin
to INTVCC forces the VCO to a fixed high frequency of
535kHz. Other frequencies between 50kHz and 900kHz can
be programmed using a resistor between FREQ and GND.
An internal 20µA pull-up current develops the voltage to
be used by the VCO to control the frequency
PLLIN/MODE (Pin 4/Pin 6): External Synchronization
Input to Phase Detector and Forced Continuous Mode
Input. When an external clock is applied to this pin, the
phase-locked loop will force the rising TG1 signal to be
synchronized with the rising edge of the external clock.
When not synchronizing to an external clock, this input,
which acts on both controllers, determines how the
LTC3868-1 operates at light loads. Pulling this pin to
ground selects Burst Mode operation. An internal 100k
resistor to ground also invokes Burst Mode operation
when the pin is floated. Tying this pin to INTVCC forces
continuous inductor current operation. Tying this pin to
a voltage greater than 1.2V and less than INTVCC – 1.3V
selects pulse-skipping operation.
SGND (Pin 5, Exposed Pad Pin 29/Pin 7): Small-signal
ground common to both controllers, must be routed
separately from high current grounds to the common (–)
terminals of the CIN capacitors. The exposed pad (QFN
only) must be soldered to the PCB for rated thermal
performance.
RUN1, RUN2 (Pin 6, Pin 8/Pin 7, Pin 9): Digital Run
Control Inputs for Each Controller. Forcing either of these
pins below 1.26V shuts down that controller. Forcing both
of these pins below 0.7V shuts down the entire LTC3868‑1,
reducing quiescent current to approximately 8µA. Do not
float these pins.
38681fb
LTC3868-1
Pin Functions
(QFN/SSOP)
INTVCC (Pin 17/Pin 19): Output of the Internal Linear Low
Dropout Regulator. The driver and control circuits are
powered from this voltage source. Must be decoupled to
power ground with a minimum of 4.7µF ceramic or other
low ESR capacitor. Do not use the INTVCC pin for any
other purpose.
EXTVCC (Pin 18/Pin 20): External Power Input to an
Internal LDO Connected to INTVCC. This LDO supplies
INTVCC power, bypassing the internal LDO powered from
VIN whenever EXTVCC is higher than 4.7V. See EXTVCC
Connection in the Applications Information section. Do
not exceed 14V on this pin.
PGND (Pin 19/Pin 21): Driver Power Ground. Connects to
the sources of bottom (synchronous) N-channel MOSFETs
and the (–) terminal(s) of CIN.
VIN (Pin 20/Pin 22): Main Supply Pin. A bypass capacitor should be tied between this pin and the signal ground
pin.
TG1, TG2 (Pin 24, Pin 26/Pin 13, Pin 15): High Current
Gate Drives for Top N-Channel MOSFETs. These are the
outputs of floating drivers with a voltage swing equal to
INTVCC – 0.5V superimposed on the switch node voltage
SW.
PGOOD1 (Pin 25/Pin 27): Open-Drain Logic Output.
PGOOD1 is pulled to ground when the voltage on the VFB1
pin is not within ±10% of its set point.
SS1, SS2 (Pin 26, Pin 28/Pin 12, Pin 14): External SoftStart Input. The LTC3868-1 regulates the VFB1,2 voltage
to the smaller of 0.8V or the voltage on the SS1,2 pin. An
internal 1µA pull-up current source is connected to this
pin. A capacitor to ground at this pin sets the ramp time
to final regulated output voltage. This pin is also used as
the short-circuit latchoff timer.
ITH1, ITH2 (Pin 27, Pin 1/Pin 11, Pin 13): Error Amplifier
Outputs and Switching Regulator Compensation Points.
Each associated channel’s current comparator trip point
increases with this control voltage.
BG1, BG2 (Pin 21, Pin 23/Pin 16, Pin 18): High Current
Gate Drives for Bottom (Synchronous) N-Channel
MOSFETs. Voltage swing at these pins is from ground
to INTVCC.
VFB1, VFB2 (Pin 28, Pin 2/Pin 10, Pin 12): Receives the
remotely sensed feedback voltage for each controller from
an external resistive divider across the output.
BOOST1, BOOST2 (Pin 22, Pin 24/Pin 15, Pin 17):
Bootstrapped Supplies to the Topside Floating Drivers.
Capacitors are connected between the BOOST and SW pins
and Schottky diodes are tied between the BOOST and INTVCC
pins. Voltage swing at the BOOST pins is from INTVCC to
(VIN + INTVCC).
SENSE1+, SENSE2+ (Pin 1, Pin 3/Pin 9, Pin 11): The (+)
input to the differential current comparators are normally
connected to DCR sensing networks or current sensing
resistors. The ITH pin voltage and controlled offsets between
the SENSE– and SENSE+ pins in conjunction with RSENSE
set the current trip threshold.
SW1, SW2 (Pin 23, Pin 25/Pin 14, Pin 16): Switch Node
Connections to Inductors.
38681fb
LTC3868-1
FUNCTIONAL Diagram
INTVCC
DUPLICATE FOR SECOND
CONTROLLER CHANNEL
PGOOD1
BOOST
DROP
OUT
DET
0.88V
VFB1
+
–
+
0.72V
S
Q
R
Q
TOP ON
SWITCH
LOGIC BOT
INTVCC
BG
VOUT
CLK2
0.425V
–
CLK1
+
SLEEP
–
ICMP
PFD
+
+
–+
+–
–
SYNC
DET
PLLIN/MODE
IR
–
SENSE+
2(VFB)
0.45V
100k
SENSE–
SLOPE COMP
VFB
VIN
+
EA
–
OV
–
5.1V
LDO
EN
LDO
EN
0.5µA
+
–
SHDN
RST
2(VFB)
0.80V
TRACK/SS
0.88V
SGND
INTVCC
RUN
RA
CC
ITH
CC2
FOLDBACK
1µA
11V
RB
+
EXTVCC
5.1V
RSENSE
L
3mV
4.7V
COUT
PGND
VCO
CIN
SW
20µA
FREQ
CB
D
BOT
SHDN
DB
TG
TOP
VIN
SHORT CKT
LATCH-OFF
RC
SS
CSS
SHDN
10µA
38681 FD
38681fb
10
LTC3868-1
Operation (Refer to the Functional Diagram)
Main Control Loop
The LTC3868-1 uses a constant frequency, current mode
step-down architecture with the two controller channels
operating 180 degrees out of phase. During normal operation, each external top MOSFET is turned on when the
clock for that channel sets the RS latch, and is turned off
when the main current comparator, ICMP, resets the RS
latch. The peak inductor current at which ICMP trips and
resets the latch is controlled by the voltage on the ITH pin,
which is the output of the error amplifier, EA. The error
amplifier compares the output voltage feedback signal at
the VFB pin (which is generated with an external resistor
divider connected across the output voltage, VOUT , to
ground) to the internal 0.800V reference voltage. When the
load current increases, it causes a slight decrease in VFB
relative to the reference, which causes the EA to increase
the ITH voltage until the average inductor current matches
the new load current.
After the top MOSFET is turned off each cycle, the bottom
MOSFET is turned on until either the inductor current starts
to reverse, as indicated by the current comparator IR, or
the beginning of the next clock cycle.
INTVCC/EXTVCC Power
Power for the top and bottom MOSFET drivers and most
other internal circuitry is derived from the INTVCC pin. When
the EXTVCC pin is left open or tied to a voltage less than
4.7V, the VIN LDO (low dropout linear regulator) supplies
5.1V from VIN to INTVCC. If EXTVCC is taken above 4.7V,
the VIN LDO is turned off and an EXTVCC LDO is turned on.
Once enabled, the EXTVCC LDO supplies 5.1V from EXTVCC
to INTVCC. Using the EXTVCC pin allows the INTVCC power
to be derived from a high efficiency external source such
as one of the LTC3868-1 switching regulator outputs.
Each top MOSFET driver is biased from the floating bootstrap capacitor, CB, which normally recharges during each
cycle through an external diode when the top MOSFET
turns off. If the input voltage VIN decreases to a voltage
close to VOUT , the loop may enter dropout and attempt
to turn on the top MOSFET continuously. The dropout
detector detects this and forces the top MOSFET off for
about one-twelfth of the clock period every tenth cycle to
allow CB to recharge.
Shutdown and Start-Up (RUN1, RUN2 and
SS1, SS2 Pins)
The two channels of the LTC3868-1 can be independently
shut down using the RUN1 and RUN2 pins. Pulling either of
these pins below 1.26V shuts down the main control loop
for that controller. Pulling both pins below 0.7V disables
both controllers and most internal circuits, including the
INTVCC LDOs. In this state, the LTC3868-1 draws only 8µA
of quiescent current.
The RUN pin may be externally pulled up or driven directly
by logic. When driving the RUN pin with a low impedance
source, do not exceed the absolute maximum rating of
8V. The RUN pin has an internal 11V voltage clamp that
allows the RUN pin to be connected through a resistor to a
higher voltage (for example, VIN), so long as the maximum
current into the RUN pin does not exceed 100µA.
The start-up of each controller’s output voltage VOUT is
controlled by the voltage on the SS pin for that channel.
When the voltage on the SS pin is less than the 0.8V
internal reference, the LTC3868-1 regulates the VFB voltage to the SS pin voltage instead of the 0.8V reference.
This allows the SS pin to be used to program a soft-start
by connecting an external capacitor from the SS pin to
SGND. An internal 1µA pull-up current charges this capacitor creating a voltage ramp on the SS pin. As the SS
voltage rises linearly from 0V to 0.8V (and beyond up to
the absolute maximum rating of 6V), the output voltage
VOUT rises smoothly from zero to its final value.
Short-Circuit Latchoff
After the controller has been started and been given adequate time to ramp up the output voltage, the SS capacitor is used in a short-circuit timeout circuit. Specifically,
once the voltage on the SS pin rises above 2V (the arming
threshold), the short-circuit timeout circuit is enabled (see
Figure 1). If the output voltage falls below 70% of its nominal regulated voltage, the SS capacitor begins discharging with a net 9µA pulldown current on the assumption
that the output is in an overcurrent and/or short-circuit
condition. If the condition lasts long enough to allow the
SS pin voltage to fall below 1.5V (the latchoff threshold),
the controller will shut down (latch off) until the RUN pin
voltage or the VIN voltage is recycled.
38681fb
11
LTC3868-1
Operation (Refer to the Functional Diagram)
current is provided due to internal current foldback and
actual power wasted is low due to the efficient nature of
the current mode switching regulator. Foldback current
limiting is disabled during the soft-start interval (as long
as the VFB voltage is keeping up with the SS voltage).
INTVCC
SS VOLTAGE
2V
1.5V
0.8V
LATCHOFF
COMMAND
SS PIN
CURRENT
Light Load Current Operation (Burst Mode Operation,
Pulse-Skipping or Forced Continuous Mode)
(PLLIN/MODE Pin)
0V
1µA
1µA
–9µA
OUTPUT
VOLTAGE
LATCHOFF
ENABLE
ARMING
tLATCH
38681 F01
SOFT-START INTERVAL
Figure 1. Latchoff Timing Diagram
The delay time from when a short-circuit occurs until the
controller latches off can be calculated using the following equation
tLATCH ~ CSS (VSS – 1.5V)/9µA
where VSS is the initial voltage (must be greater than 2V)
on the SS pin at the time the short-circuit occurs. Normally
the SS pin voltage will have been pulled up to the INTVCC
voltage (5.1V) by the internal 1µA pull-up current.
Note that the two controllers on the LTC3868-1 have separate, independent short-circuit latchoff circuits. Latchoff
can be overridden/defeated by connecting a resistor 150k
or less from the SS pin to INTVCC. This resistor provides
enough pull-up current to overcome the 9µA pull-down
current present during a short-circuit. Note that this resistor also shortens the soft-start period.
Foldback Current
On the other hand, when the output voltage falls to less
than 70% of its nominal level, foldback current limiting
is also activated, progressively lowering the peak current
limit in proportion to the severity of the overcurrent or
short-circuit condition. Even if a short-circuit is present
and the short-circuit latchoff is not yet enabled (when
SS voltage has not yet reached 2V), a safe, low output
The LTC3868-1 can be enabled to enter high efficiency
Burst Mode operation, constant frequency pulse-skipping
mode, or forced continuous conduction mode at low load
currents. To select Burst Mode operation, tie the PLLIN/
MODE pin to ground. To select forced continuous operation, tie the PLLIN/MODE pin to INTVCC. To select pulseskipping mode, tie the PLLIN/MODE pin to a DC voltage
greater than 1.2V and less than INTVCC – 1.3V.
When a controller is enabled for Burst Mode operation,
the minimum peak current in the inductor is set to approximately 30% of the maximum sense voltage even
though the voltage on the ITH pin indicates a lower value.
If the average inductor current is higher than the load
current, the error amplifier, EA, will decrease the voltage
on the ITH pin. When the ITH voltage drops below 0.425V,
the internal sleep signal goes high (enabling sleep mode)
and both external MOSFETs are turned off.
In sleep mode, much of the internal circuitry is turned off,
reducing the quiescent current. If one channel is shut down
and the other channel is in sleep mode, the LTC3868-1
draws only 170µA of quiescent current. If both channels
are in sleep mode, the LTC3868-1 draws only 300µA of quiescent current. In sleep mode, the load current is supplied
by the output capacitor. As the output voltage decreases,
the EA’s output begins to rise. When the output voltage
drops enough, the ITH pin is reconnected to the output
of the EA, the sleep signal goes low, and the controller
resumes normal operation by turning on the top external
MOSFET on the next cycle of the internal oscillator.
When a controller is enabled for Burst Mode operation,
the inductor current is not allowed to reverse. The reverse
38681fb
12
LTC3868-1
Operation (Refer to the Functional Diagram)
current comparator, IR, turns off the bottom external
MOSFET just before the inductor current reaches zero,
preventing it from reversing and going negative. Thus,
the controller is in discontinuous operation.
In forced continuous operation or when clocked by an
external clock source to use the phase-locked loop (see
Frequency Selection and Phase-Locked Loop section),
the inductor current is allowed to reverse at light loads
or under large transient conditions. The peak inductor
current is determined by the voltage on the ITH pin, just
as in normal operation. In this mode, the efficiency at light
loads is lower than in Burst Mode operation. However,
continuous operation has the advantages of lower output
voltage ripple and less interference to audio circuitry. In
forced continuous mode, the output ripple is independent
of load current.
When the PLLIN/MODE pin is connected for pulse-skipping
mode, the LTC3868-1 operates in PWM pulse-skipping
mode at light loads. In this mode, constant frequency
operation is maintained down to approximately 1% of
designed maximum output current. At very light loads, the
current comparator, ICMP, may remain tripped for several
cycles and force the external top MOSFET to stay off for
the same number of cycles (i.e., skipping pulses). The
inductor current is not allowed to reverse (discontinuous
operation). This mode, like forced continuous operation,
exhibits low output ripple as well as low audio noise and
reduced RF interference when compared to Burst Mode
operation. It provides higher light load efficiency than
forced continuous mode, but not nearly as high as Burst
Mode operation.
Frequency Selection and Phase-Locked Loop
(FREQ and PLLIN/MODE Pins)
The selection of switching frequency is a trade off between
efficiency and component size. Low frequency operation increases efficiency by reducing MOSFET switching
losses, but requires larger inductance and/or capacitance
to maintain low output ripple voltage.
The switching frequency of the LTC3868-1’s controllers
can be selected using the FREQ pin.
If the PLLIN/MODE pin is not being driven by an external
clock source, the FREQ pin can be tied to SGND, tied to
INTVCC or programmed through an external resistor. Tying
FREQ to SGND selects 350kHz while tying FREQ to INTVCC
selects 535kHz. Placing a resistor between FREQ and
SGND allows the frequency to be programmed between
50kHz and 900kHz.
A phase-locked loop (PLL) is available on the LTC3868-1
to synchronize the internal oscillator to an external clock
source that is connected to the PLLIN/MODE pin. The
phase detector adjusts the voltage (through an internal
lowpass filter) of the VCO input to align the turn-on of
controller 1’s external top MOSFET to the rising edge of
the synchronizing signal. Thus, the turn-on of controller
2’s external top MOSFET is 180 degrees out of phase to
the rising edge of the external clock source.
The VCO input voltage is prebiased to the operating frequency set by the FREQ pin before the external clock is
applied. If prebiased near the external clock frequency,
the PLL loop only needs to make slight changes to the
VCO input in order to synchronize the rising edge of the
external clock’s to the rising edge of TG1. The ability to
prebias the loop filter allows the PLL to lock-in rapidly
without deviating far from the desired frequency.
The typical capture range of the phase-locked loop is from
approximately 55kHz to 1MHz, with a guarantee over all
manufacturing variations to be between 75kHz and 850kHz.
In other words, the LTC3868-1’s PLL is guaranteed to lock
to an external clock source whose frequency is between
75kHz and 850kHz.
The typical input clock thresholds on the PLLIN/MODE
pin are 1.6V (rising) and 1.1V (falling).
Output Overvoltage Protection
An overvoltage comparator guards against transient overshoots as well as other more serious conditions that may
overvoltage the output. When the VFB pin rises by more
than 10% above its regulation point of 0.800V, the top
MOSFET is turned off and the bottom MOSFET is turned
on until the overvoltage condition is cleared.
38681fb
13
LTC3868-1
Operation (Refer to the Functional Diagram)
Power Good (PGOOD) Pin
The PGOOD1 pin is connected to an open drain of an
internal N-channel MOSFET. The MOSFET turns on and
pulls the PGOOD1 pin low when the corresponding VFB1 pin
voltage is not within ±10% of the 0.8V reference voltage.
The PGOOD1 pin is also pulled low when the RUN1 pin
is low (shut down). When the VFB1 pin voltage is within
the ±10% requirement, the MOSFET is turned off and the
pin is allowed to be pulled up by an external resistor to a
source no greater than 6V.
pulses increased the total RMS current flowing from the
input capacitor, requiring the use of more expensive input
capacitors and increasing both EMI and losses in the input
capacitor and battery.
Theory and Benefits of 2-Phase Operation
With 2-phase operation, the two channels of the dual
switching regulator are operated 180 degrees out of phase.
This effectively interleaves the current pulses drawn by the
switches, greatly reducing the overlap time where they add
together. The result is a significant reduction in total RMS
input current, which in turn allows less expensive input
capacitors to be used, reduces shielding requirements for
EMI and improves real world operating efficiency.
Why the need for 2-phase operation? Up until the
2-phase family, constant frequency dual switching regulators operated both channels in phase (i.e., single phase
operation). This means that both switches turned on at
the same time, causing current pulses of up to twice the
amplitude of those for one regulator to be drawn from the
input capacitor and battery. These large amplitude current
Figure 2 compares the input waveforms for a representative
single phase dual switching regulator to the LTC3868-1
2-phase dual switching regulator. An actual measurement of the RMS input current under these conditions
shows that 2-phase operation dropped the input current
from 2.53ARMS to 1.55ARMS. While this is an impressive
reduction in itself, remember that the power losses are
5V SWITCH
20V/DIV
3.3V SWITCH
20V/DIV
INPUT CURRENT
5A/DIV
INPUT VOLTAGE
500mV/DIV
IIN(MEAS) = 2.53ARMS
IIN(MEAS) = 1.55ARMS
38681 F01
Figure 2. Input Waveforms Comparing Single-Phase (a) and 2-Phase (b) Operation for Dual Switching Regulators
Converting 12V to 5V and 3.3V at 3A Each. The Reduced Input Ripple with the 2-Phase Regulator Allows
Less Expensive Input Capacitors, Reduces Shielding Requirements for EMI and Improves Efficiency
38681fb
14
LTC3868-1
Operation (Refer to the Functional Diagram)
proportional to IRMS2, meaning that the actual power wasted
is reduced by a factor of 2.66. The reduced input ripple
voltage also means less power is lost in the input power
path, which could include batteries, switches, trace/connector resistances and protection circuitry. Improvements
in both conducted and radiated EMI also directly accrue as
a result of the reduced RMS input current and voltage.
Of course, the improvement afforded by 2-phase operation is a function of the dual switching regulator’s relative
duty cycles which, in turn, are dependent upon the input
voltage VIN (Duty Cycle = VOUT/VIN). Figure 3 shows how
the RMS input current varies for single-phase and 2-phase
operation for 3.3V and 5V regulators over a wide input
voltage range.
It can readily be seen that the advantages of 2-phase operation are not just limited to a narrow operating range,
for most applications is that 2-phase operation will reduce
the input capacitor requirement to that for just one channel
operating at maximum current and 50% duty cycle.
3.0
SINGLE PHASE
DUAL CONTROLLER
INPUT RMS CURRENT (A)
2.5
2.0
1.5
2-PHASE
DUAL CONTROLLER
1.0
0.5
0
VO1 = 5V/3A
VO2 = 3.3V/3A
0
10
20
30
INPUT VOLTAGE (V)
40
38681 F03
Figure 3. RMS Input Current Comparison
38681fb
15
LTC3868-1
Applications Information
The Typical Application on the first page is a basic
LTC3868‑1 application circuit. LTC3868-1 can be configured
to use either DCR (inductor resistance) sensing or low
value resistor sensing. The choice between the two current sensing schemes is largely a design trade off between
cost, power consumption and accuracy. DCR sensing is
becoming popular because it saves expensive current
sensing resistors and is more power efficient, especially
in high current applications. However, current sensing
resistors provide the most accurate current limits for the
controller. Other external component selection is driven
by the load requirement, and begins with the selection of
RSENSE (if RSENSE is used) and inductor value. Next, the
power MOSFETs and Schottky diodes are selected. Finally,
input and output capacitors are selected.
SENSE+ and SENSE– Pins
The SENSE+ and SENSE– pins are the inputs to the current comparators. The common mode voltage range on
these pins is 0V to 16V (Absolute Maximum), enabling
the LTC3868-1 to regulate output voltages up to a nominal
14V (allowing margin for tolerances and transients).
programmed current limit unpredictable. If inductor DCR
sensing is used (Figure 5b), resistor R1 should be placed
close to the switching node, to prevent noise from coupling
into sensitive small-signal nodes.
TO SENSE FILTER,
NEXT TO THE CONTROLLER
COUT
38681 F04
INDUCTOR OR RSENSE
Figure 4. Sense Lines Placement with Inductor or Sense Resistor
VIN
INTVCC
BOOST
TG
SW
LTC3868-1
Filter components mutual to the sense lines should be
placed close to the LTC3868-1, and the sense lines should
run close together to a Kelvin connection underneath the
current sense element (shown in Figure 4). Sensing current elsewhere can effectively add parasitic inductance
and capacitance to the current sense element, degrading
the information at the sense terminals and making the
VOUT
BG
SENSE+
PLACE CAPACITOR NEAR
SENSE PINS
SENSE–
The SENSE+ pin is high impedance over the full common
mode range, drawing at most ±1µA. This high impedance
allows the current comparators to be used in inductor
DCR sensing.
The impedance of the SENSE– pin changes depending on
the common mode voltage. When SENSE– is less than
INTVCC – 0.5V, a small current of less than 1µA flows out
of the pin. When SENSE– is above INTVCC + 0.5V, a higher
current (~550µA) flows into the pin. Between INTVCC – 0.5V
and INTVCC + 0.5V, the current transitions from the smaller
current to the higher current.
VIN
SGND
38681 F05a
(5a) Using a Resistor to Sense Current
VIN
INTVCC
VIN
BOOST
INDUCTOR
TG
L
SW
LTC3868-1
DCR
VOUT
BG
R1
SENSE+
C1*
R2
SENSE–
SGND
*PLACE C1 NEAR
SENSE PINS
(R1||R2) • C1 =
L
DCR
RSENSE(EQ) = DCR
R2
R1 + R2
38681 F05b
(5b) Using the Inductor DCR to Sense Current
Figure 5. Current Sensing Methods
38681fb
16
LTC3868-1
Applications Information
Low Value Resistor Current Sensing
A typical sensing circuit using a discrete resistor is shown
in Figure 5a. RSENSE is chosen based on the required
output current.
The current comparator has a maximum threshold
VSENSE(MAX) of 50mV. The current comparator threshold
voltage sets the peak of the inductor current, yielding a
maximum average output current, IMAX, equal to the peak
value less half the peak-to-peak ripple current, ∆IL. To
calculate the sense resistor value, use the equation:
RSENSE =
VSENSE(MAX)
IMAX +
∆IL
2
When using the controller in very low dropout conditions,
the maximum output current level will be reduced due to
the internal compensation required to meet stability criterion for buck regulators operating at greater than 50%
duty factor. A curve is provided in the Typical Performance
Characteristics section to estimate this reduction in
peak output current depending upon the operating duty
factor.
Inductor DCR Sensing
For applications requiring the highest possible efficiency
at high load currents, the LTC3850 is capable of sensing
the voltage drop across the inductor DCR, as shown in
Figure 5b. The DCR of the inductor represents the small
amount of DC resistance of the copper wire, which can be
less than 1mΩ for today’s low value, high current inductors.
In a high current application requiring such an inductor,
power loss through a sense resistor would cost several
points of efficiency compared to inductor DCR sensing.
If the external R1||R2 • C1 time constant is chosen to be
exactly equal to the L/DCR time constant, the voltage drop
across the external capacitor is equal to the drop across
the inductor DCR multiplied by R2/(R1 + R2). R2 scales the
voltage across the sense terminals for applications where
the DCR is greater than the target sense resistor value.
To properly dimension the external filter components, the
DCR of the inductor must be known. It can be measured
using a good RLC meter, but the DCR tolerance is not
always the same and varies with temperature; consult the
manufacturers’ data sheets for detailed information.
Using the inductor ripple current value from the Inductor
Value Calculation section, the target sense resistor value
is:
RSENSE(EQUIV) =
VSENSE(MAX)
IMAX +
∆IL
2
To ensure that the application will deliver full load current
over the full operating temperature range, choose the
minimum value for the maximum current sense threshold
voltage (VSENSE(MAX)) in the Electrical Characteristics
table.
Next, determine the DCR of the inductor. When provided,
use the manufacturer’s maximum value, usually given at
20°C. Increase this value to account for the temperature
coefficient of copper resistance, which is approximately
0.4%/°C. A conservative value for TL(MAX) is 100°C.
To scale the maximum inductor DCR to the desired sense
resistor (RD) value, use the divider ratio:
RD =
RSENSE(EQUIV )
DCRMAX at TL(MAX )
C1 is usually selected to be in the range of 0.1µF to 0.47µF.
This forces R1||R2 to around 2k, reducing error that might
have been caused by the SENSE+ pin’s ±1µA current.
The equivalent resistance R1||R2 is scaled to the room
temperature inductance and maximum DCR:
R1|| R2 =
L
DCR at 20°C • C1
(
)
The sense resistor values are:
R1 =
R1 • RD
R1|| R2
; R2 =
RD
1 – RD
38681fb
17
LTC3868-1
Applications Information
The maximum power loss in R1 is related to duty cycle,
and will occur in continuous mode at the maximum input
voltage:
PLOSS R1 =
( VIN(MAX) – VOUT ) • VOUT
R1
Ensure that R1 has a power rating higher than this value.
If high efficiency is necessary at light loads, consider this
power loss when deciding whether to use DCR sensing or
sense resistors. Light load power loss can be modestly
higher with a DCR network than with a sense resistor, due
to the extra switching losses incurred through R1. However,
DCR sensing eliminates a sense resistor, reduces conduction losses and provides higher efficiency at heavy loads.
Peak efficiency is about the same with either method.
Inductor Value Calculation
The operating frequency and inductor selection are interrelated in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
of MOSFET gate charge losses. In addition to this basic
trade-off, the effect of inductor value on ripple current and
low current operation must also be considered.
The inductor value has a direct effect on ripple current.
The inductor ripple current ∆IL decreases with higher
inductance or higher frequency and increases with higher
VIN:
ΔIL =
 V 
1
VOUT 1– OUT 
( f) (L)  VIN 
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.3(IMAX). The maximum
∆IL occurs at the maximum input voltage.
The inductor value also has secondary effects. The transition to Burst Mode operation begins when the average
inductor current required results in a peak current below
30% of the current limit determined by RSENSE. Lower
inductor values (higher ∆IL) will cause this to occur at
lower load currents, which can cause a dip in efficiency in
the upper range of low current operation. In Burst Mode
operation, lower inductance values will cause the burst
frequency to decrease.
Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite or molypermalloy
cores. Actual core loss is independent of core size for a
fixed inductor value, but it is very dependent on inductance
value selected. As inductance increases, core losses go
down. Unfortunately, increased inductance requires more
turns of wire and therefore copper losses will increase.
Ferrite designs have very low core loss and are preferred
for high switching frequencies, so design goals can
concentrate on copper loss and preventing saturation.
Ferrite core material saturates hard, which means that
inductance collapses abruptly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple. Do
not allow the core to saturate!
Power MOSFET and Schottky Diode
(Optional) Selection
Two external power MOSFETs must be selected for each
controller in the LTC3868-1: one N-channel MOSFET for
the top (main) switch, and one N-channel MOSFET for the
bottom (synchronous) switch.
The peak-to-peak drive levels are set by the INTVCC voltage.
This voltage is typically 5.1V during start-up (see EXTVCC
Pin Connection). Consequently, logic-level threshold
MOSFETs must be used in most applications. The only
exception is if low input voltage is expected (VIN < 4V);
then, sub-logic level threshold MOSFETs (VGS(TH) < 3V)
should be used. Pay close attention to the BVDSS specification for the MOSFETs as well; many of the logic-level
MOSFETs are limited to 30V or less.
38681fb
18
LTC3868-1
Applications Information
Selection criteria for the power MOSFETs include the
on-resistance, RDS(ON), Miller capacitance, CMILLER, input
voltage and maximum output current. Miller capacitance,
CMILLER, can be approximated from the gate charge curve
usually provided on the MOSFET manufacturers’ data
sheet. CMILLER is equal to the increase in gate charge
along the horizontal axis while the curve is approximately
flat divided by the specified change in VDS. This result is
then multiplied by the ratio of the application applied VDS
to the gate charge curve specified VDS. When the IC is
operating in continuous mode the duty cycles for the top
and bottom MOSFETs are given by:
Main Switch Duty Cycle =
VOUT
VIN
Synchronous Switch Duty Cycle =
VIN − VOUT
VIN
The MOSFET power dissipations at maximum output
current are given by:
V
2
PMAIN = OUT (IMAX ) (1+ δ) RDS(ON) +
VIN

2 I
 (R ) (C
( VIN)  MAX
)•
2  DR MILLER

1
1 
+

( f)
 VINTVCC – VTHMIN VTHMIN 
PSYNC =
VIN – VOUT
2
IMAX ) (1+ δ) RDS(ON)
(
VIN
where δ is the temperature dependency of RDS(ON) and
RDR (approximately 2Ω) is the effective driver resistance
at the MOSFET’s Miller threshold voltage. VTHMIN is the
typical MOSFET minimum threshold voltage.
Both MOSFETs have I2R losses while the topside N-channel
equation includes an additional term for transition losses,
which are highest at high input voltages. For VIN < 20V
the high current efficiency generally improves with larger
MOSFETs, while for VIN > 20V the transition losses rapidly
increase to the point that the use of a higher RDS(ON) device
with lower CMILLER actually provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during
a short-circuit when the synchronous switch is on close
to 100% of the period.
The term (1+ δ) is generally given for a MOSFET in the
form of a normalized RDS(ON) vs Temperature curve, but
δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs.
The optional Schottky diodes D1 and D2 shown in Figure 10
conduct during the dead-time between the conduction of
the two power MOSFETs. This prevents the body diode of
the bottom MOSFET from turning on, storing charge during
the dead-time and requiring a reverse recovery period that
could cost as much as 3% in efficiency at high VIN. A 1A
to 3A Schottky is generally a good compromise for both
regions of operation due to the relatively small average
current. Larger diodes result in additional transition losses
due to their larger junction capacitance.
CIN and COUT Selection
The selection of CIN is simplified by the 2-phase architecture and its impact on the worst-case RMS current drawn
through the input network (battery/fuse/capacitor). It can be
shown that the worst-case capacitor RMS current occurs
when only one controller is operating. The controller with
the highest (VOUT)(IOUT) product needs to be used in the
formula shown in Equation 1 to determine the maximum
RMS capacitor current requirement. Increasing the output current drawn from the other controller will actually
decrease the input RMS ripple current from its maximum
value. The out-of-phase technique typically reduces the
input capacitor’s RMS ripple current by a factor of 30%
to 70% when compared to a single phase power supply
solution.
In continuous mode, the source current of the top MOSFET
is a square wave of duty cycle (VOUT)/(VIN). To prevent
large voltage transients, a low ESR capacitor sized for the
maximum RMS current of one channel must be used. The
maximum RMS capacitor current is given by:
CIN Required IRMS ≈
IMAX 
1/2 (1)
V
V
–
V
(
)
(
)
VIN  OUT IN OUT 
38681fb
19
LTC3868-1
Applications Information
Equation 1 has a maximum at VIN = 2VOUT , where IRMS
= IOUT/2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturers’ ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet
size or height requirements in the design. Due to the high
operating frequency of the LTC3868-1, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
The benefit of the LTC3868-1 2-phase operation can be
calculated by using Equation 1 for the higher power controller and then calculating the loss that would have resulted
if both controller channels switched on at the same time.
The total RMS power lost is lower when both controllers
are operating due to the reduced overlap of current pulses
required through the input capacitor’s ESR. This is why
the input capacitor’s requirement calculated above for the
worst-case controller is adequate for the dual controller
design. Also, the input protection fuse resistance, battery
resistance, and PC board trace resistance losses are also
reduced due to the reduced peak currents in a 2-phase
system. The overall benefit of a multiphase design will
only be fully realized when the source impedance of the
power supply/battery is included in the efficiency testing.
The sources of the top MOSFETs should be placed within
1cm of each other and share a common CIN(s). Separating
the sources and CIN may produce undesirable voltage and
current resonances at VIN.
A small (0.1µF to 1µF) bypass capacitor between the chip
VIN pin and ground, placed close to the LTC3868-1, is
also suggested. A 10Ω resistor placed between CIN (C1)
and the VIN pin provides further isolation between the
two channels.
The selection of COUT is driven by the effective series
resistance (ESR). Typically, once the ESR requirement
is satisfied, the capacitance is adequate for filtering. The
output ripple (∆VOUT) is approximated by:
ΔVOUT


1
≈ ΔIL ESR +

8 • f • COUT 

where f is the operating frequency, COUT is the output
capacitance and ∆IL is the ripple current in the inductor.
The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.
Setting Output Voltage
The LTC3868-1 output voltages are each set by an external feedback resistor divider carefully placed across the
output, as shown in Figure 6. The regulated output voltage
is determined by:
 R 
VOUT = 0.8V 1+ B 
 RA 
To improve the frequency response, a feedforward capacitor, CFF , may be used. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
VOUT
RB
1/2 LTC3868-1
CFF
VFB
RA
38681 F05
Figure 6. Setting Output Voltage
Soft-Start (SS Pins)
The start-up of each VOUT is controlled by the voltage on
the respective SS pin. When the voltage on the SS pin
is less than the internal 0.8V reference, the LTC3868-1
regulates the VFB pin voltage to the voltage on the SS pin
instead of 0.8V. The SS pin can be used to program an
external soft-start function.
Soft-start is enabled by simply connecting a capacitor from
the SS pin to ground, as shown in Figure 7. An internal
1µA current source charges the capacitor, providing a
1/2 LTC3868-1
SS
CSS
SGND
38681 F06
Figure 7. Using the SS Pin to Program Soft-Start
38681fb
20
LTC3868-1
Applications Information
linear ramping voltage at the SS pin. The LTC3868-1 will
regulate the VFB pin (and hence VOUT) according to the
voltage on the SS pin, allowing VOUT to rise smoothly from
0V to its final regulated value. The total soft-start time will
be approximately:
0.8 V
tSS = CSS •
1µA
INTVCC Regulators
The LTC3868-1 features two separate internal P-channel
low dropout linear regulators (LDO) that supply power
at the INTVCC pin from either the VIN supply pin or the
EXTVCC pin depending on the connection of the EXTVCC
pin. INTVCC powers the gate drivers and much of the
LTC3868-1’s internal circuitry. The VIN LDO and the EXTVCC
LDO regulate INTVCC to 5.1V. Each of these can supply a
peak current of 50mA and must be bypassed to ground
with a minimum of 4.7µF low ESR capacitor. No matter
what type of bulk capacitor is used, an additional 1µF ceramic capacitor placed directly adjacent to the INTVCC and
PGND IC pins is highly recommended. Good bypassing
is needed to supply the high transient currents required
by the MOSFET gate drivers and to prevent interaction
between the channels.
High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the
maximum junction temperature rating for the LTC3868-1
to be exceeded. The INTVCC current, which is dominated
by the gate charge current, may be supplied by either
the VIN LDO or the EXTVCC LDO. When the voltage on
the EXTVCC pin is less than 4.7V, the VIN LDO is enabled.
Power dissipation for the IC in this case is highest and is
equal to VIN • IINTVCC. The gate charge current is dependent on operating frequency as discussed in the Efficiency
Considerations section. The junction temperature can be
estimated by using the equations given in Note 2 of the
Electrical Characteristics. For example, the LTC3868-1
INTVCC current is limited to less than 22mA from a 28V
supply when not using the EXTVCC supply at 70°C ambient
temperature in the SSOP package:
To prevent the maximum junction temperature from being exceeded, the input supply current must be checked
while operating in forced continuous mode (PLLIN/MODE
= INTVCC) at maximum VIN.
When the voltage applied to EXTVCC rises above 4.7V, the
VIN LDO is turned off and the EXTVCC LDO is enabled. The
EXTVCC LDO remains on as long as the voltage applied to
EXTVCC remains above 4.5V. The EXTVCC LDO attempts
to regulate the INTVCC voltage to 5.1V, so while EXTVCC
is less than 5.1V, the LDO is in dropout and the INTVCC
voltage is approximately equal to EXTVCC. When EXTVCC
is greater than 5.1V, up to an absolute maximum of 14V,
INTVCC is regulated to 5.1V.
Using the EXTVCC LDO allows the MOSFET driver and
control power to be derived from one of the LTC3868-1’s
switching regulator outputs (4.7V ≤ VOUT ≤ 14V) during
normal operation and from the VIN LDO when the output is out of regulation (e.g., start-up, short-circuit). If
more current is required through the EXTVCC LDO than
is specified, an external Schottky diode can be added
between the EXTVCC and INTVCC pins. In this case, do
not apply more than 6V to the EXTVCC pin and make sure
that EXTVCC ≤ VIN.
Significant efficiency and thermal gains can be realized
by powering INTVCC from the output, since the VIN current resulting from the driver and control currents will be
scaled by a factor of (Duty Cycle)/(Switcher Efficiency).
For 5V to 14V regulator outputs, this means connecting
the EXTVCC pin directly to VOUT . Tying the EXTVCC pin to
a 8.5V supply reduces the junction temperature in the
previous example from 125°C to:
TJ = 70°C + (45mA)(8.5V)(90°C/W) = 87°C
However, for 3.3V and other low voltage outputs, additional circuitry is required to derive INTVCC power from
the output.
TJ = 70°C + (22mA)(28V)(90°C/W) = 125°C
38681fb
21
LTC3868-1
Applications Information
The following list summarizes the four possible connections for EXTVCC:
1. EXTVCC Left Open (or Grounded). This will cause INTVCC
to be powered from the internal 5.1V regulator resulting in an efficiency penalty of up to 10% at high input
voltages.
2. EXTVCC Connected Directly to VOUT . This is the normal
connection for a 5V to 14V regulator and provides the
highest efficiency.
3. EXTVCC Connected to an External Supply. If an external
supply is available in the 5V to 14V range, it may be
used to power EXTVCC. Ensure that EXTVCC < VIN.
4. EXTVCC Connected to an Output-Derived Boost Network.
For 3.3V and other low voltage regulators, efficiency
gains can still be realized by connecting EXTVCC to an
output-derived voltage that has been boosted to greater
than 4.7V. This can be done with the capacitive charge
pump shown in Figure 8. Ensure that EXTVCC < VIN.
CIN
BAT85
VIN
BAT85
MTOP
VN2222LL
TG1
1/2 LTC3868-1
EXTVCC
L
SW
RSENSE
BAT85
VOUT
MBOT
BG1
D
PGND
COUT
38681 F08
Figure 8. Capacitive Charge Pump for EXTVCC
Topside MOSFET Driver Supply (CB, DB)
External bootstrap capacitors, CB, connected to the BOOST
pins supply the gate drive voltages for the topside MOSFETs.
Capacitor CB in the Functional Diagram is charged though
external diode DB from INTVCC when the SW pin is low.
When one of the topside MOSFETs is to be turned on, the
driver places the CB voltage across the gate-source of the
desired MOSFET. This enhances the top MOSFET switch
and turns it on. The switch node voltage, SW, rises to VIN
and the BOOST pin follows. With the topside MOSFET
on, the boost voltage is above the input supply: VBOOST =
VIN + VINTVCC. The value of the boost capacitor, CB, needs
to be 100 times that of the total input capacitance of the
topside MOSFET(s). The reverse breakdown of the external
Schottky diode must be greater than VIN(MAX).
When adjusting the gate drive level, the final arbiter is the
total input current for the regulator. If a change is made
and the input current decreases, then the efficiency has
improved. If there is no change in input current, then there
is no change in efficiency.
Fault Conditions: Current Limit and Current Foldback
When the output current hits the current limit, the output
voltage begins to drop. If the output voltage falls below
70% of its nominal output level, then the maximum sense
voltage is progressively lowered to about one-half of its
maximum selected value. Under short-circuit conditions
with very low duty cycles, the LTC3868-1 will begin cycle
skipping in order to limit the short-circuit current. In this
situation the bottom MOSFET will be dissipating most of
the power but less than in normal operation. The shortcircuit ripple current is determined by the minimum ontime, tON(MIN), of the LTC3868-1 (≈90ns), the input voltage
and inductor value:
V 
ΔIL(SC) = tON(MIN)  IN 
 L 
The resulting average short-circuit current is:
ISC =
50% • ILIM(MAX)
RSENSE
1
– ∆IL(SC)
2
Fault Conditions: Overvoltage Protection (Crowbar)
The overvoltage crowbar is designed to blow a system
input fuse when the output voltage of the regulator rises
much higher than nominal levels. The crowbar causes huge
currents to flow, that blow the fuse to protect against a
shorted top MOSFET if the short occurs while the controller is operating.
A comparator monitors the output for overvoltage conditions. The comparator detects faults greater than 10%
38681fb
22
LTC3868-1
Applications Information
above the nominal output voltage. When this condition
is sensed, the top MOSFET is turned off and the bottom
MOSFET is turned on until the overvoltage condition is
cleared. The bottom MOSFET remains on continuously
for as long as the overvoltage condition persists; if VOUT
returns to a safe level, normal operation automatically
resumes.
1000
900
FREQUENCY (kHz)
800
A shorted top MOSFET will result in a high current condition
which will open the system fuse. The switching regulator
will regulate properly with a leaky top MOSFET by altering
the duty cycle to accommodate the leakage.
Phase-Locked Loop and Frequency Synchronization
The LTC3868-1 has an internal phase-locked loop (PLL)
comprised of a phase frequency detector, a lowpass filter,
and a voltage-controlled oscillator (VCO). This allows the
turn-on of the top MOSFET of controller 1 to be locked to
the rising edge of an external clock signal applied to the
PLLIN/MODE pin. The turn-on of controller 2’s top MOSFET
is thus 180 degrees out of phase with the external clock.
The phase detector is an edge sensitive digital type that
provides zero degrees phase shift between the external
and internal oscillators. This type of phase detector does
not exhibit false lock to harmonics of the external clock.
If the external clock frequency is greater than the internal
oscillator’s frequency, fOSC, then current is sourced continuously from the phase detector output, pulling up the VCO
input. When the external clock frequency is less than fOSC,
current is sunk continuously, pulling down the VCO input.
If the external and internal frequencies are the same but
exhibit a phase difference, the current sources turn on for
an amount of time corresponding to the phase difference.
The voltage at the VCO input is adjusted until the phase
and frequency of the internal and external oscillators are
identical. At the stable operating point, the phase detector
output is high impedance and the internal filter capacitor,
CLP , holds the voltage at the VCO input.
Typically, the external clock (on the PLLIN/MODE pin)
input high threshold is 1.6V, while the input low threshold
is 1.1V.
700
600
500
400
300
200
100
0
15 25 35 45 55 65 75 85 95 105 115 125
FREQ PIN RESISTOR (kΩ)
38681 F09
Figure 9. Relationship Between Oscillator Frequency
and Resistor Value at the FREQ Pin
Rapid phase locking can be achieved by using the FREQ
pin to set a free-running frequency near the desired
synchronization frequency. The VCO’s input voltage is
prebiased at a frequency corresponding to the frequency
set by the FREQ pin. Once prebiased, the PLL only needs
to adjust the frequency slightly to achieve phase lock
and synchronization. Although it is not required that the
free-running frequency be near external clock frequency,
doing so will prevent the operating frequency from passing
through a large range of frequencies as the PLL locks.
Note that the LTC3868-1 can only be synchronized to an
external clock whose frequency is within range of the
LTC3868-1’s internal VCO, which is nominally 55kHz
to 1MHz. This is guaranteed to be between 75kHz and
850kHz.
Table 2 summarizes the different states in which the FREQ
pin can be used.
Table 2
FREQ PIN
PLLIN/MODE PIN
FREQUENCY
0V
DC Voltage
350kHz
INTVCC
DC Voltage
535kHz
Resistor
DC Voltage
50kHz–900kHz
Any of the Above
External Clock
Phase–Locked to
External Clock
38681fb
23
LTC3868-1
Applications Information
Minimum On-Time Considerations
Minimum on-time, tON(MIN), is the smallest time duration that the LTC3868-1 is capable of turning on the top
MOSFET. It is determined by internal timing delays and the
gate charge required to turn on the top MOSFET. Low duty
cycle applications may approach this minimum on-time
limit and care should be taken to ensure that:
tON(MIN) <
VOUT
VIN f
()
If the duty cycle falls below what can be accommodated
by the minimum on-time, the controller will begin to skip
cycles. The output voltage will continue to be regulated,
but the ripple voltage and current will increase.
The minimum on-time for the LTC3868-1 is approximately
95ns. However, as the peak sense voltage decreases the
minimum on-time gradually increases up to about 130ns.
This is of particular concern in forced continuous applications with low ripple current at light loads. If the duty cycle
drops below the minimum on-time limit in this situation,
a significant amount of cycle skipping can occur with correspondingly larger current and voltage ripple.
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage of input power.
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of
the losses in LTC3868-1 circuits: 1) IC VIN current, 2)
INTVCC regulator current, 3) I2R losses, 4) topside MOSFET
transition losses.
1. The VIN current is the DC input supply current given
in the Electrical Characteristics table, which excludes
MOSFET driver and control currents. VIN current typically results in a small (<0.1%) loss.
2. INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge, dQ, moves
from INTVCC to ground. The resulting dQ/dt is a current
out of INTVCC that is typically much larger than the
control circuit current. In continuous mode, IGATECHG
= f(QT + QB), where QT and QB are the gate charges of
the topside and bottom side MOSFETs.
Supplying INTVCC from an output-derived power source
through EXTVCC will scale the VIN current required
for the driver and control circuits by a factor of (Duty
Cycle)/(Efficiency). For example, in a 20V to 5V application, 10mA of INTVCC current results in approximately
2.5mA of VIN current. This reduces the midcurrent loss
from 10% or more (if the driver was powered directly
from VIN) to only a few percent.
3. I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor, and input and output capacitor ESR. In continuous
mode the average output current flows through L and
RSENSE, but is chopped between the topside MOSFET
and the synchronous MOSFET. If the two MOSFETs have
approximately the same RDS(ON), then the resistance
of one MOSFET can simply be summed with the resistances of L, RSENSE and ESR to obtain I2R losses. For
example, if each RDS(ON) = 30mΩ, RL = 50mΩ, RSENSE
= 10mΩ and RESR = 40mΩ (sum of both input and
output capacitance losses), then the total resistance
is 130mΩ. This results in losses ranging from 3% to
13% as the output current increases from 1A to 5A for
a 5V output, or a 4% to 20% loss for a 3.3V output.
Efficiency varies as the inverse square of VOUT for the
same external components and output power level. The
combined effects of increasingly lower output voltages
and higher currents required by high performance digital
systems is not doubling but quadrupling the importance
of loss terms in the switching regulator system!
38681fb
24
LTC3868-1
Applications Information
4. Transition losses apply only to the topside MOSFET(s),
and become significant only when operating at high
input voltages (typically 15V or greater). Transition
losses can be estimated from:
Transition Loss = (1.7) • VIN • 2 • IO(MAX) • CRSS • f
Other hidden losses such as copper trace and internal
battery resistances can account for an additional 5%
to 10% efficiency degradation in portable systems. It
is very important to include these system level losses
during the design phase. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at
the switching frequency. A 25W supply will typically
require a minimum of 20µF to 40µF of capacitance
having a maximum of 20mΩ to 50mΩ of ESR. The
LTC3868-1 2-phase architecture typically halves this
input capacitance requirement over competing solutions. Other losses including Schottky conduction losses
during dead-time and inductor core losses generally
account for less than 2% total additional loss.
Checking Transient Response
The regulator loop response can be checked by looking at
the load current transient response. Switching regulators
take several cycles to respond to a step in DC (resistive)
load current. When a load step occurs, VOUT shifts by
an amount equal to ∆ILOAD (ESR), where ESR is the effective series resistance of COUT . ∆ILOAD also begins to
charge or discharge COUT generating the feedback error
signal that forces the regulator to adapt to the current
change and return VOUT to its steady-state value. During
this recovery time VOUT can be monitored for excessive
overshoot or ringing, which would indicate a stability
problem. OPTI-LOOP compensation allows the transient
response to be optimized over a wide range of output
capacitance and ESR values. The availability of the ITH pin
not only allows optimization of control loop behavior, but
it also provides a DC coupled and AC filtered closed-loop
response test point. The DC step, rise time and settling
at this test point truly reflects the closed-loop response.
Assuming a predominantly second order system, phase
margin and/or damping factor can be estimated using the
percentage of overshoot seen at this pin. The bandwidth
can also be estimated by examining the rise time at the
pin. The ITH external components shown in Figure 12
circuit will provide an adequate starting point for most
applications.
The ITH series RC-CC filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and
the particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
gain and phase. An output current pulse of 20% to 80%
of full-load current having a rise time of 1µs to 10µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop.
Placing a resistive load and a power MOSFET directly
across the output capacitor and driving the gate with an
appropriate signal generator is a practical way to produce
a realistic load step condition. The initial output voltage
step resulting from the step change in output current may
not be within the bandwidth of the feedback loop, so this
signal cannot be used to determine phase margin. This
is why it is better to look at the ITH pin signal which is in
the feedback loop and is the filtered and compensated
control loop response.
The gain of the loop will be increased by increasing RC
and the bandwidth of the loop will be increased by decreasing CC. If RC is increased by the same factor that CC
is decreased, the zero frequency will be kept the same,
thereby keeping the phase shift the same in the most
critical frequency range of the feedback loop. The output
voltage settling behavior is related to the stability of the
closed-loop system and will demonstrate the actual overall
supply performance.
A second, more severe transient is caused by switching
in loads with large (>1µF) supply bypass capacitors. The
discharged bypass capacitors are effectively put in parallel
with COUT , causing a rapid drop in VOUT . No regulator can
alter its delivery of current quickly enough to prevent this
sudden step change in output voltage if the load switch
resistance is low and it is driven quickly. If the ratio of
38681fb
25
LTC3868-1
Applications Information
CLOAD to COUT is greater than 1:50, the switch rise time
should be controlled so that the load rise time is limited
to approximately 25 • CLOAD. Thus a 10µF capacitor would
require a 250µs rise time, limiting the charging current
to about 200mA.
Design Example
The power dissipation on the topside MOSFET can be easily
estimated. Choosing a Fairchild FDS6982S dual MOSFET
results in: RDS(ON) = 0.035Ω/0.022Ω, CMILLER = 215pF. At
maximum input voltage with T(estimated) = 50°C:
PMAIN =
ΔIL(NOM) =

VOUT 
V
1– OUT 
( f) (L)  VIN(NOM) 
A 3.9µH inductor will produce 29% ripple current. The
peak inductor current will be the maximum DC value plus
one half the ripple current, or 6.88A. Increasing the ripple
current will also help ensure that the minimum on-time
of 95ns is not violated. The minimum on-time occurs at
maximum VIN:
VOUT
3.3V
tON(MIN) =
=
= 4299ns
VIN(MAX ) ( f) 22V (350kHz )
The equivalent RSENSE resistor value can be calculated by
using the minimum value for the maximum current sense
threshold (43mV):
RSENSE ≤
43mV
= 0.006Ω
6.88A
( )
(
As a design example for one channel, assume VIN =
12V(nominal), VIN = 22V (max), VOUT = 3.3V, IMAX = 6A,
VSENSE(MAX) = 50mV and f = 350kHz.
The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQ pin
to GND, generating 350kHz operation. The minimum
inductance for 30% ripple current is:
2
3.3V
6A 1+ 0.005 50°C – 25°C 
22V
2 6A
2.5Ω 215pF •
0.035Ω + 22V
2

1 
1
 5V – 2.3V + 2.3V  350kHz = 433mW


) (
(
)(
)
(
(
)
)(
)
)
A short-circuit to ground will result in a folded back current of:
ISC =
(
)
25mV
1  95ns 22V 
– 
 = 3.9A
0.006Ω 2  3.9µH 
with a typical value of RDS(ON) and δ = (0.005/°C)(25°C)
= 0.125. The resulting power dissipated in the bottom
MOSFET is:
(
PSYNC = 3.9A
)2 (1.125)(0.022Ω) = 376mW
which is less than under full-load conditions.
CIN is chosen for an RMS current rating of at least 3A at
temperature assuming only this channel is on. COUT is
chosen with an ESR of 0.02Ω for low output ripple. The
output ripple in continuous mode will be highest at the
maximum input voltage. The output voltage ripple due to
ESR is approximately:
VORIPPLE = RESR(∆IL) = 0.02Ω(1.75A) = 35mVP-P
Choosing 1% resistors: RA = 25k and RB = 78.1k yields
an output voltage of 3.299V.
38681fb
26
LTC3868-1
Applications Information
PC Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the IC. These items are also illustrated graphically in the
layout diagram of Figure 10. Figure 11 illustrates the current
waveforms present in the various branches of the 2-phase
synchronous regulators operating in the continuous mode.
Check the following in your layout:
1. Are the top N-channel MOSFETs MTOP1 and MTOP2
located within 1cm of each other with a common drain
connection at CIN? Do not attempt to split the input
decoupling for the two channels as it can cause a large
resonant loop.
2. Are the signal and power grounds kept separate? The
combined IC signal ground pin and the ground return
of CINTVCC must return to the combined COUT (–) terminals. The path formed by the top N-channel MOSFET,
Schottky diode and the CIN capacitor should have short
leads and PC trace lengths. The output capacitor (–)
terminals should be connected as close as possible
to the (–) terminals of the input capacitor by placing
the capacitors next to each other and away from the
Schottky loop described above.
3. Do the LTC3868-1 VFB pins’ resistive dividers connect
to the (+) terminals of COUT? The resistive divider
must be connected between the (+) terminal of COUT
and signal ground. The feedback resistor connections
should not be along the high current input feeds from
the input capacitor(s).
v
4. Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The filter capacitor between
SENSE+ and SENSE– should be as close as possible
to the IC. Ensure accurate current sensing with Kelvin
connections at the SENSE resistor.
5. Is the INTVCC decoupling capacitor connected close
to the IC, between the INTVCC and the power ground
pins? This capacitor carries the MOSFET drivers’ current peaks. An additional 1µF ceramic capacitor placed
immediately next to the INTVCC and PGND pins can help
improve noise performance substantially.
6. Keep the switching nodes (SW1, SW2), top gate nodes
(TG1, TG2), and boost nodes (BOOST1, BOOST2) away
from sensitive small-signal nodes, especially from
the opposites channel’s voltage and current sensing
feedback pins. All of these nodes have very large and
fast moving signals and therefore should be kept on
the output side of the LTC3868-1 and occupy minimum
PC trace area.
7. Use a modified star ground technique: a low impedance,
large copper area central grounding point on the same
side of the PC board as the input and output capacitors
with tie-ins for the bottom of the INTVCC decoupling
capacitor, the bottom of the voltage feedback resistive
divider and the SGND pin of the IC.
PC Board Layout Debugging
Start with one controller on at a time. It is helpful to use
a DC-50MHz current probe to monitor the current in the
inductor while testing the circuit. Monitor the output
switching node (SW pin) to synchronize the oscilloscope
to the internal oscillator and probe the actual output voltage
as well. Check for proper performance over the operating
voltage and current range expected in the application. The
frequency of operation should be maintained over the input
voltage range down to dropout and until the output load
drops below the low current operation threshold—typically 10% of the maximum designed current level in Burst
Mode operation.
The duty cycle percentage should be maintained from cycle
to cycle in a well-designed, low noise PCB implementation.
Variation in the duty cycle at a subharmonic rate can suggest noise pickup at the current or voltage sensing inputs
or inadequate loop compensation. Overcompensation of
the loop can be used to tame a poor PC layout if regulator bandwidth optimization is not required. Only after
each controller is checked for its individual performance
should both controllers be turned on at the same time.
A particularly difficult region of operation is when one
controller channel is nearing its current comparator trip
point when the other channel is turning on its top MOSFET.
This occurs around 50% duty cycle on either channel due
to the phasing of the internal clocks and may cause minor
duty cycle jitter.
38681fb
27
LTC3868-1
Applications Information
Reduce VIN from its nominal level to verify operation
of the regulator in dropout. Check the operation of the
undervoltage lockout circuit by further lowering VIN while
monitoring the outputs to verify operation.
Investigate whether any problems exist only at higher output currents or only at higher input voltages. If problems
coincide with high input voltavges and low output currents,
look for capacitive coupling between the BOOST, SW, TG,
and possibly BG connections and the sensitive voltage
and current pins. The capacitor placed across the current
sensing pins needs to be placed immediately adjacent to
the pins of the IC. This capacitor helps to minimize the
effects of differential noise injection due to high frequency
capacitive coupling. If problems are encountered with
high current output loading at lower input voltages, look
for inductive coupling between CIN, Schottky and the top
MOSFET components to the sensitive current and voltage
sensing traces. In addition, investigate common ground
path voltage pickup between these components and the
SGND pin of the IC.
An embarrassing problem, which can be missed in an
otherwise properly working switching regulator, results
when the current sensing leads are hooked up backwards.
The output voltage under this improper hookup will still
be maintained but the advantages of current mode control
will not be realized. Compensation of the voltage loop will
be much more sensitive to component selection. This
behavior can be investigated by temporarily shorting out
the current sensing resistor—don’t worry, the regulator
will still maintain control of the output voltage.
SS1
LTC3868-1
ITH1
VFB1
RPU1
PGOOD1
PGOOD1
VPULL-UP
(<6V)
L1
SENSE1+
TG1
SENSE1–
SW1
CB1
FREQ
M1
BOOST1
M2
RSENSE
VOUT1
D1
BG1
RUN1
RUN2
EXTVCC
INTVCC
SENSE2+
BG2
ITH2
SS2
RIN
1µF
CERAMIC
COUT1
PGND
SENSE2–
VFB2
CVIN
VOUT1
CINTVCC
VIN
CIN
COUT2
1µF
CERAMIC
M3
BOOST2
SW2
GND
+
SGND
VIN
+
PLLIN/MODE
+
fIN
M4
D2
CB2
RSENSE
TG2
VOUT2
L2
38681 F10
Figure 10. Recommended Printed Circuit Layout Diagram
38681fb
28
LTC3868-1
Applications Information
SW1
L1
D1
RSENSE1
VOUT1
COUT1
RL1
VIN
RIN
CIN
SW2
BOLD LINES INDICATE
HIGH SWITCHING
CURRENT. KEEP LINES
TO A MINIMUM LENGTH.
D2
L2
RSENSE2
VOUT2
COUT2
RL2
38681 F11
Figure 11. Branch Current Waveforms
38681fb
29
LTC3868-1
Typical ApplicationS
RB1
215k
CF1
15pF
LTC3868-1
SENSE1+
C1
1nF
RA1
68.1k
SENSE1–
VFB1
CITH1A 150pF
INTVCC
PGOOD1
100k
MBOT1
BG1
SW1
CITH1 820pF
ITH1
CSS1 0.1µF
D1
INTVCC
PLLIN/MODE
SGND
TG2
EXTVCC
RUN1
BOOST2
RUN2
FREQ
RITH2 27k
CITH2A 100pF
SW2
ITH2
BG2
VIN
9V TO 24V
D2
MTOP2
CB2
0.47µF
L2
7.2µH
RSENSE2
10mΩ
MBOT2
VOUT2
8.5V
3A
COUT2
150µF
VFB2
SENSE2–
C2
1nF
CF2
39pF
SS2
CIN
22µF
CINT
4.7µF
PGND
CSS2 0.1µF
COUT1
150µF
VOUT1
3.3V
5A
MTOP1
TG1
VIN
SS1
RA2
44.2k
RSENSE1
7mΩ
CB1
0.47µF
BOOST1
RITH1 15k
CITH2 680pF
L1
3.3µH
SENSE2+
RB2
442k
38681 F12
COUT1, COUT2: SANYO 10TPD150M
L1: SUMIDA CDEP105-3R2M
L2: SUMIDA CDEP105-7R2M
MTOP1, MTOP2, MBOT1, MBOT2: VISHAY Si7848DP
Start-Up
Efficiency vs Output Current
SW Node Waveforms
100
90
EFFICIENCY (%)
80
70
VOUT = 8.5V
VOUT2
2V/DIV
VOUT = 3.3V
SW1
5V/DIV
60
50
VOUT1
2V/DIV
40
SW2
5V/DIV
30
20
10
VIN = 12V
Burst Mode OPERATION
0
0.00001 0.0001 0.001 0.01
0.1
OUTPUT CURRENT (A)
1
10
20ms/DIV
38681 F12c
1µs/DIV
38681 F12d
38681 F12b
Figure 12. High Efficiency Dual 8.5V/3.3V Step-Down Converter
38681fb
30
LTC3868-1
Typical ApplicationS
High Efficiency Dual 2.5V/3.3V Step-Down Converter
RB1
143k
CF1
22pF
RA1
68.1k
C1
1nF
LTC3868-1
SENSE1+
SENSE1–
VFB1
CITH1A 100pF
INTVCC
PGOOD1
100k
MBOT1
BG1
SW1
RITH1 22k
CITH1 820pF
ITH1
CSS1 0.01µF
RITH2 15k
CITH2A 150pF
RA2
68.1k
CF2
15pF
C2
1nF
COUT1
150µF
VOUT1
2.5V
5A
MTOP1
TG1
D1
VIN
INTVCC
PLLIN/MODE
SGND
TG2
EXTVCC
RUN1
BOOST2
RUN2
FREQ
SS2
SW2
ITH2
BG2
CIN
22µF
CINT
4.7µF
PGND
CSS2 0.01µF
RSENSE1
7mΩ
CB1
0.47µF
BOOST1
SS1
CITH2 820pF
L1
2.4µH
VIN
4V TO 24V
D2
MTOP2
CB2
0.47µF
L2
3.2µH
RSENSE2
7mΩ
MBOT2
VOUT2
3.3V
COUT2 5A
150µF
VFB2
SENSE2–
SENSE2+
RB2
215k
COUT1, COUT2: SANYO 10TPD150M
L1: SUMIDA CDEP105-2R5
L2: SUMIDA CDEP105-3R2M
MTOP1, MTOP2, MBOT1, MBOT2: VISHAY Si7848DP
38681 F13
38681fb
31
LTC3868-1
Typical ApplicationS
High Efficiency Dual 12V/5V Step-Down Converter
RB1
422k
CF1
33pF
RA1
34k
C1
1nF
SENSE1+
SENSE1–
INTVCC
PGOOD1
100k
MBOT1
BG1
VFB1
CITH1A 100pF
SW1
RITH1 33k
CITH1 680pF
CSS1 0.01µF
D1
LTC3868-1
VIN
INTVCC
RFREQ
60k
CSS2 0.01µF
RITH2 17k
CITH2A 100pF
RA2
75k
CF2
15pF
C2
1nF
PGND
PLLIN/MODE
SGND
TG2
EXTVCC
RUN1
BOOST2
RUN2
FREQ
SS2
SW2
ITH2
BG2
VFB2
SENSE2–
SENSE2+
COUT1
47µF
VOUT1
12V
3A
MTOP1
TG1
ITH1
RSENSE1
10mΩ
CB1
0.47µF
BOOST1
SS1
CITH2 680pF
L1
8.8µH
CIN
22µF
CINT
4.7µF
VIN
12.5V TO 24V
D2
MTOP2
CB2
0.47µF
L2
4.3µH
RSENSE2
7mΩ
MBOT2
VOUT2
5V
COUT2 5.5A
150µF
COUT1: KEMET T525D476M016E035
COUT2: SANYO 10TPD150M
L1: SUMIDA CDEP105-8R8M
L2: SUMIDA CDEP105-4R3M
MTOP1, MTOP2, MBOT1, MBOT2: VISHAY Si7848DP
RB2
393k
38681 TA02a
38681fb
32
LTC3868-1
Typical ApplicationS
High Efficiency Dual 1V/1.2V Step-Down Converter
RB1
28.7k
CF1
56pF
RA1
115k
C1
1nF
SENSE1+
SENSE1–
INTVCC
PGOOD1
L1
MBOT1 0.47µH
BG1
VFB1
CITH1A 220pF
SW1
RITH1 3.93k
CITH1 1000pF
RFREQ
60k
CSS2 0.01µF
RITH2 3.43k
CITH2A 220pF
RA2
115k
VIN
CF2
56pF
RB2
57.6k
C2
1nF
INTVCC
PGND
PLLIN/MODE
SGND
TG2
EXTVCC
RUN1
BOOST2
RUN2
FREQ
SS2
SW2
ITH2
BG2
VFB2
SENSE2–
SENSE2+
COUT1
220µF
×2
VOUT1
1V
8A
D1
LTC3868-1
CSS1 0.01µF
MTOP1
TG1
ITH1
RSENSE1
4mΩ
CB1
0.47µF
BOOST1
SS1
CITH2 1000pF
100k
CIN
22µF
CINT
4.7µF
VIN
12V
D2
MTOP2
CB2
0.47µF
L2
0.47µH
MBOT2
RSENSE2
4mΩ
VOUT2
1.2V
COUT2 8A
220µF
×2
COUT1, COUT2: SANYO 2RSTPE220M
L1: SUMIDA CDEP105-3R2M
L2: SUMIDA CDEP105-7R2M
MTOP1, MTOP2: RENESAS RJK0305
MBOT1, MBOT2: RENESAS RJK0328
38681 TA03a
38681fb
33
LTC3868-1
Typical ApplicationS
High Efficiency Dual 1V/1.2V Step-Down Converter with Inductor DCR Current Sensing
RB1
28.7k
CF1
56pF
RA1
115k
RS1 1.18k
C1
0.1µF
SENSE1+
SENSE1–
INTVCC
PGOOD1
CITH1A 200pF
L1
0.47µH
MBOT1
BG1
VFB1
SW1
RITH1 3.93k
CITH1 1000pF
RFREQ
65k
CSS2 0.01µF
RITH2 3.93k
CITH2A 220pF
RA2
115k
CF2
56pF
RB2
57.6k
C2
0.1µF
VOUT1
1V
8A
D1
LTC3868-1
CSS1 0.01µF
MTOP1
TG1
ITH1
COUT1
220µF
×2
CB1
0.47µF
BOOST1
VIN
SS1
CITH2 1000pF
100k
INTVCC
PGND
PLLIN/MODE
SGND
TG2
EXTVCC
RUN1
BOOST2
RUN2
FREQ
SS2
SW2
ITH2
BG2
CIN
22µF
CINT
4.7µF
VIN
12V
D2
MTOP2
CB2
0.47µF
L2
0.47µH
VOUT2
1.2V
COUT2 8A
220µF
×2
MBOT2
VFB2
SENSE2–
SENSE2+
COUT1, COUT2: SANYO 2R5TPE220M
L1, L2: SUMIDA IHL P2525CZERR47M06
MTOP1, MTOP2: RENESAS RJK0305
MBOT1, MBOT2: RENESAS RJK0328
RS2 1.18k
38681 TA05
38681fb
34
LTC3868-1
Package Description
UFD Package
28-Lead Plastic QFN (4mm × 5mm)
(Reference LTC DWG # 05-08-1712 Rev B)
0.70 p0.05
4.50 p 0.05
3.10 p 0.05
2.50 REF
2.65 p 0.05
3.65 p 0.05
PACKAGE
OUTLINE
0.25 p0.05
0.50 BSC
3.50 REF
4.10 p 0.05
5.50 p 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 p 0.10
(2 SIDES)
0.75 p 0.05
R = 0.05
TYP
PIN 1 NOTCH
R = 0.20 OR 0.35
s 45o CHAMFER
2.50 REF
R = 0.115
TYP
27
28
0.40 p 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
5.00 p 0.10
(2 SIDES)
3.50 REF
3.65 p 0.10
2.65 p 0.10
(UFD28) QFN 0506 REV B
0.200 REF
0.00 – 0.05
0.25 p 0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WXXX-X).
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
38681fb
35
LTC3868-1
Package Description
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 p.005
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.033
(0.838)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 p.0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 p .004
s 45o
(0.38 p 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
.0532 – .0688
(1.35 – 1.75)
9 10 11 12 13 14
.004 – .0098
(0.102 – 0.249)
0o – 8o TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN28 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
38681fb
36
LTC3868-1
Revision History
(Revision history begins at Rev B)
REV
DATE
DESCRIPTION
PAGE NUMBER
B
12/09
Change to Absolute Maximum Ratings
Change to Electrical Characteristics
Change to Typical Performance Characteristics
Change to Pin Functions
Text Changes to Operation Section
Text Changes to Applications Information Section
2
3, 4
6
8, 9
11, 12, 13
21, 22, 23, 24, 26
Change to Figure 10
28
Changes to Related Parts
38
38681fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
37
LTC3868-1
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LTC3857/LTC3857-1
Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
Phase-Lockable Fixed Operating Frequency 50kHz to 900kHz,
4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 24V, IQ = 50µA,
LTC3858/LTC3858-1
Low IQ, Dual Output 2-Phase Synchronous Step-Down
DC/DC Controllers with 99% Duty Cycle
Phase-Lockable Fixed Operating Frequency 50kHz to 900kHz,
4V ≤ VIN ≤ 24V, 0.8V ≤ VOUT ≤ 14V, IQ = 170µA,
LTC3834/LTC3834-1
Low IQ, Synchronous Step-Down DC/DC Controllers
Phase-Lockable Fixed Operating Frequency 140kHz to 650kHz,
4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 30µA,
LTC3835/LTC3835-1
Low IQ, Synchronous Step-Down DC/DC Controllers
Phase-Lockable Fixed Operating Frequency 140kHz to 650kHz,
4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 10V, IQ = 80µA,
LT3845
Low IQ, High Voltage Synchronous Step-Down
DC/DC Controller
Adjustable Fixed Operating Frequency 100kHz to 500kHz,
4V ≤ VIN ≤ 60V, 1.23V ≤ VOUT ≤ 36V, IQ = 120µA, TSSOP-16
LT3800
Low IQ, High Voltage Synchronous Step-Down
DC/DC Controller
Fixed 200kHz Operating Frequency, 4V ≤ VIN ≤ 60V,
1.23V ≤ VOUT ≤ 36V, IQ = 100µA, TSSOP-16
LTC3824
Low IQ, High Voltage DC/DC Controller, 100% Duty Cycle
Selectable Fixed 200kHz to 600kHz Operating Frequency,
4V ≤ VIN ≤ 60V, 0.8V ≤ VOUT ≤ VIN, IQ = 40µA, MSOP-10E
LTC3850/LTC3850-1
LTC3850-2
Dual 2-Phase, High Efficiency Synchronous Step-Down
DC/DC Controllers, RSENSE or DCR Current Sensing and
Tracking
Phase-Lockable Fixed Operating Frequency 250kHz to 780kHz,
4V ≤ VIN ≤ 30V, 0.8V ≤ VOUT ≤ 5.25V
LTC3855
Dual, Multiphase, Synchronous DC/DC Step-Down
Controller with Diffamp and DCR Temperature
Compensation
Phase-Lockable Fixed Frequency 250kHz to 770kHz,
4.5V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 12.5V
LTC3853
Triple Output, Multiphase Synchronous Step-Down DC/DC Phase-Lockable Fixed Operating Frequency 250kHz to 750kHz,
Controller, RSENSE or DCR Current Sensing and Tracking
4V ≤ VIN ≤ 24V, VOUT Up to 13.5V
Small Footprint Wide VIN Range Synchronous Step-Down Fixed 400kHz Operating Frequency, 4.5V ≤ VIN ≤ 38V,
DC/DC Controller
0.8V ≤ VOUT ≤ 5.25V, 2mm × 3mm QFN-12, MSOP-12
LTC3854
LTC3775
High Frequency Synchronous Voltage Mode Step-Down
DC/DC Controller
Fast Transient Response, tON(MIN) = 30ns, 4V ≤ VIN ≤ 38V,
0.6V ≤ VOUT ≤ 0.8VIN, MSOP-16E, 3mm × 3mm QFN-16
LTC3851A/
LTC3851A-1
No RSENSE ™ Wide VIN Range Synchronous Step-Down
DC/DC Controllers
Phase-Lockable Fixed Operating Frequency 250kHz to 750kHz,
4V ≤ VIN ≤ 38V, 0.8V ≤ VOUT ≤ 5.25V, MSOP-16E, 3mm × 3mm
QFN-16, SSOP-16
LTC3878/LTC3879
No RSENSE Constant On-Time Synchronous Step-Down
DC/DC Controllers
Very Fast Transient Response, tON(MIN) = 43ns, 4V ≤ VIN ≤ 38V,
VOUT Up 90% of VIN, MSOP-16E, 3mm × 3mm QFN-16, SSOP-16
38681fb
38 Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
●
FAX: (408) 434-0507 ● www.linear.com
LT 0110 REV B • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 2009
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