Intersil CA3318CM Cmos video speed, 8-bit, flash a/d converter Datasheet

CA3318
CMOS Video Speed,
8-Bit, Flash A/D Converter
August 1997
Features
Description
• CMOS Low Power with SOS Speed (Typ). . . . . . . . 150mW
The CA3318 is a CMOS parallel (FLASH) analog-to-digital
converter designed for applications demanding both low
power consumption and high speed digitization.
• Parallel Conversion Technique
• 15MHz Sampling Rate (Conversion Time) . . . . . . . 67ns
• 8-Bit Latched Three-State Output with Overflow Bit
• Accuracy (Typ) . . . . . . . . . . . . . . . . . . . . . . . . . . ±1 LSB
• Single Supply Voltage . . . . . . . . . . . . . . . . . . 4V to 7.5V
• 2 Units in Series Allow 9-Bit Output
• 2 Units in Parallel Allow 30MHz Sampling Rate
Applications
•
•
•
•
•
•
•
•
•
•
TV Video Digitizing (Industrial/Security/Broadcast)
High Speed A/D Conversion
Ultrasound Signature Analysis
Transient Signal Analysis
High Energy Physics Research
General-Purpose Hybrid ADCs
Optical Character Recognition
Radar Pulse Analysis
Motion Signature Analysis
µP Data Acquisition Systems
The CA3318 operates over a wide full scale input voltage
range of 4V up to 7.5V with maximum power consumption
depending upon the clock frequency selected. When
operated from a 5V supply at a clock frequency of 15MHz,
the typical power consumption of the CA3318 is 150mW.
The intrinsic high conversion rate makes the CA3318 ideally
suited for digitizing high speed signals. The overflow bit
makes possible the connection of two or more CA3318s in
series to increase the resolution of the conversion system. A
series connection of two CA3318s may be used to produce a
9-bit high speed converter. Operation of two CA3318s in
parallel doubles the conversion speed (i.e., increases the
sampling rate from 15MHz to 30MHz).
256 paralleled auto balanced voltage comparators measure
the input voltage with respect to a known reference to
produce the parallel bit outputs in the CA3318.
255 comparators are required to quantize all input voltage
levels in this 8-bit converter, and the additional comparator is
required for the overflow bit.
Ordering Information
PART NUMBER LINEARITY (INL, DNL)
TEMP. RANGE (oC)
SAMPLING RATE
PACKAGE
PKG. NO.
CA3318CE
±1.5 LSB
15MHz (67ns)
-40 to 85
24 Ld PDIP
E24.6
CA3318CM
±1.5 LSB
15MHz (67ns)
-40 to 85
24 Ld SOIC
M24.3
CA3318CD
±1.5 LSB
15MHz (67ns)
-40 to 85
24 Ld SBDIP
D24.6
Pinout
CA3318
(PDIP, SBDIP, SOIC)
TOP VIEW
(LSB) B1 1
24 VAA + (ANA. SUP.)
B2 2
23 3/4R
B3 3
22 VREF +
B4 4
21 VIN
B5 5
20 p
B6 6
19 PHASE
B7 7
18 CLK
17 VAA - (ANA. GND)
(MSB) B8 8
16 VIN
OVERFLOW 9
1/ R 10
4
15 VREF -
(DIG. GND) VSS 11
14 CE1
(DIG. SUP.) VDD 12
13 CE2
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
http://www.intersil.com or 407-727-9207 | Copyright © Intersil Corporation 1999
4-9
File Number
3103.1
CA3318
Functional Block Diagram
VAA+
ANALOG
SUPPLY
24
φ2 φ1
φ1
φ1
φ1
φ2
21
D
Q
D
Q
COUNT
256
D Q
CAB
# 256
LATCH
256
D
Q
D
Q
D Q
COUNT
193
ENCODER
LOGIC
ARRAY
BIT 7
LATCH
LATCH
BIT 6
D Q
R
D
Q
D
Q
COUNT
129
6
CLK
BIT 5
CAB
# 129
20
7
CLK
R
1/ REF
2
8
CLK
D Q
CAB
# 193
= 7Ω
9
BIT 8
(MSB)
LATCH
256
R
23
12
CLK
R = 2Ω
3/ REF
4
DIGITAL
SUPPLY
THREESTATE
OUTPUT
REGISTER DRIVERS OVERFLOW
VIN
VREF + 1
/2 R
22
VDD
φ1
D Q
= 30Ω
LATCH
R
5
CLK
LATCH
BIT 4
R
1/ REF
4
D
D
Q
D Q
= 4Ω
4
CLK
CAB
# 65
10
BIT 3
LATCH
R
VIN
Q
COUNT
65
D Q
LATCH
3
CLK
16
≅ 2K
R
VREF 15
D
CAB
(NOTE 1)
COMPARATOR #1
1/ R
2
Q
D
COUNT
1
BIT 2
D Q
2
CLK
LATCH
1
BIT 1
(LSB)
LATCH
11
D Q
≅ 50K
φ1 (AUTO BALANCE)
CLOCK
Q
1
CLK
18
PHASE
VAA17
CE1
φ2 (SAMPLE UNKNOWN)
19
14
ANALOG
GND
CE2
13
NOTE:
1. Cascaded Auto Balance (CAB).
VSS
DIGITAL
GND
4-10
11
CA3318
Absolute Maximum Ratings
Thermal Information
DC Supply Voltage Range (VDD or VAA+) . . . . . . . . . . -0.5V to +8V
(Referenced to VSS or VAA- Terminal, Whichever is More Negative)
Input Voltage Range
CE2 and CE1 . . . . . . . . . . . . . . . . . . . . VAA- -0.5V to VDD + 0.5V
Clock, Phase, VREF -, 1/2 Ref . . . . . . . VAA- -0.5V to VAA+ + 0.5V
Clock, Phase, VREF -, 1/4 Ref . . . . . . . . VSS- -0.5V to VDD + 0.5V
VIN , 3/4 REF, VREF + . . . . . . . . . . . . . . VAA- -0.5V to VAA- + 7.5V
Output Voltage Range, . . . . . . . . . . . . . . . VSS - 0.5V to VDD + 0.5V
Bits 1-8, Overflow (Outputs Off)
DC Input Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±20mA
Clock, Phase, CE1, CE2, VIN , Bits 1-8, Overflow
Thermal Resistance (Typical, Note 1)
θJA (oC/W) θJC (oC/W)
SBDIP Package . . . . . . . . . . . . . . . . . . . .
60
22
PDIP Package . . . . . . . . . . . . . . . . . . . . .
60
N/A
SOIC Package . . . . . . . . . . . . . . . . . . . . .
75
N/A
Maximum Junction Temperature
Ceramic Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175oC
Plastic Packages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150oC
Maximum Storage Temperature Range . . . . . . . . . .-65oC to 150oC
Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . . 265oC
(SOIC - Lead Tips Only)
Operating Conditions
Operating Voltage Range (VDD or VAA+) . . . 4V (Min) to 7.5V (Max)
Recommended VAA + Operating Range . . . . . . . . . . . . . . . VDD ±1V
Recommended VAA - Operating Range . . . . . . . . . . . . . . . VSS ±1V
Operating Temperature Range (TA) . . . . . . . . . . . . . . -40oC to 85oC
CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation
of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied.
NOTE:
1. θJA is measured with the component mounted on an evaluation PC board in free air.
Electrical Specifications
At 25oC, VAA+ = VDD = 5V, VREF + = 6.4V, VREF - = VAA- = VSS , CLK = 15MHz,
All Reference Points Adjusted, Unless Otherwise Specified
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
Resolution
8
-
-
Bits
Integral Linearity Error
-
-
± 1.5
LSB
SYSTEM PERFORMANCE
Differential Linearity Error
-
-
+1, -0.8
LSB
VIN = VREF- + 1/2 LSB
VIN = VREF+ - 1/2 LSB
-0.5
4.5
6.4
LSB
-1.5
0
1.5
LSB
Maximum Input Bandwidth
(Note 1) CA3318
2.5
5.0
-
MHz
Maximum Conversion Speed
CLK = Square Wave
15
17
-
MSPS
Signal to Noise Ratio (SNR)
fS = 15MHz, fIN = 100kHz
-
47
-
dB
fS = 15MHz, fIN = 4MHz
-
43
-
dB
Offset Error, Unadjusted
Gain Error Unadjusted
DYNAMIC CHARACTERISTICS
RMSSignal
= -------------------------------RMSNoise
Signal to Noise Ratio (SINAD)
RMSSignal
= -----------------------------------------------------------RMSNoise+Distortion
Total Harmonic Distortion, THD
fS = 15MHz, fIN = 100kHz
-
45
-
dB
fS = 15MHz, fIN = 4MHz
-
35
-
dB
fS = 15MHz, fIN = 100kHz
-
-46
-
dBc
fS = 15MHz, fIN = 4MHz
-
-36
-
dBc
fS = 15MHz, fIN = 100kHz
-
7.2
-
Bits
fS = 15MHz, fIN = 4MHz
-
5.5
-
Bits
Differential Gain Error
Unadjusted
-
2
-
%
Differential Phase Error
Unadjusted
-
1
-
%
Notes 2, 4
4
-
7
V
-
30
-
pF
VIN = 5V, VREF+ = 5V
-
-
3.5
mA
270
500
800
Ω
Effective Number of Bits (ENOB)
ANALOG INPUTS
Full Scale Range, VIN and (VREF+) - (VREF -)
Input Capacitance, VIN
Input Current, VIN , (See Text)
REFERENCE INPUTS
Ladder Impedance
4-11
CA3318
Electrical Specifications
At 25oC, VAA+ = VDD = 5V, VREF + = 6.4V, VREF - = VAA- = VSS , CLK = 15MHz,
All Reference Points Adjusted, Unless Otherwise Specified (Continued)
PARAMETER
TEST CONDITIONS
MIN
TYP
MAX
UNITS
DIGITAL INPUTS
Low Level Input Voltage, VOL
CE1, CE2
Phase, CLK
High Level Input Voltage, VIN
CE1, CE2
Phase, CLK
Input Leakage Current, II (Except CLK Input)
Note 4
-
-
0.2VDD
V
Note 4
-
-
0.2VAA
V
V
Note 4
0.7VDD
-
-
Note 4
0.7VAA
-
-
V
Note 3
-
±0.2
±5
µA
-
3
-
pF
Input Capacitance, CI
DIGITAL OUTPUTS
Output Low (Sink) Current
VO = 0.4V
4
10
-
mA
Output High (Source) Current
VO = 4.5V
-4
-6
-
mA
Three-State Output Off-State Leakage Current, IOZ
-
±0.2
±5
µA
Output Capacitance, CO
-
4
-
pF
33
-
∞
ns
25
-
500
ns
-
15
-
ns
TIMING CHARACTERISTICS
Auto Balance Time (φ1)
Sample Time (φ2)
Note 4
Aperture Delay
Aperture Jitter
-
100
-
ps
Data Valid Time, tD
Note 4
-
50
65
ns
Data Hold Time, tH
Note 4
25
40
-
ns
Output Enable Time, tEN
-
18
-
ns
Output Disable Time, tDIS
-
18
-
ns
POWER SUPPLY CHARACTERISTICS
Device Current (IDD + IA) (Excludes IREF)
Continuous Conversion (Note 4)
-
30
60
mA
Auto Balance (φ1)
-
30
60
mA
NOTES:
1. A full scale sine wave input of greater than fCLOCK/2 or the specified input bandwidth (whichever is less) may cause an erroneous code.
The -3dB bandwidth for frequency response purposes is greater than 30MHz.
2. VIN (Full Scale) or VREF+ should not exceed VAA+ + 1.5V for accuracy.
3. The clock input is a CMOS inverter with a 50kΩ feedback resistor and may be AC coupled with 1VP-P minimum source.
4. Parameter not tested, but guaranteed by design or characterization.
Timing Waveforms
DECODED DATA IS SHIFTED
TO OUTPUT REGISTERS
COMPARATOR DATA IS LATCHED
CLOCK (PIN 18)
IF PHASE (PIN 19)
IS LOW
CLOCK IF
PHASE IS HIGH
φ1
φ2
SAMPLE
N
AUTO
BALANCE
φ2
SAMPLE
N+1
φ1
φ2
AUTO
BALANCE
SAMPLE
N+2
tD
tH
DATA
N-2
DATA
N-1
FIGURE 1. INPUT TO OUTPUT TIMING DIAGRAM
4-12
DATA
N
CA3318
Timing Waveforms
(Continued)
CE1
CE2
tDIS
tEN
tDIS
BITS 1 - 8
DATA
tEN
DATA
DATA
HIGH
IMPEDANCE
OF
HIGH
IMPEDANCE
DATA
HIGH
IMPEDANCE
FIGURE 2. OUTPUT ENABLE TIMING DIAGRAM
AUTO
BALANCE
AUTO
BALANCE
SAMPLE
N
CLOCK
NO MAX
LIMIT
25ns
MIN
SAMPLE
N+1
33ns
MIN
25ns
MIN
50ns
MIN
DATA
FIGURE 3A. STANDBY IN INDEFINITE AUTO BALANCE (SHOWN WITH PHASE = LOW)
CLOCK
SAMPLE
N
500ns
MAX
AUTO
BALANCE
33ns
MIN
SAMPLE
N+1
25ns
MIN
AUTO
BALANCE
SAMPLE
N+2
50ns
TYP
DATA
N-1
DATA
FIGURE 3B. STANDBY IN SAMPLE (SHOWN WITH PHASE = LOW)
FIGURE 3. PULSE MODE OPERATION
4-13
DATA
N
CA3318
Typical Performance Curves
40
28
35
27
IDD (mA)
IDD (mA)
30
25
26
25
20
24
15
10
10
0
20
23
-50
30
-25
25
0
50
TEMPERATURE (oC)
fS (MHz)
FIGURE 4. DEVICE CURRENT vs SAMPLE FREQUENCY
1.00
fS = 15MHz, fI = 1MHz
7.8
fS = 15MHz
0.90
7.6
NON-LINEARITY (LSB)
0.80
7.4
ENOB (LSB)
100
FIGURE 5. DEVICE CURRENT vs TEMPERATURE
8.0
7.2
7.0
6.8
6.6
0.70
0.50
0.40
0.30
0.20
6.2
0.10
0
10
20
30
40
50
70
60
80
INL
0.60
6.4
6.0
-40 -30 -20 -10
DNL
0
-40 -30 -20 -10
90
0
TEMPERATURE (oC)
1.08
1.80
0.96
1.60
NON-LINEARITY (LSB)
1.00
0.84
INL
0.60
0.48
0.36
DNL
0.24
20
30
40
50
60
70
80
90
FIGURE 7. NON-LINEARITY vs TEMPERATURE
1.20
0.72
10
TEMPERATURE (oC)
FIGURE 6. ENOB vs TEMPERATURE
NON-LINEARITY (LSB)
75
fS = 15MHz
1.40
1.20
1.00
INL
0.80
0.60
DNL
0.40
0.20
0.12
0
0
0
5
10
15
20
0
25
fS (MHz)
FIGURE 8. NON-LINEARITY vs SAMPLE FREQUENCY
1
2
3
4
VREF (V)
5
6
FIGURE 9. NON-LINEARITY vs REFERENCE VOLTAGE
4-14
7
CA3318
Typical Performance Curves
(Continued)
8.0
7.6
fS = 15MHz
7.2
ENOB (LSB)
6.8
6.4
6.0
5.6
5.2
4.8
4.4
4.0
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
fI (MHz)
FIGURE 10. ENOB vs INPUT FREQUENCY
Pin Descriptions
CHIP ENABLE TRUTH TABLE
PIN
NAME
1
B1
Bit 1 (LSB)
DESCRIPTION
2
B2
Bit 2
3
B3
Bit 3
4
B4
Bit 4
5
B5
Bit 5
6
B6
Bit 6
7
B7
Bit 7
8
B8
Bit 8 (MSB)
Output Data Bits
(High = True)
Overflow
CE1
CE2
B1 - B8
OF
0
1
Valid
Valid
1
1
Three-State
Valid
X
0
Three-State
Three-State
X = Don’t Care
Theory of Operation
A sequential parallel technique is used by the CA3318
converter to obtain its high speed operation. The sequence
consists of the “Auto-Balance” phase, φ1, and the “Sample
Unknown” phase, φ2. (Refer to the circuit diagram.) Each
conversion takes one clock cycle (see Note). With the phase
control (pin 19) high, the “Auto-Balance” (φ1) occurs during
the high period of the clock cycle, and the “Sample Unknown”
(φ2) occurs during the low period of the clock cycle.
9
OF
10
1/ R
4
Reference Ladder 1/4 Point
11
VSS
Digital Ground
12
VDD
Digital Power Supply, +5V
13
CE2
Three-State Output Enable Input,
Active Low, See Truth Table.
NOTE: The device requires only a single phase clock The terminology
of φ1 and φ2 refers to the high and low periods of the same clock.
14
CE1
Three-State Output Enable Input
Active High. See Truth Table.
15
VREF -
Reference Voltage Negative Input
During the “Auto-Balance” phase, a transmission switch is
used to connect each of the first set of 256 commutating
capacitors to their associated ladder reference tap. Those
tap voltages will be as follows:
16
VIN
Analog Signal Input
17
VAA-
Analog Ground
18
CLK
Clock Input
19
PHASE
20
1/ R
2
21
VIN
Sample clock phase control input.
When PHASE is low, “Sample
Unknown” occurs when the clock is
low and “Auto Balance” occurs when
the clock is high (see text).
VTAP (N) = [(N/256) VREF] - (1/512) VREF]
= [(2N - 1)/512] VREF ,
Where:
VTAP (n) = reference ladder tap voltage at point n,
VREF
= voltage across VREF - to VREF +,
N
= tap number (1 through 256).
Reference Ladder Midpoint
Analog Signal Input
22
VREF+
23
3/ R
4
Reference Ladder 3/4 Point
24
VAA+
Analog Power Supply, +5V
Reference Voltage Positive Input
The other side of these capacitors are connected to singlestage amplifiers whose outputs are shorted to their inputs by
switches. This balances the amplifiers at their intrinsic trip
points, which is approximately (VAA+ - VAA-)/2. The first set
of capacitors now charges to their associated tap voltages.
4-15
CA3318
At the same time a second set of commutating capacitors
and amplifiers is also auto-balanced. The balancing of the
second-stage amplifier at its intrinsic trip point removes any
tracking differences between the first and second amplifier
stages. The cascaded auto-balance (CAB) technique, used
here, increases comparator sensitivity and temperature
tracking.
In the “Sample Unknown” phase, all ladder tap switches and
comparator shorting switches are opened. At the same time
VlN is switched to the first set of commutating capacitors.
Since the other end of the capacitors are now looking into an
effectively open circuit, any input voltage that differs from the
previous tap voltage will appear as a voltage shift at the
comparator amplifiers. All comparators that had tap voltages
greater than VlN will go to a “high” state at their outputs. All
comparators that had tap voltages lower than VlN will go to a
“low” state.
The status of all these comparator amplifiers is AC coupled
through the second-stage comparator and stored at the end
of this phase (φ2) by a latching amplifier stage. The latch
feeds a second latching stage, triggered at the end of φ1.
This delay allows comparators extra settling time. The status
of the comparators is decoded by a 256 to 9-bit decoder
array, and the results are clocked into a storage register at
the end of the next φ2.
A 3-stage buffer is used at the output of the 9 storage registers which are controlled by two chip-enable signals. CE1
will independently disable B1 through B6 when it is in a high
state. CE2 will independently disable B1 through B8 and the
OF buffers when it is in the low state.
To facilitate usage of this device, a phase control input is provided which can effectively complement the clock as it enters
the chip.
Continuous-Clock Operation
One complete conversion cycle can be traced through the
CA3318 via the following steps. (Refer to timing diagram.)
With the phase control in a “low” state, the rising edge of the
clock input will start a “sample” phase. During this entire
“high” state of the clock, the comparators will track the input
voltage and the first-stage latches will track the comparator
outputs. At the falling edge of the clock, all 256 comparator
outputs are captured by the 256 latches. This ends the “sample” phase and starts the “auto-balance” phase for the comparators. During this “low” state of the clock, the output of the
latches settles and is captured by a second row of latches
when the clock returns high. The second-stage latch output
propagates through the decode array, and a 9-bit code
appears at the D inputs of the output registers. On the next
falling edge of the clock, this 9-bit code is shifted into the output registers and appears with time delay tD as valid data at
the output of the three-state drivers. This also marks the end
of the next “sample” phase, thereby repeating the conversion
process for this next cycle.
Pulse-Mode Operation
The CA3318 needs two of the same polarity clock edges to
complete a conversion cycle: If, for instance, a negative
going clock edge ends sample “N”, then data “N” will appear
after the next negative going edge. Because of this requirement, and because there is a maximum sample time of
500ns (due to capacitor droop), most pulse or intermittent
sample applications will require double clock pulsing.
If an indefinite standby state is desired, standby should be in
auto-balance, and the operation would be as in Figure 3A.
If the standby state is known to last less than 500ns and
lowest average power is desired, then operation could be as
in Figure 3B.
Increased Accuracy
In most cases the accuracy of the CA3318 should be
sufficient without any adjustments. In applications where
accuracy is of utmost importance, five adjustments can be
made to obtain better accuracy, i.e., offset trim; gain trim;
and 1/4 , 1/2 and 3/4 point trim.
Offset Trim
In general, offset correction can be done in the preamp
circuitry by introducing a DC shift to VlN or by the offset trim
of the op amp. When this is not possible the VREF - input can
be adjusted to produce an offset trim. The theoretical input
voltage to produce the first transition is 1/2 LSB. The equation is as follows:
VlN (0 to 1 transition) = 1/2 LSB = 1/2 (VREF/256)
= VREF /512.
If VlN for the first transition is less than the theoretical, then a
single-turn 50Ω pot connected between VREF - and ground
will accomplish the adjustment. Set VlN to 1/2 LSB and trim
the pot until the 0-to-1 transition occurs.
If VlN for the first transition is greater than the theoretical,
then the 50Ω pot should be connected between VREF - and a
negative voltage of about 2 LSBs. The trim procedure is as
stated previously.
Gain Trim
In general, the gain trim can also be done in the preamp
circuitry by introducing a gain adjustment for the op amp.
When this is not possible, then a gain adjustment circuit
should be made to adjust the reference voltage. To perform
this trim, VlN should be set to the 255 to overflow transition.
That voltage is 1/3 LSB less than VREF + and is calculated as
follows:
VlN (255 to 256 transition) = VREF - VREF /512
= VREF(511/512).
To perform the gain trim, first do the offset trim and then
apply the required VlN for the 255 to overflow transition. Now
adjust VREF + until that transition occurs on the outputs.
4-16
CA3318
+10V TO 30V
INPUT
The first step for connecting a 9-bit circuit is to totem-pole
the ladder networks, as illustrated in Figure 13. Since the
absolute resistance value of each ladder may vary, external
trim of the mid-reference voltage may be required.
+
3
18Ω
2
1
(NOTE)
5K
IOT
6
4
VREF+
8
CA3085E
7
(PIN 22)
CW
10µF, TAN
(NOTE)
1.5K
+
4.7µF,
TAN/IOV
NOTE: Bypass VREF+ to analog GND near A/D with 0.1µF ceramic
cap. Parts noted should have low temperature drift.
FIGURE 11. TYPICAL VOLTAGE REFERENCE SOURCE FOR
DRIVING VREF+ INPUT
Grounding/Bypassing
1/ Point Trims
4
The 1/4 , 1/2 and 3/4 points on the reference ladder are
brought out for linearity adjusting or if the user wishes to
create a nonlinear transfer function. The 1/4 points can be
driven by the reference drivers shown (Figure 12) or by 2-K
pots connected between VREF + and VREF -. The 1/2 (mid-)
point should be set first by applying an input of 257/512 x
(VREF) and adjusting for an output changing from 128 to
129. Similarly the 1/4 and 3/4 points can be set with inputs of
129/512 and 385/512 x (VREF) and adjusting for counts of
192 to 193 and 64 to 65. (Note that the points are actually
1/ , 1/ and 3 / of full scale +1 LSB.)
4
2
4
VREF+
(PIN 22)
+10V TO +30V
510Ω
4
3
1K
IOT
CW
1K
IOT
CW
1K
IOT
2
11
+
5
+
6
3/ REF
4
(PIN 23)
7
10Ω
1/ REF
2
(PIN 20)
8
10Ω
1/ REF
4
The analog and digital supply grounds of a system should be
kept separate and only connected at the A/D. This keeps
digital ground noise out of the analog data to be converted.
Reference drivers, input amps, reference taps, and the VAA
supply should be bypassed at the A/D to the analog side of
the ground. See Figure 15 for a block diagram of this concept. All capacitors shown should be low impedance 0.1µF
ceramics and should be mounted as close to the A/D as possible. If VAA+ is derived from VDD , a small (10Ω resistor or
inductor and additional filtering (4.7µF tantalum) may be
used to keep digital noise out of the analog system.
Input Loading
The CA3318 outputs a current pulse to the VlN terminal at
the start of every sample period. This is due to capacitor
charging and switch feedthrough and varies with input voltage and sampling rate. The signal source must be capable
of recovering from the pulse before the end of the sample
period to guarantee a valid signal for the A/D to convert.
Suitable high speed amplifiers include the HA-5033,
HA-2542; and CA3450. Figure 16 is an example of an amplifier which recovers fast enough for sampling at 15MHz.
Output Loading
+
9
10Ω
-
10
CW
1
-
The overflow output of the lower device now becomes the
ninth bit. When it goes high, all counts must come from the
upper device. When it goes low, all counts must come from
the lower device. This is done simply by connecting the lower
overtlow signal to the CE1 control of the lower A/D converter
and the CE2 control of the upper A/D converter. The threestate outputs of the two devices (bits 1 through 8) are now
connected in parallel to complete the circuitry. The complete
circuit for a 9-bit A/D converter is shown in Figure 13.
-
The CMOS digital output stage, although capable of driving
large loads, will reflect these loads into the local ground. It is
recommended that a local QMOS buffer such as
CD74HC541 E be used to isolate capacitive loads.
(PIN 10)
510Ω
NOTES:
Definitions
1. All Op Amps = 3/4 CA324E.
2. Bypass all reference points to analog ground near A/D with 0.1µF
ceramic caps.
3. Adjust VREF+ first, then 1/3 , 3/4 and 1/4 points.
FIGURE 12. TYPICAL 1/4 POINT DRIVERS FOR ADJUSTING
LINEARITY (USE FOR MAXIMUM LINEARITY)
9-Bit Resolution
To obtain 9-bit resolution, two CA3318s can be wired together.
Necessary ingredients include an open-ended ladder network, an overflow indicator, three-state outputs, and chipenable controls - all of which are available on the CA3318.
Dynamic Performance Definitions
Fast Fourier Transform (FFT) techniques are used to evaluate
the dynamic performance of the converter. A low distortion sine
wave is applied to the input, it is sampled, and the output is
stored in RAM. The data is then transformed into the frequency
domain with a 4096 point FFT and analyzed to evaluate the
dynamic performance of the A/D. The sine wave input to the
part is -0.5dB down from fullscale for all these tests.
Signal-to-Noise (SNR)
SNR is the measured RMS signal to RMS noise at a
specified input and sampling frequency. The noise is the
RMS sum of all of the spectral components except the
fundamental and the first five harmonics.
4-17
CA3318
Signal-to-Noise + Distortion Ratio (SINAD)
Total Harmonic Distortion (THD)
SINAD is the measured RMS signal to RMS sum of all other
spectral components below the Nyquist frequency excluding DC.
THD is the ratio of the RMS sum of the first 5 harmonic
components to the RMS value of the measured input signal.
Effective Number of Bits (ENOB)
The effective number of bits (ENOB) is derived from the
SINAD data. ENOB is calculated from:
ENOB = (SINAD - 1.76 + VCORR)/6.02,
where:
VCORR = 0.5dB.
+6.4V REF
+5V
VREF+
OF
NC
VAA+
VDD
+5V
VAA-
BIT 8
VIN
BIT 1
VIN
CL
A
VIN1
0V TO 6.4V
PH
CE2
CE1
MID-POINT
DRIVER
6.4V REF
VREF-
VSS
D
A
VREF+
+5V
VDD
CE2
+5V
A
VIN
CE1
VIN
OF
BIT 9
BIT 8
BIT 8
BIT 1
BIT 1
CLOCK
VAA+
CL
VAAVREF -
PH
VSS
PHASE
D
A
FIGURE 13. USING TWO CA3318s FOR 9-BIT RESOLUTION
4-18
CA3318
4.7µF/10V TANTALUM
+
+5V (ANALOG SUPPLY)
A
+4V TO +6.5V
REFERENCE
OPTIONAL CAP
(SEE TEXT)
0.01µF
CLOCK
SOURCE
VAA+
BIT 1
3/ REF
4
BIT 2
VREF+
BIT 3
VIN
BIT 4
1/ REF
2
BIT 5
PHASE
BIT 6
CLK
BIT 7
VAA-
BIT 8
OVF
VIN
INPUT SIGNAL
1/ REF
4
VREF-
AMPLIFIER/BUFFER
(SEE TEXT)
A
D
DIGITAL
OUTPUT
CE1
VSS
CE2
VDD
A
D
+
CA3318
4.7µF
TANTALUM/10V
+5V (DIGITAL SUPPLY)
FIGURE 14. TYPICAL CIRCUIT CONFIGURATION FOR THE CA3318 WITH NO LINEARITY ADJUST
VIN
AMP
SIGNAL
SOURCE
REF
TO
DIGITAL
SYSTEM
OUTPUT
DRIVERS
VIN
VREF+
SIGNAL
GROUND
REFERENCE
TAPS
VDD
VAA+
VREF VAA-
-
VSS
SYSTEM
DIGITAL
GROUND
ANALOG +
SUPPLIES
VAA
SUPPLY
VDD
SUPPLY
FIGURE 15. TYPICAL SYSTEM GROUNDING/BYPASSING
+8V
75Ω
1VP-P
VIDEO
INPUT
10Ω
14
75Ω
0.001µF
7
11
5pF
8
13
3
16
A/D FLASH
INPUT
21
12
4
390
5
0.001µF
10Ω
NOTE: Ground-planing and tight layout
are extremely important.
10Ω
6
CA3450
9
750
0.1
-4V
110
0V TO -10V
OFFSET SOURCE
RS < 10Ω
FIGURE 16. TYPICAL HIGH BANDWIDTH AMPLIFIER FOR DRIVING THE CA3318
4-19
CA3318
TABLE 1. OUTPUT CODE TABLE
(NOTE 1)
INPUT VOLTAGE
BINARY OUTPUT CODE
CODE
DESCRIPTION
VREF
6.40V (V)
VREF
5.12V (V)
OF
MSB
B8
B7
B6
B5
B4
B3
B2
LSB
B1
DECIMAL
COUNT
Zero
0.00
0.00
0
0
0
0
0
0
0
0
0
0
1 LSB
0.025
0.02
0
0
0
0
0
0
0
0
1
1
2 LSB
0.05
0.04
0
0
0
0
0
0
0
1
0
2
•
•
•
•
•
•
•
•
•
1/ Full Scale
4
1.60
1.28
•
•
•
•
•
•
•
•
•
1/ Full Scale - 1 LSB
2
3.175
2.54
0
0
1
1
1
1
1
1
1
127
1/ Full Scale
2
3.20
2.56
0
1
0
0
0
0
0
0
0
128
1/ Full Scale + 1 LSB
2
3.225
2.58
0
1
0
0
0
0
0
0
1
129
•
•
•
•
•
•
•
•
•
3/ Full Scale
4
4.80
3.84
•
•
•
•
•
•
•
•
•
Full Scale - 1 LSB
6.35
5.08
0
1
1
1
1
1
1
1
0
254
Full Scale
6.375
5.10
0
1
1
1
1
1
1
1
1
255
Over Flow
6.40
5.12
1
1
1
1
1
1
1
1
1
511
•
•
•
0
0
1
0
0
•
•
•
0
0
0
0
•
•
•
•
•
•
•
•
•
0
1
1
0
0
64
•
•
•
0
0
0
0
•
•
•
192
•
•
•
NOTE: 1. The voltages listed above are the ideal centers of each output code shown as a function of its associated reference voltage.
Reducing Power
Clock Input
Most power is consumed while in the auto-balance state.
When operating at lower than 15MHz clock speed, power
can be reduced by stretching the sample (φ2) time. The constraints are a minimum balance time (φ1) of 33ns, and a
maximum sample time of 500ns. Longer sample times cause
droop in the auto-balance capacitors. Power can also be
reduced in the reference string by switching the reference on
only during auto-balance.
The Clock and Phase inputs feed buffers referenced to VAA+
and VAA-. Phase should be tied to one of these two potentials, while the clock (if DC coupled) should be driven at least
from 0.2 to 0.7 x (VAA+ - VAA-). The clock may also be AC
coupled with at least a 1VP-P swing. This allows TTL drive
levels or 5V QMOS levels when VAA+ is greater than 5V.
All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification.
Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design and/or specifications at any time without
notice. Accordingly, the reader is cautioned to verify that data sheets are current before placing orders. Information furnished by Intersil is believed to be accurate
and reliable. However, no responsibility is assumed by Intersil or its subsidiaries for its use; nor for any infringements of patents or other rights of third parties which
may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Intersil or its subsidiaries.
For information regarding Intersil Corporation and its products, see web site http://www.intersil.com
4-20
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