LINER LTC1779ES6 250ma current mode step-down dc/dc converter in thinsot Datasheet

LTC1779
250mA Current Mode
Step-Down DC/DC Converter
in ThinSOT
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FEATURES
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DESCRIPTIO
High Efficiency: Up to 94%
250mA Output Current
Wide VIN Range: 2.5V to 9.8V
550kHz Constant Frequency Operation
Burst ModeTM Operation at Light Load
Low Dropout: 100% Duty Cycle
0.8V Reference Allows Low Output Voltages
±2.5% Reference Accuracy
Current Mode Operation for Excellent Line and Load
Transient Response
Low Quiescent Current: 135µA
Shutdown Mode Draws Only 8µA Supply Current
Low Profile (1mm) ThinSOTTM Package
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APPLICATIO S
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The LTC1779 boasts a ±2.5% output voltage accuracy and
consumes only 135µA of quiescent current. For applications where efficiency is a prime consideration, the LTC1779
is configured for Burst Mode operation, which enhances
efficiency at low output current.
To further maximize the life of a battery source, the
internal P-channel MOSFET is turned on continuously in
dropout (100% duty cycle). In shutdown, the device draws
a mere 8µA. High constant operating frequency of 550kHz
allows the use of a small external inductor.
1- or 2-Cell Lithium-Ion-Powered Applications
Cellular Telephones
Wireless Modems
Portable Computers
Distributed 3.3V, 2.5V or 1.8V Power Systems
Scanners
The LTC1779 is available in a low profile (1mm) ThinSOT
package.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and ThinSOT are trademarks of Linear Technology Corporation.
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The LTC®1779 is a constant frequency current mode stepdown DC/DC converter in a 6-lead ThinSOT package. The
part operates with a 2.5V to 9.8V input and can provide up
to 250mA of output current. Current mode control provides excellent AC and DC load and line regulation. The
device incorporates an accurate undervoltage lockout feature that shuts down the LTC1779 when the input voltage
falls below 2V.
TYPICAL APPLICATIO
1
20k
2
SW
ITH/RUN
LTC1779
GND
VIN
+
VFB
SENSE –
D1
5
100pF
3
L1
22µH
6
4
R1
10Ω
C2
47µF
6V
100
90
169k
VOUT
2.5V
100mA
78.7k
1779 F01a
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: IR10BQ015
L1: COILTRONICS UP1B220
Figure 1. LTC1779 High Efficiency 2.5V/100mA Step-Down Converter
VIN = 3.3V
80
EFFICIENCY (%)
C3
0.1µF
Efficiency vs Load Current
VIN
2.5V
TO 9.8V
C1
10µF
16V
VIN = 6V
70
VIN = 9.8V
60
50
40
30
0.1
VOUT = 2.5V
RSENSE = 10Ω
1
10
100
LOAD CURRENT (mA)
1000
1779 F01b
1
LTC1779
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Note 1)
Input Supply Voltage (VIN).........................– 0.3V to 10V
SENSE –, SW Voltages .................. – 0.3V to (VIN + 0.3V)
VFB, ITH/RUN Voltages ..............................– 0.3V to 2.4V
SW Peak Output Current (< 10µs) .......................... 0.5A
Storage Ambient Temperature Range ... – 65°C to 150°C
Operating Temperature Range (Note 2) ...–40°C to 85°C
Junction Temperature (Note 3) ............................. 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
ITH/RUN 1
6 SW
GND 2
5 VIN
VFB 3
LTC1779ES6
4 SENSE
–
S6 PART MARKING
S6 PACKAGE
6-LEAD PLASTIC SOT-23
LTLP
TJMAX = 150°C, θJA = 230°C/ W
Consult LTC Marketing for parts specified with wider operating temperature
ranges.
ELECTRICAL CHARACTERISTICS
The ● denotes specifications that apply over the full operating temperature
range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)
PARAMETER
CONDITIONS
MIN
Input DC Supply Current
Normal Operation
Shutdown
UVLO
Typicals at VIN = 4.2V (Note 4)
2.5V ≤ VIN ≤ 9.8V
2.5V ≤ VIN ≤ 9.8V, VITH/RUN = 0V
VIN < UVLO Threshold
Undervoltage Lockout Threshold
VIN Falling
VIN Rising
●
Shutdown Threshold (at ITH/RUN)
Start-Up Current Source
VITH/RUN = 0V
Regulated Feedback Voltage
(Note 5) 0°C to 70°C
(Note 5) – 40°C to 85°C
Output Voltage Line Regulation
2.5V ≤ VIN ≤ 9.8V (Note 5)
Output Voltage Load Regulation
ITH/RUN Sinking 5µA (Note 5)
ITH/RUN Sourcing 5µA (Note 5)
VFB Input Current
(Note 5)
Overvoltage Protect Threshold
Measured at VFB
1.60
TYP
MAX
UNITS
135
8
7
240
22
13
µA
µA
µA
2.0
2.1
2.5
V
V
0.325
0.5
V
●
0.15
0.25
0.5
0.85
µA
●
●
0.780
0.770
0.800
0.800
0.820
0.830
V
V
0
3
–3
2.5
2.5
0.820
mV/V
mV/µA
mV/µA
5
25
0.860
0.895
nA
V
Overvoltage Protect Hysteresis
30
Overtemperature Protect Threshold
170
°C
Overtemperature Protect Hysteresis
15
°C
Oscillator Frequency
VFB = 0.8V
VFB = 0V
RDS(ON) of Internal P-Channel FET
Peak Current Sense Voltage
550
100
650
VIN = 4.2V, ISW = 100mA
0.85
1.4
(Note 6)
120
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1779E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the – 40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • θJ°C/W)
2
500
mV
kHz
kHz
Ω
mV
Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 5: The LTC1779 is tested in a feedback loop that servos VFB to the
output of the error amplifier.
Note 6: Peak current sense voltage is reduced dependent upon duty cycle
to a percentage of value as given in Figure 2.
LTC1779
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TYPICAL PERFOR A CE CHARACTERISTICS
Reference Voltage
vs Temperature
15
VIN = 4.2V
810
805
800
795
790
785
9
2.12
6
2.08
3
0
–3
–6
780
775
–55 –35 –15
2.16
1.88
5 25 45 65 85 105 125
TEMPERATURE (°C)
1.80
–55 –35 –15
1779 G03
RDS(ON) of Internal P-Channel FET
vs Input Voltage
1.85
VIN = 4.2V
RDS(ON) of Internal P-Channel FET
vs Temperature
1.85
ISW = 100mA
SENSE – = VIN
1.70
ISW = 100mA
1.70 SENSE – = V
IN
520
1.55
1.55
480
1.40
1.40
440
1.25
400
0.95
320
0.80
280
0.65
240
0.50
200
–55 –35 –15
1.25
TA = 125°C
1.10
360
VIN = 2.4V
0.95
TA = 25°C
0.65
0.50
3
4
5
7
8
6
INPUT VOLTAGE (V)
1779 G04
VIN = 6V
0.80
TA = –55°C
2
VIN = 4.2V
1.10
0.35
5 25 45 65 85 105 125
TEMPERATURE (°C)
5 25 45 65 85 105 125
TEMPERATURE (°C)
1779 G02
RDS(ON) (Ω)
ITH/RUN VOLTAGE (mV)
1.92
1.84
Shutdown Threshold
vs Temperature
560
1.96
–9
1779 G01
600
2.00
–12
–15
–55 –35 –15
5 25 45 65 85 105 125
TEMPERATURE (°C)
VIN FALLING
2.04
RDS(ON) (Ω)
VFB VOLTAGE (mV)
815
2.20
VIN = 4.2V
12
NORMALIZED FREQUENCY (%)
820
TRIP VOLTAGE (V)
825
Undervoltage Lockout Trip
Voltage vs Temperature
Normalized Oscillator Frequency
vs Temperature
9
10
1779 G05
0.35
–55 –35 –15
VIN = 9.8V
VIN = 8.4V
5 25 45 65 85 105 125
TEMPERATURE (°C)
1779 G06
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PI FU CTIO S
ITH/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run control input. The current comparator threshold
increases with this control voltage. Nominal voltage range
for this pin is 0.7V to 1.9V. Forcing this pin below 0.325V
causes the device to be shut down. In shutdown all
functions are disabled and the internal P-channel MOSFET
is turned off. The SW pin will be high impedance.
GND (Pin 2): Ground Pin.
VFB (Pin 3): Receives the feedback voltage from an external resistive divider across the output.
SENSE – (Pin 4): The Negative Input to the Current Comparator. Can be connected to VIN for default minimum
peak current of 250mA. Connecting a resistor between
SENSE – and VIN specifies a lower peak current. (See
Applications Information for specifying resistor value.)
VIN (Pin 5): Supply Pin. Must be closely decoupled to GND
Pin 2.
SW (Pin 6): Switching Node and Drain of Internal
P-Channel Power MOSFET. Connects to external inductor and catch diode.
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LTC1779
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FU CTIO AL DIAGRA
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SENSE –
VIN
4
5
+
OVERTEMP
DETECT
ICMP
–
2Ω
VIN
RS1
SLOPE
COMP
OSC
1×
SWITCHING
LOGIC AND
BLANKING
CIRCUIT
R
Q
S
24×
SW
6
–
FREQ
FOLDBACK
BURST
CMP
+
0.3V
+
SHORT-CIRCUIT
DETECT
SLEEP
–
0.15V
OVP
+
–
VREF
+
60mV
+
VREF
0.8V
VIN
EAMP
0.5µA
VFB
+
–
1 ITH/RUN
3
VIN
VIN
0.3V
–
0.325V
VOLTAGE
REFERENCE
+
SHDN
CMP
VREF
0.8V
–
GND
SHDN
UV
2
UNDERVOLTAGE
LOCKOUT
1.2V
1779FD
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OPERATIO
(Refer to Functional Diagram)
Main Control Loop
The LTC1779 is a constant frequency current mode switching regulator. During normal operation, the internal
P-channel power MOSFET is turned on each cycle when
the oscillator sets the RS latch (RS1) and turned off when
the current comparator (ICMP) resets the latch. The peak
inductor current at which ICMP resets the RS latch is
controlled by the voltage on the ITH/RUN pin, which is the
output of the error amplifier EAMP. An external resistive
divider connected between VOUT and ground allows the
EAMP to receive an output feedback voltage VFB. When the
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load current increases, it causes a slight decrease in VFB
relative to the 0.8V reference, which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.
The main control loop is shut down by pulling the ITH/RUN
pin low. Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 325mV, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
As the external compensation network continues to charge
LTC1779
(Refer to Functional Diagram)
up, the corresponding output current trip level follows,
allowing normal operation.
Comparator OVP guards against transient overshoots
> 7.5% by turning off the internal P-channel power
MOSFET and keeping it off until the fault is removed.
Burst Mode Operation
The LTC1779 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
set as if VITH/RUN = 1V (at low duty cycles) even though
the voltage at the ITH/RUN pin is at a lower value. If the
inductor’s average current is greater than the load requirement, the voltage at the ITH/RUN pin will drop. When the
ITH/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the internal MOSFET. The sleep signal
goes low when the ITH/RUN voltage goes above 0.925V
and the LTC1779 resumes normal operation. The next
oscillator cycle will turn the internal MOSFET on and the
switching cycle repeats.
Dropout Operation
When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means that
the internal P-channel MOSFET will remain on for more
than one oscillator cycle since the inductor current has not
ramped up to the threshold set by EAMP. Further reduction in input supply voltage will eventually cause the
P-channel MOSFET to be turned on 100%, i.e., DC. The
output voltage will then be determined by the input voltage
minus the voltage drop across the MOSFET, the sense
resistor and the inductor.
Undervoltage Lockout
To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorporated into the LTC1779. When the input supply voltage
drops below approximately 2.0V, the P-channel MOSFET
and all circuitry is turned off except the undervoltage
block, which draws only several microamperes.
Short-Circuit Protection
When the output is shorted to ground, the frequency of the
oscillator will be reduced to about 100kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s frequency will gradually increase to its designed rate when
the feedback voltage again approaches 0.8V.
Overvoltage Protection
As a further protection, the overvoltage comparator in the
LTC1779 will turn the internal MOSFET off when the
feedback voltage has risen 7.5% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 30mV.
Slope Compensation and Inductor’s Peak Current
The inductor’s peak current is determined by:
IPK =
M( VITH/RUN – 0.7)
10 RSENSE + 2Ω
(
)
when the LTC1779 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope compensation begins and effectively reduces the peak inductor current. The amount of reduction is given by the curves
in Figure 2.
110
100
90
SF = IOUT/IOUT(MAX) (%)
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OPERATIO
80
70
60
50
IRIPPLE = 0.4IPK
AT 5% DUTY CYCLE
IRIPPLE = 0.2IPK
AT 5% DUTY CYCLE
40
30
20
VIN = 4.2V
10
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1779 F02
Figure 2. Maximum Output Current vs Duty Cycle
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LTC1779
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OPERATIO
(Refer to Functional Diagram)
The variable M is the ratio of the total switch current to that
portion of the switch current that flows through RSENSE. M
is a function of both RSENSE and ROUT of the internal power
switch, which in turn, is a strong function of supply
voltage. For values of M refer to Figure 3. In order to
guarantee the desired IPK over the full range of supply
voltage, the minimum value of M, corresponding to the
minimum supply voltage seen in the application, should
be chosen. Note that the selection of RSENSE, and hence
the resulting M, is an iterative process. For most applications, a value of RSENSE between 0Ω and 20Ω will be
chosen.
60
RSENSE = 18.2Ω
55
RSENSE = 14Ω
M (mA/mA)
50
45
RSENSE = 10Ω
40
RSENSE = 6.2Ω
35
RSENSE = 2Ω
30
RSENSE = 0Ω
25
20
0
1
2
3 4 5 6 7 8
SUPPLY VOLTAGE (V)
9
10
1779 F03
Figure 3. M vs Supply Voltage
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APPLICATIO S I FOR ATIO
The basic LTC1779 application circuit is shown in Figure
1. External component selection is driven by the load
requirement and begins with the selection of L1 and
RSENSE (= R1). Next, the output diode D1 is selected
followed by CIN (= C1)and COUT(= C2).
Inductor Value Calculation
The inductance value has a direct effect on ripple current.
The ripple current, IRIPPLE, decreases with higher inductance or frequency and increases with higher VIN. The
inductor’s peak-to-peak ripple current is given by:
IRIPPLE =
( VIN – VOUT )VOUT
VIN • ƒ • L
where f is the operating frequency fixed at 550kHz in the
LTC1779.
A smaller value of L results in higher current ripple and
output voltage ripple as well as greater core losses. Larger
values of L decrease the ripple, but require finding physically larger inductors since maximum DC current rating
decreases significantly as inductance increases within
inductor product types. Generally, by choosing the desired ripple current based on the maximum output current, the inductor value can be calculated from the previous
equation. It is typical to choose the inductor so that the
ripple current is about 40% of the maximum output
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current at maximum input voltage. Use the following
equations to calculate L:
IRIPPLE = 0.4 • IOUT(MAX)
L=
(VIN(MAX) – VOUT ) • VOUT
VIN(MAX) • ƒ • IRIPPLE
IL(MAX) = IOUT(MAX) +
IRIPPLE
2
and then choose an appropriate L and recalculate the
ripple current.
In Burst Mode operation on the LTC1779, the ripple
current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peakto-peak ripple current must not exceed:
IRIPPLE ≤
M(0.030)
(RSENSE + 2Ω)
This implies a minimum inductance of:
LMIN =
 VOUT + VD 
 M(0.030)   VIN + VD 
f

 (RSENSE + 2Ω) 
VIN − VOUT
(Use VIN(MAX) = VIN)
LTC1779
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APPLICATIO S I FOR ATIO
A smaller value than L MIN could be used in the circuit;
however, the inductor current will not be continuous
during burst periods.
understand how it is going to work over the entire input
voltage range.
Inductor Core Selection
RSENSE Selection for Output Current
The selection of RSENSE determines the output current
limit, the maximum possible output current before the
internal current limit threshold is reached. IOUT(MAX), the
maximum specified output current in a design, must be
less than ICL. With the current comparator monitoring the
voltage developed across RSENSE, the threshold of the
comparator determines the inductor’s peak current. The
maximum output current, ICL, the LTC1779 can provide is
given by:
 SF   0.12V  IRIPPLE
ICL = M 

–
 100   RSENSE + 2Ω 
2
where SF and M are as defined in the previous section,
Figures 2 and 3. Typically, RSENSE is chosen between 0Ω
and 20Ω. Current limit is at a minimum at minimum input
voltage and maximum at maximum input voltage. Both
conditions should be considered in a design where current
limit is important.
To calculate several current limit conditions and choose
the best sense resistor for your design, first use minimum
input voltage. Calculate the duty cycle at minimum input
voltage.
DC =
VOUT
VIN(MIN)
Choose the slope factor, SF, from Figure 2 based on the
duty cycle. The ripple current calculated at minimum input
voltage and the chosen L should be used in the current
limit equation (see Inductor Value Calculation). Figure 3
provides several values of RSENSE and their corresponding
M values at different input voltages. Select the minimum
input voltage and calculate the resulting minimum current
limit settings.
The process must be repeated for maximum current limit
using duty cycle, slope factor, ripple current and mirror
ratio based on maximum input voltage in order to choose
the best sense resistor for a particular design and to
Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mu® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manufacturer is Kool Mu. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.
Output Diode Selection
The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
internal P-channel switch duty cycle. At high input voltages the diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of
the time. The most stressful condition for the diode is
when the output is short-circuited. Under this condition
the diode must safely handle IPK at close to 100% duty
cycle. Therefore, it is important to adequately specify the
diode peak current and average power dissipation so as
not to exceed the diode ratings.
Kool Mu is a registered trademark of Magnetics, Inc.
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LTC1779
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APPLICATIO S I FOR ATIO
Under normal load conditions, the average current conducted by the diode is:
V −V 
ID =  IN OUT  IOUT
 VIN + VD 
The allowable forward voltage drop in the diode is calculated from the maximum short-circuit current as:
VF ≈
PD
ICL (MAX)
where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.
A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.
CIN and COUT Selection
In continuous mode, the source current of the internal
P-channel MOSFET is a square wave of duty cycle
(VOUT + VD)/(VIN + VD). To prevent large voltage transients, a low ESR input capacitor sized for the maximum
RMS current must be used. The maximum RMS capacitor current is given by:
CIN Required IRMS ≈ IMAX
[VOUT (VIN − VOUT )]1/ 2
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT /2. This simple worst-case condition is commonly
used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC1779, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.
8
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering.
The output ripple (∆VOUT) is approximated by:

1 
∆VOUT ≈ IRIPPLE  ESR +


8 fCOUT 
where f is the operating frequency, COUT is the output
capacitance and IRIPPLE is the ripple current in the inductor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for COUT has been
met, the RMS current rating generally far exceeds the
IRIPPLE(P-P) requirement.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
Nichicon PL series and Panasonic SP.
Low Supply Operation
Although the LTC1779 can function down to approximately 2.0V, the maximum allowable output current is
reduced when VIN decreases below 3V. Figure 3 shows the
amount of change as the supply is reduced down to 2V.
Also shown in Figure 4 is the effect of VIN on VREF as VIN
goes below 2.3V.
LTC1779
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APPLICATIO S I FOR ATIO
Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1779 circuits: 1) LTC1779 DC bias current,
2) MOSFET gate charge current, 3) I2R losses and 4)
voltage drop of the output diode.
NORMALIZED VOLTAGE (%)
105
VREF
100
VITH
95
90
1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.
85
80
75
2.0
2.2
3.0
2.4
2.6
2.8
INPUT VOLTAGE (V)
1779 F04
Figure 4. Line Regulation of VREF and VITH
Setting Output Voltage
The LTC1779 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 5). By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:
 R2 
VOUT = 0.8 1 + 
 R1 
For most applications, an 80k resistor is suggested for R1.
To prevent stray pickup, locate resistors R1 and R2 close
to LTC1779.
VOUT
LTC1779
3
VFB
R2
R1
1779 F05
Figure 5. Setting Output Voltage
Efficiency Considerations
The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:
Efficiency = 100% – (η1 + η2 + η3 + ...)
where η1, η2, etc. are the individual losses as a percentage of input power.
2. MOSFET gate charge current results from switching
the gate capacitance of the internal power MOSFET.
Each time the MOSFET gate is switched from low to
high to low again, a packet of charge dQ moves from
VIN to ground. The resulting dQ/dt is a current out of
VIN which is typically much larger than the DC supply
current. In continuous mode, IGATECHG = f(Qp).
3. I2R losses are predicted from the DC resistances of the
internal MOSFET, inductor and current shunt. In continuous mode the average output current flows through
L but is “chopped” between the internal P-channel
MOSFET in series with RSENSE and the output diode.
The MOSFET RDS(ON) plus RSENSE multiplied by duty
cycle can be summed with the resistances of L and
RSENSE to obtain I2R losses.
4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the
load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of
0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.
5. Transition losses apply to the internal MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:
Transition Loss = 2(VIN)2IO(MAX)CRSS(f)
Other losses including CIN and COUT ESR dissipative
losses, and inductor core losses, generally account for
less than 2% total additional loss.
9
LTC1779
U
W
U U
APPLICATIO S I FOR ATIO
Foldback Current Limiting
PC Board Layout Checklist
As described in the Output Diode Selection, the worstcase dissipation occurs with a short-circuited output
when the diode conducts the current limit value almost
continuously. To prevent excessive heating in the diode,
foldback current limiting can be added to reduce the
current in proportion to the severity of the fault.
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1779. These items are illustrated graphically in the
layout diagram in Figure 7. Check the following in your
layout:
Foldback current limiting is implemented by adding diodes DFB1 and DFB2 between the output and the ITH/RUN
pin as shown in Figure 6. In a hard short (VOUT = 0V), the
current will be reduced to approximately 50% of the
maximum output current.
R2
ITH /RUN VFB
2. Is the input decoupling capacitor (0.1µF) connected
closely between VIN (Pin 5) and ground (Pin 2)?
3. Keep the switching node SW away from sensitive small
signal nodes.
VOUT
LTC1779
1. Large switch currents flow into the input capacitor CIN,
the power switch and the Schottky diode D1. The loop
formed by these components should be as small as
possible.
+
DFB1
4. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1 and R2 must be
connected between the (+) plate of COUT and signal
ground. Locate R1 and R2 close to the VFB pin.
R1
DFB2
1779 F06
Figure 6. Foldback Current Limiting
VIN
+
CIN
1
ITH/RUN
SW
L1
SW
6
VOUT
LTC1779
RITH
2
3
CITH
5
GND
VIN
VFB
4
SENSE –
+
RS
0.1µF
D1
COUT
R1
BOLD LINES INDICATE HIGH CURRENT PATHS
1779 F07
Figure 7. LTC1779 Layout Diagram (See PC Board Layout Checklist)
10
R2
LTC1779
U
PACKAGE DESCRIPTION
S6 Package
6-Lead Plastic SOT-23
(LTC DWG # 05-08-1634)
(LTC DWG # 05-08-1636)
2.80 – 3.10
(.110 – .118)
(NOTE 3)
SOT-23
(Original)
SOT-23
(ThinSOT)
A
.90 – 1.45
(.035 – .057)
1.00 MAX
(.039 MAX)
A1
.00 – 0.15
(.00 – .006)
.01 – .10
(.0004 – .004)
A2
.90 – 1.30
(.035 – .051)
.80 – .90
(.031 – .035)
L
.35 – .55
(.014 – .021)
.30 – .50 REF
(.012 – .019 REF)
2.60 – 3.00
(.102 – .118)
1.50 – 1.75
(.059 – .069)
(NOTE 3)
PIN ONE ID
.95
(.037)
REF
.25 – .50
(.010 – .020)
(6PLCS, NOTE 2)
.20
(.008)
A
DATUM ‘A’
L
NOTE:
1. CONTROLLING DIMENSION: MILLIMETERS
MILLIMETERS
2. DIMENSIONS ARE IN
(INCHES)
.09 – .20
(.004 – .008)
(NOTE 2)
A2
1.90
(.074)
REF
A1
S6 SOT-23 0401
3. DRAWING NOT TO SCALE
4. DIMENSIONS ARE INCLUSIVE OF PLATING
5. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
6. MOLD FLASH SHALL NOT EXCEED .254mm
7. PACKAGE EIAJ REFERENCE IS:
SC-74A (EIAJ) FOR ORIGINAL
JEDEL MO-193 FOR THIN
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
11
LTC1779
U
TYPICAL APPLICATIONS
Efficiency vs Load Current
LTC1779 Minimal Component Count,
Single Li-Ion to 1.8V/250mA Step-Down Converter
20k
2
SW
ITH/RUN
LTC1779
VIN
GND
6
+
D1
5
85
C2
47µF
6V
100k
SENSE –
VFB
VIN = 3.6V
80
VIN = 4.2V
75
70
65
100pF
3
VOUT
1.8V
250mA
VIN = 2.7V
90
EFFICIENCY (%)
1
95
VIN
2.7V
TO 4.2V
C1
10µF
16V
L1
10µH
100
80.6k
4
60
VOUT = 1.8V
RSENSE = 0Ω
55
1779 TA01a
C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: IR10BQ015
L1: COILTRONICS UP1B100
50
0.1
1
10
100
LOAD CURRENT (mA)
1k
1779 TA01b
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT 1375/LT1376
1.5A, 500kHz Step-Down Switching Regulators
High Frequency, Small Inductor, High Efficiency
LT1616
600mA Step-Down Switching Regulator
1.4MHz, 4V to 25V Input, ThinSOT Package
LTC1624
High Efficiency SO-8 N-Channel Switching Regulator Controller
8-Pin N-Channel Drive, 3.5V ≤ VIN ≤ 36V
®
TM
LTC1625
No RSENSE Synchronous Step-Down Regulator
High Efficiency, No Sense Resistor
LTC1627
Low Voltage, Monolithic Synchronous Step-Down Regulator
Low Supply Voltage Range: 2.65V to 8V, IOUT = 0.5A
LT1676/LT1776
Wide Input Range Step-Down Switching Regulators
60V Input, 700mA Internal Switches
LTC1735
Single, High Efficiency, Low Noise Synchronous Switching Controller High Efficiency 5V to 3.3V Conversion at up to 15A
LT1767
1.5A, 1.4MHz Step-Down DC/DC Converter
Higher Current, 8-Lead MSOP Package
LTC1771
Ultralow Supply Current Step-Down DC/DC Controller
10µA IQ, 93% Efficiency, 1.23V ≤ VOUT ≤ 18V,
2.8V ≤ VIN ≤ 20V
LTC1772
Constant Frequency Current Mode Step-Down DC/DC Controller
VIN = 2.5V to 9.8V, IOUT Up to 2A, ThinSOT Package
LTC1773
95% Efficient Synchronous Step-Down Controller
2.65V ≤ VIN ≤ 8.5V, 0.8V ≤ VOUT ≤ VIN, Current Mode, 550kHz
LTC1877
High Efficiency Monolithic Step-Down Regulator
550kHz, MS8, VIN Up to 10V, IQ = 10µA,
IOUT to 600mA at VIN = 5V
LTC1878
High Efficiency Monolithic Step-Down Regulator
550kHz, MS8, VIN Up to 6V, IQ = 10µA,
IOUT to 600mA at VIN = 3.3V
LTC3400
1.2MHz Synchronous Step-Up DC/DC Converter in ThinSOT
92% Efficiency, VIN = 0.5V to 6V, VOUT = 2.6V to 5V
LTC3401
Single Cell, High Current (1A), Micropower, Synchronous 3MHz
Step-Up DC/DC Converter
VIN = 0.5V to 5V, Up to 97% Efficiency Synchronizable
Oscillator from 100kHz to 3MHz
LTC3402
Single Cell, High Current (2A), Micropower, Synchronous 3MHz
Step-Up DC/DC Converter
VIN = 0.7V to 5V, Up to 95% Efficiency Synchronizable
Oscillator from 100kHz to 3MHz
LTC3404
1.4MHz High Efficiency, Monolithic Synchronous Step-Down
Regulator
Up to 95% Efficiency, 100% Duty Cycle, IQ = 10µA,
VIN = 2.65V to 6V
No RSENSE is a trademark of Linear Technology Corporation.
12
Linear Technology Corporation
1779f LT/TP 0701 2K • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2000
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