NSC LM34917A Ultra small 33v, 1.25a constant on-time buck switching regulator with intelligent current limit Datasheet

LM34917A
Ultra Small 33V, 1.25A Constant On-Time Buck Switching
Regulator with Intelligent Current Limit
General Description
The LM34917A Step-Down Switching Regulator features all
the functions needed to implement a low cost, efficient, buck
bias regulator capable of supplying at least 1.25A to the load.
To reduce excessive switch current due to the possibility of a
saturating inductor the valley current limit threshold changes
with input and output voltages, and the on-time is reduced
when current limit is detected. This buck regulator contains
an N-Channel Buck Switch, and is available in the 12 pin micro SMD package. The constant on-time feedback regulation
scheme requires no loop compensation, results in fast load
transient response, and simplifies circuit implementation. The
operating frequency remains constant with line and load variations due to the inverse relationship between the input voltage and the on-time. The valley current limit results in a
smooth transition from constant voltage to constant current
mode when current limit is detected, reducing the frequency
and output voltage, without the use of foldback. Additional
features include: VCC under-voltage lock-out, input over-voltage shutdown, thermal shutdown, gate drive under-voltage
lock-out, and maximum duty cycle limit.
Features
■ Functional Input Voltage Range: 8V to 33V
■ Micro SMD package
■ Input Over-Voltage Shutdown at ≊35V
■ Transient Capability to 50V
■ Integrated N-Channel buck switch
■ Valley current limit varies with VIN and VOUT to reduce
■
■
■
■
■
■
■
■
■
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excessive inductor current
On-time is reduced when in current limit
Integrated start-up regulator
No loop compensation required
Ultra-Fast transient response
Maximum switching frequency: 2 MHz
Operating frequency remains nearly constant with load
current and input voltage variations
Programmable soft-start
Precision internal reference
Adjustable output voltage
Thermal shutdown
Typical Applications
■ High Efficiency Point-Of-Load (POL) Regulator
■ Non-Isolated Buck Regulator
■ Secondary High Voltage Post Regulator
Package
■ micro SMD Package
Basic Step Down Regulator
20216601
© 2008 National Semiconductor Corporation
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LM34917A Ultra Small 33V, 1.25A Constant On-Time Buck Switching Regulator with Intelligent
Current Limit
May 8, 2008
LM34917A
Connection Diagrams
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20216630
Bump Side
Top View
Ordering Information
Order Number
Package Type
NSC Package
Drawing
Junction Temperature Range
Supplied As
LM34917ATL
12-Bump micro
SMD
TLA12UNA
−40°C to + 125°C
250 Units Tape and Reel
LM34917ATLX
3k Units Tape and Reel
Pin Descriptions
Pin Number
Name
Description
Application Information
A1
SGND
Sense Ground
Re-circulating current flows into this pin to the current sense
resistor.
A2
RTN
Circuit Ground
Ground for all internal circuitry other than the current limit
detection.
A3
FB
Feedback input from the regulated
output
Internally connected to the regulation and over-voltage
comparators. The regulation level is 2.5V.
B1
ISEN
Current sense
The re-circulating current flows out of this pin to the freewheeling diode.
B2
RON/SD
On-time control and shutdown
An external resistor from VIN to this pin sets the buck switch
on-time. Grounding this pin shuts down the regulator.
B3
SS
Softstart
An internal current source charges an external capacitor to
2.5V, providing the softstart function.
C1,C2
VIN
Input supply voltage
Operating input range is 8.0V to 33V, with over-voltage
C3
VCC
Output from the startup regulator
Nominally regulated at 7.0V. Connect a 0.1 µF capacitor
from this pin to RTN. An external voltage (8V to 14V) can be
applied to this pin to reduce internal dissipation. An internal
diode connects VCC to VIN.
D1,D2
SW
Switching Node
Internally connected to the buck switch source. Connect to
the inductor, diode, and bootstrap capacitor.
D3
BST
Boost pin for bootstrap capacitor
Connect a 0.022 µF capacitor from SW to this pin. The
capacitor is charged each off-time via an internal diode.
shutdown internally set at ≊35V. Transient capability is 50V.
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If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN to RTN
BST to RTN
SW to RTN (Steady State)
BST to VCC
VIN to SW
BST to SW
VCC to RTN
SGND to RTN
50V
64V
-1.5V
50V
50V
14V
14V
-0.3V to +0.3V
Operating Ratings
VIN Voltage
Junction Temperature
LM34917A
Current out of ISEN
SS to RTN
All Other Inputs to RTN
ESD Rating (Note 2)
Human Body Model
Storage Temperature Range
Junction Temperature
Absolute Maximum Ratings (Note 1)
See text
-0.3V to 4V
-0.3 to 7V
2kV
-65°C to +150°C
150°C
(Note 1)
8.0V to 33V
−40°C to + 125°C
Electrical Characteristics Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the
junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are guaranteed through test, design, or statistical
correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for reference purposes only.
Unless otherwise stated the following conditions apply: VIN = 12V, RON = 200kΩ. See (Note 4) and (Note 5).
Symbol
Parameter
Conditions
Min
Typ
Max
Units
6.6
7.0
7.4
V
Start-Up Regulator, VCC
VCCReg
VCC regulated output
Vin > 9V
VIN-VCC dropout voltage
ICC = 0 mA,
VCC = UVLOVCC + 250 mV
1.3
V
VCC output impedance
VIN = 8V
150
Ω
VIN = 12V
0.75
VCC current limit (Note 3)
VCC = 0V
11
mA
VCC under-voltage lockout
threshold
VCC increasing
5.45
V
UVLOVCC hysteresis
VCC decreasing
145
mV
UVLOVCC filter delay
100 mV overdrive
IIN operating current
Non-switching, FB = 3V
IIN shutdown current
(0 mA ≤ ICC ≤ 5 mA)
UVLOVCC
3
µs
0.68
0.95
mA
RON/SD = 0V
85
160
µA
0.33
0.7
Ω
4
4.62
Switch Characteristics
Rds(on)
Buck Switch Rds(on)
ITEST = 200 mA
UVLOGD
Gate Drive UVLO
VBST - VSW Increasing
2.65
V
UVLOGD hysteresis
450
mV
VSS
Pull-up voltage
2.5
V
ISS
Internal current source
11.6
µA
Restart threshold after OVP
shutdown
0.18
V
Softstart Pin
VRES
Current Limit
ILIM
Threshold
VIN = 8V, VFB = 2.4V
1.15
1.35
1.55
VIN = 30V, VFB = 2.4V
1.05
1.2
1.45
VIN = 30V, VFB = 1.0V
0.95
1.15
1.35
Response time
150
3
A
ns
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LM34917A
Symbol
Parameter
Conditions
Min
Typ
Max
Units
2.1
2.8
3.5
µs
On Timer
tON - 1
On-time (normal operation)
VIN = 10V, RON = 200 kΩ
tON - 2
On-time (normal operation)
VIN = 32V, RON = 200 kΩ
860
ns
tON - 3
On-time (current limit)
VIN = 10V, RON = 200 kΩ
1.13
µs
Shutdown threshold at RON/
SD
Voltage at RON/SD rising
Shutdown Threshold
hysteresis
Voltage at RON/SD falling
0.3
0.65
1.0
V
40
mV
90
ns
Off Timer
tOFF
Minimum Off-time
Regulation and Over-Voltage Comparators (FB Pin)
VREF
FB regulation threshold
SS pin = steady state
2.445
FB over-voltage threshold
FB bias current
FB = 3V
2.50
2.550
V
2.9
V
10
nA
Input Over-Voltage Shutdown
VIN(OV)
Shutdown voltage threshold at VIN increasing
VIN
33.0
34.8
36.9
V
Thermal Shutdown
TSD
Thermal shutdown temperature Junction temperature rising
175
°C
Thermal shutdown hysteresis
20
°C
58
°C/W
Thermal Resistance
θJA
Junction to Ambient
0 LFPM Air Flow
Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the
device is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.
Note 2: The human body model is a 100pF capacitor discharged through a 1.5kΩ resistor into each pin.
Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading
Note 4: For detailed information on soldering micro SMD packages, refer to AN-1112 available from National Semiconductor Corporation.
Note 5: Typical specifications represent the most likely parametric norm at 25°C operation.
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LM34917A
Typical Performance Characteristics
Unless otherwise specified the following conditions apply: TJ = 25°C
Efficiency at 1.5 MHz
Efficiency at 2 MHz
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VCC vs. VIN
ON-Time vs. VIN and RON
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Valley Current Limit Threshold vs. VFB and VIN
Voltage at the RON/SD Pin
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LM34917A
VCC vs. ICC
ICC vs Externally Applied VCC
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Shutdown and Operating Current Into VIN
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LM34917A
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Typical Application Circuit and Block Diagram
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LM34917A
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FIGURE 1. Startup Sequence
LM34917A can be applied in numerous applications to efficiently regulate down higher voltages. Additional features
include: Thermal shutdown, VCC under-voltage lock-out, gate
drive under-voltage lock-out, and maximum duty cycle limit.
Functional Description
The LM34917A Step Down Switching Regulator features all
the functions needed to implement a low cost, efficient buck
bias power converter capable of supplying at least 1.25A to
the load. This high voltage regulator contains an N-Channel
buck switch, is easy to implement, and is available in the micro
SMD package. The regulator’s operation is based on a constant on-time control scheme where the on-time is inversely
proportional to the input voltage. This feature results in the
operating frequency remaining relatively constant with load
and input voltage variations. The feedback control scheme
requires no loop compensation resulting in very fast load transient response. The valley current limit scheme protects
against excessively high currents if the output is short circuited when VIN is high. To aid in controlling excessive switch
current due to a possible saturating inductor the valley current
limit threshold changes with input and output voltages, and
the on-time is reduced by approximately 50% when current
limit is detected. An over-voltage detection at VIN stops the
circuit's switching when the input voltage exceeds 34.8V. The
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Control Circuit Overview
The LM34917A buck DC-DC regulator employs a control
scheme based on a comparator and a one-shot on-timer, with
the output voltage feedback (FB) compared to an internal reference (2.5V). If the FB voltage is below the reference the
buck switch is switched on for a time period determined by
the input voltage and a programming resistor (RON). Following
the on-time the switch remains off until the FB voltage falls
below the reference, but for a time not less than the minimum
off-time forced by the LM34917A. The buck switch is then
switched on for another on-time period.
When in regulation, the LM34917A operates in continuous
conduction mode at heavy load currents and discontinuous
conduction mode at light load currents. In continuous conduction mode the inductor’s current is always greater than
8
ON-Time Timer
The on-time for the LM34917A is determined by the RON resistor and the input voltage (VIN), calculated from:
(4)
The inverse relationship with VIN results in a nearly constant
frequency as VIN is varied. To set a specific continuous conduction mode switching frequency (fSW), the RON resistor is
determined from the following:
(1)
The buck switch duty cycle is equal to:
(2)
In discontinuous conduction mode, where the inductor’s current reaches zero during the off-time forcing a longer-thannormal off-time, the operating frequency is lower than in
continuous conduction mode, and varies with load current.
Conversion efficiency is maintained at light loads since the
switching losses reduce with the reduction in load and frequency. The approximate discontinuous operating frequency
can be calculated as follows:
(5)
Equations 1, 4 and 5 are valid only during normal operation i.e., the circuit is not in current limit. When the LM34917A
operates in current limit, the on-time is reduced by approximately 50%. This feature reduces the peak inductor current
which may be excessively high if the load current and the input
voltage are simultaneously high. This feature operates on a
cycle-by-cycle basis until the load current is reduced and the
output voltage resumes its normal regulated value. Equations
1, 4 and 5 have a ±25% tolerance.
(3)
Remote Shutdown
where RL = the load resistance, and L1 is the circuit’s inductor.
The output voltage is set by the two feedback resistors (R1,
R2 in the Block Diagram). The regulated output voltage is
calculated as follows:
The LM34917A can be remotely shut down by taking the
RON/SD pin below 0.65V. See Figure 2. In this mode the SS
pin is internally grounded, the on-timer is disabled, and bias
currents are reduced. Releasing the RON/SD pin allows the
circuit to resume operation after the SS pin voltage is below
0.18V. The voltage at the RON/SD pin is normally between
1.4V and 3.5V, depending on VIN and the RON resistor.
VOUT = 2.5 x (R1 + R2) / R2
Output voltage regulation is based on supplying ripple voltage
to the feedback input (FB pin) in phase with the SW pin. The
LM34917A requires a minimum of 25 mVp-p of ripple voltage
at the FB pin. The ripple is generated as a triangle wavefrom
at the junction of R3 and C8 as the SW pin switches high and
low, and fed to the FB pin by C7.
If the voltage at FB rises above 2.9V, due to a transient at
VOUT or excessive inductor current which creates higher than
normal ripple at VOUT, the internal over-voltage comparator
immediately shuts off the internal buck switch. The next ontime starts when the voltage at FB falls below 2.5V and the
inductor current falls below the current limits threshold.
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FIGURE 2. Remote Shutdown
Input Over-Voltage Shutdown
If the input voltage at VIN increases above 34.8V an internal
comparator disables the buck switch and the on-timer, and
grounds the soft-start pin. Normal operation resumes when
the VIN voltage reduces below 34.8V, and when the soft-start
voltage (at the SS pin) has reduced below 0.18V.
Current Limit
Current limit detection occurs during the off-time by monitoring the recirculating current flowing out of the ISEN pin.
Referring to the Block Diagram, during the off-time the inductor current flows through the load, into SGND, through the
internal sense resistor, out of ISEN and through D1 to the
inductor. If that current exceeds the current limit threshold the
current limit comparator output delays the start of the next on-
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LM34917A
zero, and the operating frequency remains relatively constant
with load and line variations. The minimum load current for
continuous conduction mode is one-half the inductor’s ripple
current amplitude. The approximate operating frequency is
calculated as follows:
LM34917A
time period. The next on-time starts when the current out of
ISEN is below the threshold and the voltage at FB falls below
2.5V. The operating frequency is typically lower due to longerthan-normal off-times.
The valley current limit threshold is a function of the input
voltage (VIN) and the output voltage sensed at FB, as shown
in the graph “Valley Current Limit Threshold vs. VFB and
VIN”. This feature reduces the inductor current’s peak value
at high line and load. To further reduce the inductor’s peak
current, the next cycle’s on-time is reduced by approximately
50% if the voltage at FB is below its threshold when the inductor current reduces to the current limit threshold (VOUT is
low due to current limiting).
Figure 3 illustrates the inductor current waveform during normal operation and in current limit. During the first “Normal
Operation” the load current is IOUT1, the average of the ripple
waveform. As the load resistance is reduced, the inductor
current increases until it exceeds the current limit threshold.
During the “Current Limited” portion of Figure 3, the current
limit threshold lowers since the high load current causes
VOUT (and the voltage at FB) to reduce. The on-time is reduced by approximately 50%, resulting in lower ripple amplitude for the inductor’s current. During this time the LM34917A
is in a constant current mode, with an average load current
equal to the current limit threshold + ΔI/2 (IOUT2). Normal operation resumes when the load current is reduced to IOUT3,
allowing VOUT, the current limit threshold, and the on-time to
return to their normal values. Note that in the second period
of “Normal Operation”, even though the inductor’s peak current exceeds the current limit threshold during part of each
cycle, the circuit is not in current limit since the current falls
below the threshold before the feedback voltage reduces to
its threshold to initiate the next on-time.
The peak current allowed through the buck switch, and the
ISEN pin, is 2A, and the maximum allowed average current
is 1.5A.
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FIGURE 3. Inductor Current - Normal and Current Limit Operation
N - Channel Buck Switch and Driver
Softstart
The LM34917A integrates an N-Channel buck switch and associated floating high voltage gate driver. The gate driver
circuit works in conjunction with an external bootstrap capacitor and an internal high voltage diode. A 0.022 µF capacitor
(C4) connected between BST and SW provides the voltage
to the driver during the on-time. During each off-time, the SW
pin is at approximately -1V, and C4 is recharged for the next
on-time from VCC through the internal diode. The minimum
off-time ensures a minimum time each cycle to recharge the
bootstrap capacitor.
The softstart feature allows the converter to gradually reach
a steady state operating point, thereby reducing start-up
stresses and current surges. Upon turn-on, after VCC reaches
the under-voltage threshold, an internal 11.6 µA current
source charges up the external capacitor at the SS pin to 2.5V
(t2 in Figure 1). The ramping voltage at SS (and the non-inverting input of the regulation comparator) ramps up the
output voltage in a controlled manner.
An internal switch grounds the SS pin if VCC is below the under-voltage lockout threshold, if the RON/SD pin is grounded,
or if VIN exceeds the overvoltage threshold.
Thermal Shutdown
The LM34917A should be operated so the junction temperature does not exceed 125°C. If the junction temperature increases above that, an internal Thermal Shutdown circuit
activates (typically) at 175°C, taking the controller to a low
power reset state by disabling the buck switch. This feature
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A standard value 15 µH inductor is selected. The maximum
ripple amplitude, which occurs at maximum VIN, calculates to
351 mA p-p, and the peak current is 1175 mA at maximum
load current. Ensure the selected inductor is rated for this
peak current.
C2: C2 should typically be no smaller than 3.3 µF, although
that is dependent on the frequency and the desired output
characteristics. C2 should be a low ESR good quality ceramic
capacitor. Experimentation is usually necessary to determine
the minimum value for C2, as the nature of the load may require a larger value. A load which creates significant transients requires a larger value for C2 than a non-varying load.
C1 and C5: C1’s purpose is to supply most of the switch current during the on-time, and limit the voltage ripple at VIN,
since it is assumed the voltage source feeding VIN has some
amount of source impedance.
At maximum load current, when the buck switch turns on, the
current into VIN suddenly increases to the lower peak of the
inductor’s ripple current, ramps up to the upper peak, then
drops to zero at turn-off. The average current during the ontime is the load current. For a worst case calculation, C1 must
supply this average load current during the maximum on-time,
without letting the voltage at VIN drop below ≊7.5V. The minimum value for C1 is calculated from:
Applications Information
EXTERNAL COMPONENTS
The procedure for calculating the external components is illustrated with the following design example. Referring to the
Block Diagram, the circuit is to be configured for the following
specifications:
- VOUT = 5V
- VIN = 8V to 33V
- Minimum load current = 200 mA
- Maximum load current = 1000 mA
- Switching Frequency = 1.5 MHz
- Soft-start time = 5 ms
- Output voltage ripple level: Minimum
R1 and R2: These resistors set the output voltage. The ratio
of the feedback resistors is calculated from:
R1/R2 = (VOUT/2.5V) - 1
For this example, R1/R2 = 1. R1 and R2 should be chosen
from standard value resistors in the range of 1.0 kΩ – 10 kΩ
which satisfy the above ratio. For this example, 2.49 kΩ is
chosen for R1 and R2.
RON: This resistor sets the on-time, and (by default) the
switching frequency. Since the maximum frequency is limited
by the minimum off-time forced by the LM34917A, first check
that the desired frequency is less than:
where tON is the maximum on-time, and ΔV is the allowable
ripple voltage at VIN (0.5V at VIN = 8V). C5’s purpose is to
minimize transients and ringing due to long lead inductance
leading the VIN pin. A low ESR 0.1 µF ceramic chip capacitor
must be located close to the VIN and RTN pins.
C3: The capacitor at the VCC pin provides noise filtering and
stability for the VCC regulator. C3 should be no smaller than
0.1 µF, and should be a good quality, low ESR ceramic capacitor. C3’s value, and the VCC current limit, determine a
portion of the turn-on-time (t1 in Figure 1).
C4: The recommended value for C4 is 0.022 µF. A high quality
ceramic capacitor with low ESR is recommended as C4 supplies a surge current to charge the buck switch gate at each
turn-on. A low ESR also helps ensure a complete recharge
during each off-time.
C6: The capacitor at the SS pin determines the soft-start time,
i.e. the time for the output voltage to reach its final value (t2 in
Figure 1). The capacitor value is determined from:
The RON resistor is calculated from equation 5 using the minimum input voltage:
Equation 4 is used to verify that this value resistor does not
set an on-time less than 120 ns at maximum input voltage. A
standard value 22.1 kΩ resistor is used, resulting in a nominal
frequency of 1.49 MHz. The minimum on-time is 188 ns at Vin
= 33V, and the maximum on-time is 510 ns at Vin = 8V.
L1: The main parameter affected by the inductor is the inductor current ripple amplitude (IOR). The minimum load current is used to determine the maximum allowable ripple in
order to maintain continuous conduction mode, where the
lower peak does not reach 0 mA. This is not a requirement of
the LM34917A, but serves as a guideline for selecting L1. For
this example, the maximum ripple current should be less than:
IOR(MAX) = 2 x IOUT(min) = 400 mAp-p
R3, C7, C8: The ripple amplitude at VOUT is determined by
C2’s characteristics and the inductor’s ripple current amplitude, and typically ranges from 5 mV to 30 mV over the Vin
range. Since the LM34917A’s regulation comparator requires
a minimum of 25 mVp-p ripple at the FB pin, these three components are added to generate and provide the necessary
ripple to FB in phase with the waveform at SW. R3 and C8
are chosen to generate a sawtooth waveform at their junction,
and that voltage is AC coupled to the FB pin via C7. To determine the values for R3, C7 and C8, the following procedure
is used:
(6)
For other applications, if the minimum load current is zero,
use 20% of IOUT(max) for IOUT(min) in equation 6. The ripple amplitude calculated in Equation 6 is then used in the following
equation:
Calculate VA = VOUT – (VSW x (1 – (VOUT/VIN(min)))
(7)
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LM34917A
helps prevent catastrophic failures from accidental device
overheating. When the junction temperature reduces below
155°C (typical hysteresis = 20°C), normal operation resumes.
LM34917A
where VSW is the absolute value of the voltage at the SW pin
during the off-time (typically 1V). VA, the DC voltage at the R3/
C8 junction, calculates to 4.63V, and is used in the next equation.
the SW pin may inadvertently affect the IC’s operation through
external or internal EMI. The diode must be rated for the maximum input voltage, the maximum load current, and the peak
current which occurs when the current limit and maximum
ripple current are reached simultaneously. The diode’s average power dissipation is calculated from:
PD1 = VF x IOUT x (1-D)
where VF is the diode’s forward voltage drop, and D is the ontime duty cycle.
where tON is the maximum on-time (at minimum input voltage), and ΔV is the desired ripple amplitude at the R3/C8
junction, typically 100 mV. R3 and C8 are chosen from standard value components to satisfy the above product. For this
example, 3300 pF is chosen for C8, and 5.23 kΩ is chosen
for R3. C7 is chosen large compared to C8, typically 0.1 µF.
D1: A Schottky diode is recommended. Ultra-fast recovery
diodes are not recommended as the high speed transitions at
FINAL CIRCUIT
The final circuit is shown in Figure 4, and its performance is
shown in Figure 5 and Figure 6. Current limit measured approximately 1.34A at Vin = 8V, and 1.27A at Vin = 33V. The
output ripple amplitude measured 4 mVp-p at Vin = 8V, and
14 mVp-p at Vin = 33V.
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FIGURE 4. Example Circuit
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FIGURE 5. Efficiency vs. Load Current and VIN (Circuit of Figure 4)
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LM34917A
20216634
FIGURE 6. Frequency vs. VIN (Circuit of Figure 4)
circuits of Figure 4 and Figure 7. Ripple is created at VOUT by
the inductor’s ripple current passing through R4. That ripple
voltage is coupled to the FB pin through the feedback resistors (R1, R2). Since the LM34917A requires a minimum of 25
mVp-p ripple at the FB pin, the ripple required at VOUT is higher than 25 mVp-p by the gain of the feedback resistors. The
minimum ripple current (IOR(min)) is calculated by re-arranging
Equation 7 using tON(max) and VIN(min). The minimum value for
R4 is calculated from:
ALTERNATE OUTPUT RIPPLE CONFIGURATIONS
For applications which can accept higher levels of ripple at
VOUT, the following configurations are simpler and a bit more
economical.
a) Alternate #1: In Figure 7 R3, C7 and C8 are removed, and
Cff and R4 are installed, resulting in a higher ripple level than
the circuit of Figure 4. Ripple is created at VOUT by the
inductor’s ripple current passing through R4. That ripple voltage is AC coupled to the FB pin through Cff, allowing the
minimum ripple at VOUT to be set at 25 mVp-p. The minimum
ripple current amplitude (IOR(min)) is calculated by re-arranging
Equation 7 using tON(max) and VIN(min). The minimum value for
R4 is calculated from:
The next larger standard value resistor should be used for R4.
The next larger standard value resistor should be selected for
R4 to allow for tolerances. The minimum value for Cff is determined from:
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The next larger standard value capacitor should be used for
Cff.
FIGURE 8. Maximum Ripple Configuration
c) Alternate minimum ripple configuration: The circuit in
Figure 9 is the same as that in Figure 8, except the output
voltage is taken from the junction of R4 and C2. The ripple at
VOUT is determined by the inductor’s ripple current and C2’s
characteristics. However, R4 slightly degrades the load regulation. This circuit may be suitable if the load current is fairly
constant. R4 is calculated as described in Alternate #2 above.
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FIGURE 7. Reduced Ripple Configuration
b) Alternate #2: In Figure 8, R3, C7 and C8 are removed,
and R4 is installed, resulting in a higher ripple level than the
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LM34917A
pact as possible, and all of the components must be as close
as possible to their associated pins. The two major current
loops have currents which switch very fast, and so the loops
should be as small as possible to minimize conducted and
radiated EMI. The first loop is that formed by C1, through the
VIN to SW pins, L1, C2, and back to C1.The second current
loop is formed by D1, L1, C2 and the SGND and ISEN pins.
The power dissipation within the LM34917A can be approximated by determining the total conversion loss (PIN - POUT),
and then subtracting the power losses in the free-wheeling
diode and the inductor. The power loss in the diode is approximately:
20216628
FIGURE 9. Alternate Minimum Output Ripple
Configuration
PD1 = Iout x VF x (1-D)
where Iout is the load current, VF is the diode’s forward voltage drop, and D is the on-time duty cycle. The power loss in
the inductor is approximately:
Minimum Load Current
The LM34917A requires a minimum load current of 1 mA. If
the load current falls below that level, the bootstrap capacitor
(C4) may discharge during the long off-time, and the circuit
will either shutdown, or cycle on and off at a low frequency. If
the load current is expected to drop below 1 mA in the application, R1 and R2 should be chosen low enough in value so
they provide the minimum required current at nominal VOUT.
PL1 = Iout2 x RL x 1.1
where RL is the inductor’s DC resistance, and the 1.1 factor
is an approximation for the AC losses. If it is expected that the
internal dissipation of the LM34917A will produce excessive
junction temperatures during normal operation, good use of
the PC board’s ground plane can help to dissipate heat. Additionally the use of wide PC board traces, where possible,
can help conduct heat away from the IC. Judicious positioning
of the PC board within the end product, along with the use of
any available air flow (forced or natural convection) can help
reduce the junction temperatures.
PC BOARD LAYOUT
Refer to application note AN-1112 for PC board guidelines for
the Micro SMD package.
The LM34917A regulation, over-voltage, and current limit
comparators are very fast, and respond to short duration
noise pulses. Layout considerations are therefore critical for
optimum performance. The layout must be as neat and com-
www.national.com
14
LM34917A
Physical Dimensions inches (millimeters) unless otherwise noted
Note: X1 = 1.971 mm, ±0.030 mm
X2 = 2.301 mm, ±0.030 mm
X3 = 0.60 mm, ±0.075 mm
12-Bump micro SMD Package
NS Package Number TLA12UNA
15
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LM34917A Ultra Small 33V, 1.25A Constant On-Time Buck Switching Regulator with Intelligent
Current Limit
Notes
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