Maxim MAX1846EUB High-efficiency, current-mode, inverting pwm controller Datasheet

19-2091; Rev 0; 8/01
High-Efficiency, Current-Mode,
Inverting PWM Controller
Features
♦ 90% Efficiency
The MAX1846 is available in an ultra-compact 10-pin
µMAX package. Operation at high frequency, compatibility with ceramic capacitors, and inverting topology
without transformers allow for a compact design.
Compatibility with electrolytic capacitors and flexibility
to operate down to 100kHz allow users to minimize the
cost of external components. The high-current output
drivers are designed to drive a P-channel MOSFET and
allow the converter to deliver up to 30W.
The MAX1847 features clock synchronization and shutdown functions. The MAX1847 can also be configured
to operate as an inverting flyback controller with an Nchannel MOSFET and a transformer to deliver up to
70W. The MAX1847 is available in a 16-pin QSOP
package.
Current-mode control simplifies compensation and provides good transient response. Accurate current-mode
control and over current protection are achieved
through low-side current sensing.
♦ Internal Soft-Start
♦ +3.0V to +16.5V Input Range
♦ -2V to -200V Output
♦ Drives High-Side P-Channel MOSFET
♦ 100kHz to 500kHz Switching Frequency
♦ Current-Mode, PWM Control
♦ Electrolytic or Ceramic Output Capacitor
♦ The MAX1847 also offers:
Synchronization to External Clock
Shutdown
N-Channel Inverting Flyback Option
Ordering Information
PART
TEMP. RANGE
PIN-PACKAGE
MAX1846EUB
-40°C to +85°C
10 µMAX
MAX1847EEE
-40°C to +85°C
16 QSOP
Typical Operating Circuit
POSITIVE
VIN
Applications
Cellular Base Stations
P
Networking Equipment
Optical Networking Equipment
VL
NEGATIVE
VOUT
IN
EXT
SLIC Supplies
MAX1846
MAX1847
CO DSL Line Driver Supplies
Industrial Power Supplies
COMP
CS
Automotive Electronic Power Supplies
Servers
FREQ
PGND
REF
GND
FB
Pin Configurations appear at end of data sheet.
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1846/MAX1847
General Description
MAX1846/MAX1847 high-efficiency PWM inverting controllers allow designers to implement compact, lownoise, negative-output DC-DC converters for telecom
and networking applications. Both devices operate
from +3V to +16.5V input and generate -2V to -200V
output. To minimize switching noise, both devices feature a current-mode, constant-frequency PWM control
scheme. The operating frequency can be set from
100kHz to 500kHz through a resistor.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
ABSOLUTE MAXIMUM RATINGS
IN, SHDN to GND ...................................................-0.3V to +20V
PGND to GND .......................................................-0.3V to +0.3V
VL to PGND for VIN ≤ 5.7V...........................-0.3V to (VIN + 0.3V)
VL to PGND for VIN > 5.7V .......................................-0.3V to +6V
EXT to PGND ...............................................-0.3V to (VIN + 0.3V)
REF, COMP to GND......................................-0.3V to (VL + 0.3V)
CS, FB, FREQ, POL, SYNC to GND .........................-0.3V to +6V
Continuous Power Dissipation (TA = +70°C)
10-Pin µMAX (derate 5.6mW/°C above +70°C) ...........444mW
16-Pin QSOP (derate 8.3mW/°C above +70°C)...........696mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = 0°C to +85°C, unless
otherwise noted.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
16.5
V
PWM CONTROLLER
Operating Input Voltage Range
UVLO Threshold
3.0
VIN rising
2.8
2.95
2.6
FB Threshold
No load
-12
0
12
mV
FB Input Current
VFB = -0.1V
-50
-6
50
nA
Load Regulation
CCOMP = 0.068µF, VOUT = -48V,
IOUT = 20mA to 200mA (Note 1)
-1
0
%
Line Regulation
CCOMP = 0.068µF, VOUT = -48V,
VIN = +8V to +16.5V, IOUT = 100mA
UVLO Hysteresis
60
Current-Limit Threshold
CS = GND
Supply Current
VFB = -0.1V, VIN = +3.0V to +16.5V
SHDN = GND, VIN = +3.0V to +16.5V
mV
0.04
85
CS Input Current
Shutdown Supply Current
2.74
V
VIN falling
100
%
115
mV
10
20
µA
0.75
1.2
mA
10
25
µA
1.25
1.264
V
-2
-15
mV
4.25
4.65
V
-20
-60
mV
REFERENCE AND VL REGULATOR
REF Output Voltage
IREF = 50µA
REF Load Regulation
IREF = 0 to 500µA
VL Output Voltage
IVL = 100µA
VL Load Regulation
IVL = 0.1mA to 2.0mA
2
1.236
3.85
_______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = 0°C to +85°C, unless
otherwise noted.)
OSCILLATOR
Oscillator Frequency
RFREQ = 500kΩ ±1%
90
100
110
RFREQ = 147kΩ ±1%
255
300
345
RFREQ = 76.8kΩ ±1%
Maximum Duty Cycle
500
RFREQ = 500kΩ ±1%
93
96
97
RFREQ = 147kΩ ±1%
85
88
90
%
93
%
200
ns
200
ns
550
kHz
RFREQ = 76.8kz ±1%
SYNC Input Signal Duty-Cycle
Range
80
7
Minimum SYNC Input Logic Low
Pulse Width
SYNC Input Rise/Fall Time
kHz
50
(Note 2)
SYNC Input Frequency Range
100
DIGITAL INPUTS
POL, SYNC, SHDN Input High
Voltage
2.0
V
POL, SYNC, SHDN Input Low
Voltage
POL, SYNC Input Current
POL, SYNC = GND or VL
VSHDN = +5V or GND
SHDN Input Current
VSHDN = +16.5V
-12
0.45
V
20
40
µA
-4
0
1.5
6
µA
SOFT-START
Soft-Start Clock Cycles
Cycles
1024
Soft-Start Levels
64
EXT OUTPUT
EXT Sink/Source Current
EXT On-Resistance
VIN = +5V, VEXT forced to +2.5V
1
EXT high or low, tested with 100mA load, VIN = +5V
2
5
A
EXT high or low, tested with 100mA load, VIN = +3V
5
10
Ω
Note 1: Production test correlates to operating conditions.
Note 2: Guaranteed by design and characterization.
_______________________________________________________________________________________
3
MAX1846/MAX1847
ELECTRICAL CHARACTERISTICS (continued)
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
ELECTRICAL CHARACTERISTICS
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = -40°C to +85°C,
unless otherwise noted.) (Note 3)
PARAMETER
CONDITIONS
MIN
MAX
UNITS
3.0
16.5
V
PWM CONTROLLER
Operating Input Voltage Range
UVLO Threshold
VIN rising
2.95
V
VIN falling
2.6
FB Threshold
No load
-20
20
mV
FB Input Current
VFB = -0.1V
-50
50
nA
Load Regulation
CCOMP = 0.068µF, VOUT = -48V,
IOUT= 20mA to 200mA (Note 1)
-2
0
%
85
115
mV
Current Limit Threshold
CS Input Current
CS = GND
20
µA
Supply Current
VFB = -0.1V, VIN = +3.0V to +16.5V
1.2
mA
Shutdown Supply Current
SHDN = GND, VIN = +3.0V to +16.5V
25
µA
REFERENCE AND VL REGULATOR
REF Output Voltage
IREF = 50µA
REF Load Regulation
IREF = 0 to 500µA
VL Output Voltage
IVL = 100µA
VL Load Regulation
IVL = 0.1mA to 2.0mA
1.225
3.85
1.275
V
-15
mV
4.65
V
-60
mV
OSCILLATOR
Oscillator Frequency
Maximum Duty Cycle
RFREQ = 500kΩ ±1%
84
116
RFREQ = 147kΩ ±1%
255
345
RFREQ = 500kΩ ±1%
93
98
RFREQ = 147kΩ ±1%
84
93
7
93
%
200
ns
200
ns
550
kHz
SYNC Input Signal Duty-Cycle
Range
Minimum SYNC Input Logic Low
Pulse Width
SYNC Input Rise/Fall Time
SYNC Input Frequency Range
(Note 2)
100
kHz
%
DIGITAL INPUTS
POL, SYNC, SHDN Input High
Voltage
2.0
POL, SYNC, SHDN Input Low
Voltage
4
_______________________________________________________________________________________
V
0.45
V
High-Efficiency, Current-Mode,
Inverting PWM Controller
(V SHDN = VIN = +12V, SYNC = GND, PGND = GND, RFREQ = 147kΩ ±1%, CVL = 0.47µF, CREF = 0.1µF, TA = -40°C to +85°C,
unless otherwise noted.) (Note 3)
PARAMETER
CONDITIONS
POL, SYNC Input Current
MIN
MAX
UNITS
40
µA
POL, SYNC = GND or VL
V SHDN = +5V or GND
SHDN Input Current
-12
0
V SHDN = +16.5V
µA
6
EXT OUTPUT
EXT On-Resistance
EXT high or low, IEXT = 100mA, VIN = +5V
7.5
EXT high or low, IEXT = 100mA, VIN = +3V
12
Ω
Note 3: Parameters to -40°C are guaranteed by design and characterization.
Typical Operating Characteristics
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1µF, TA = +25°C, unless otherwise
noted.)
VIN = 16.5V
50
40
70
VIN = 3.3V
60
VIN = 3V
50
40
50
40
30
30
20
20
10
10
APPLICATION CIRCUIT A
1
10
100
VOUT = -5V
1000
VOUT = -12V
APPLICATION CIRCUIT B
0
10,000
VIN = 16.5V
60
20
0
VIN = 12V
70
30
10
1
10
LOAD CURRENT (mA)
100
1000
1
10,000
10
-11.94
1.6
1.4
1.2
-11.96
-12.02
1.258
VREF (V)
IIN (mA)
-12.00
1.262
1.254
1.0
-11.98
1000
REFERENCE VOLTAGE
vs. TEMPERATURE
MAX1846/7 toc05
-11.92
100
LOAD CURRENT (mA)
SUPPLY CURRENT
vs. SUPPLY VOLTAGE
MAX1846/7 toc04
-11.90
VOUT = -48V
APPLICATION CIRCUIT C
0
LOAD CURRENT (mA)
OUTPUT VOLTAGE LOAD REGULATION
OUTPUT VOLTAGE (V)
80
MAX1846/7 toc06
60
80
90
EFFICIENCY (%)
70
VIN = 5V
90
EFFICIENCY (%)
EFFICIENCY (%)
VIN = 5V
100
MAX1846/7 toc03
MAX1846/7 toc01
90
80
EFFICIENCY vs. LOAD CURRENT
EFFICIENCY vs. LOAD CURRENT
100
MAX1846/7 toc02
EFFICIENCY vs. LOAD CURRENT
100
0.8
0.6
1.250
1.246
-12.04
0.4
-12.06
1.242
0.2
-12.08
APPLICATION CIRCUIT B
-12.10
0
100
200
300
VFB = -0.1V
VIN = 5V
400
LOAD CURRENT (mA)
1.238
0
500
600
0
2
4
6
8
VIN (V)
10
12
14
16
-40
-20
0
20
40
60
80
100
TEMPERATURE (°C)
_______________________________________________________________________________________
5
MAX1846/MAX1847
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1µF, TA = +25°C, unless otherwise
noted.)
VL VOLTAGE
vs. TEMPERATURE
VL LOAD REGULATION
4.340
4.300
1.255
4.27
4.26
VL (V)
VL (V)
VREF (V)
4.260
1.250
4.220
MAX1846/7 toc09
MAX1846/7 toc07
1.260
MAX1846/7 toc08
REFERENCE LOAD REGULATION
4.25
4.24
4.180
1.245
4.23
4.140
IVL = 0
4.100
1.240
100
200
300
400
500
0
20
40
60
80
0
100
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
TEMPERATURE (°C)
IVL (mA)
SHUTDOWN SUPPLY CURRENT
vs. TEMPERATURE
OPERATING CURRENT
vs. TEMPERATURE
SWITCHING FREQUENCY
vs. RFREQ
10
8
VIN = 3V
6
4
500
12
MAX1846/7 toc12
A
A: VIN = 3V, VOUT = -12V
400
10
fOSC (kHz)
VIN = 10V
VIN = 16.5V
14
MAX1846/7 toc11
14
MAX1846/7 toc10
8
APPLICATION CIRCUIT A
B: VIN = 5V, VOUT = -5V
C: VIN = 16.5V, VOUT = -5V
6
B
300
200
4
100
2
2
0
C
0
-40
-20
0
20
60
40
TEMPERATURE (°C)
80
100
0
-40
-20
0
20
40
60
80
100
0
TEMPERATURE (°C)
SWITCHING FREQUENCY
vs. TEMPERATURE
EXT RISE/FALL TIME
vs. CAPACITANCE
301
100
TIME (ns)
120
299
298
300
400
FALL TIME
60
296
40
295
20
600
5V/div
SHDN
0
VOUT
80
297
500
MAX1846/7 toc15
140
300
200
EXITING SHUTDOWN
160
MAX1846/7 toc13
302
100
RFREQ (kΩ)
MAX1846/7 toc14
SHUTDOWN SUPPLY CURRENT (µA)
-20
IREF (µA)
16
12
4.22
-40
OPERATING CURRENT (mA)
0
FREQUENCY (kHz)
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
5V/div
RISE TIME
RFREQ = 147kΩ ±1%
294
0
-40
6
VIN = 12V
-20
0
20
60
40
TEMPERATURE (°C)
80
100
0
2000
4000
6000
CAPACITANCE (pF)
8000
1A/div
IL
10,000
APPLICATION CIRCUIT B
1ms/div
_______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
HEAVY-LOAD SWITCHING
WAVEFORM
ENTERING SHUTDOWN
MAX1846/7 toc16
MAX1846/7 toc17
SHDN
VOUT
5V/div
0
100mV/div
5V/div
1A/div
IL
VOUT
LX
1A/div
IL
10V/div
APPLICATION CIRCUIT B
1ms/div
APPLICATION CIRCUIT B
1µs/div
ILOAD = 600mA
LIGHT-LOAD SWITCHING
WAVEFORM
MAX1846/7 toc18
VOUT
100mV/div
1A/div
IL
LX
10V/div
APPLICATION CIRCUIT B
1µs/div
ILOAD = 50mA
LOAD-TRANSIENT RESPONSE
LOAD-TRANSIENT RESPONSE
MAX1846/7 toc19
MAX1846/7 toc20
ILOAD
ILOAD
VOUT
VOUT
200mV/div
IL
500mA/div
500mV/div
IL
1A/div
APPLICATION CIRCUIT B
2ms/div
ILOAD = 10mA to 400mA
APPLICATION CIRCUIT C
400µs/div
ILOAD = 4mA to 100mA
_______________________________________________________________________________________
7
MAX1846/MAX1847
Typical Operating Characteristics (continued)
(Circuit references are from Table 1 in the Main Application Circuits section, CVL = 0.47µF, CREF = 0.1µF, TA = +25°C, unless otherwise
noted.)
High-Efficiency, Current-Mode,
Inverting PWM Controller
MAX1846/MAX1847
Pin Description
PIN
MAX1847
—
1
POL
1
2
VL
FUNCTION
Sets polarity of the EXT pin. Connect POL to GND to set EXT for use with an external
PMOS high-side FET. Connect POL to VL to set EXT for use with an external NMOS lowside FET in transformer-based applications.
VL Low-Dropout Regulator. Connect 0.47µF ceramic capacitor from VL to GND.
2
3
FREQ
Oscillator Frequency Set Input. Connect a resistor (RFREQ) from FREQ to GND to set the
internal oscillator frequency from 100kHz (RFREQ = 500kΩ) to 500kHz (RFREQ = 76.8kΩ).
RFREQ is still required if an external clock is used at SYNC. See Setting the Operating
Frequency section.
3
4
COMP
Compensation Node for Error Amp/Integrator. Connect a series resistor/capacitor network
from COMP to GND for loop compensation. See Design Procedure.
4
5
REF
5
6
FB
—
7,9
N.C.
—
8
SHDN
Shutdown Control. Drive SHDN low to turn off the DC-DC controller. Drive high or connect
to IN for normal operation.
6
10,11
GND
Analog Ground. Connect to PGND.
7
12
PGND
8
13
CS
Negative Rail for EXT Driver and Negative Current-Sense Input. Connect to GND.
Positive Current-Sense Input. Connect a current-sense resistor (RCS) between CS and
9
14
EXT
External MOSFET Gate-Driver Output. EXT swings from IN to PGND.
10
15
IN
—
8
NAME
MAX1846
16
SYNC
1.25V Reference Output. REF can source up to 500µA. Bypass with a 0.1µF ceramic
capacitor from REF to GND.
Feedback Input. Connect FB to the center of a resistor-divider connected between the
output and REF. The FB threshold is 0.
No Connection
Power-Supply Input
Operating Frequency Synchronization Control. Drive SYNC low or connect to GND to set
the internal oscillator frequency with RFREQ. Drive SYNC with a logic-level clock input
signal to externally set the converter’s operating frequency. DC-DC conversion cycles
initiate on the rising edge of the input clock signal. Note that when driving SYNC with an
external signal, RFREQ must still be connected to FREQ.
_______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
3 x 22µF
10V
VIN
+3V to +5.5V
22kΩ
FDS6375
CMSH5-40
2
0.47µF
IN
VL
8
16
SHDN
EXT
SYNC
CS
MAX1847
4
N.C.
COMP
PGND
10kΩ
3
0.22µF
5
150kΩ
VOUT
-12V AT 400mA
15
47µF
16V
10µH
DO5022P-103
14
47µF
16V
13
Sanyo
16TPB47M
7, 9
0.02Ω
1W
12
R1
95.3kΩ
1%
FREQ
REF
FB
POL
1
GND
10, 11
6
R2
10.0kΩ
1%
1200pF
0.1µF
_______________________________________________________________________________________
9
MAX1846/MAX1847
Typical Application Circuit
High-Efficiency, Current-Mode,
Inverting PWM Controller
MAX1846/MAX1847
Functional Diagram
IN
EXT
SHDN
MAX1847 ONLY
STARTUP
CIRCUITRY
PGND
EXT DRIVER
VL
VL
REGULATOR
UNDERVOLTAGE
LOCK OUT
POL
SYNC
MAX1847 ONLY
MAX1846
MAX1847
CONTROL
CIRCUITRY
OSCILLATOR
FREQ
ERROR
COMPARATOR
COMP
CS
FB
GM
CURRENTSENSE
AMPLIFIER
ERROR
AMPLIFIER
SOFT-START
PGND
SLOPE
COMP
REF
REFERENCE
GND
10
______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
PWM Controller
The architecture of the MAX1846/MAX1847 currentmode PWM controller is a Bi-CMOS multi-input system
that simultaneously processes the output-error signal,
the current-sense signal, and a slope-compensation
ramp (Functional Diagram). Slope compensation prevents subharmonic oscillation, a potential result in current-mode regulators operating at greater than 50%
duty cycle. The controller uses fixed-frequency, current-mode operation where the duty ratio is set by the
input-to-output voltage ratio. The current-mode feedback loop regulates peak inductor current as a function
of the output error signal.
Internal Regulator
The MAX1846/MAX1847 incorporate an internal lowdropout regulator (LDO). This LDO has a 4.25V output
and powers all MAX1846/MAX1847 functions (excluding EXT) for the primary purpose of stabilizing the performance of the IC over a wide input voltage range
(+3V to +16.5V). The input to this regulator is connected to IN, and the dropout voltage is typically 100mV, so
that when VIN is less than 4.35V, VL is typically VIN
minus 100mV. When the LDO is in dropout, the
MAX1846/MAX1847 still operate with VIN as low as 3V.
For best performance, it is recommended to connect
VL to IN when the input supply is less than 4.5V.
Undervoltage Lockout
The MAX1846/MAX1847 have an undervoltage lockout
circuit that monitors the voltage at VL. If VL falls below
the UVLO threshold (2.8V typ), the control logic turns the
P-channel FET off (EXT high impedance). The rest of the
IC circuitry is still powered and operating. When VL
increases to 60mV above the UVLO threshold, the IC
resumes operation from a start up condition (soft-start).
Soft-Start
The MAX1846/MAX1847 feature a “digital” soft-start
that is preset and requires no external capacitor. Upon
startup, the FB threshold decrements from the reference voltage to 0 in 64 steps over 1024 cycles of fOSC
or fSYNC. See the Typical Operating Characteristics for
a scope picture of the soft-start operation. Soft-start is
implemented: 1) when power is first applied to the IC,
2) when exiting shutdown with power already applied,
and 3) when exiting undervoltage lockout.
Shutdown (MAX1847 only)
The MAX1847 shuts down to reduce the supply current
to 10µA when SHDN is low. In this mode, the internal reference, error amplifier, comparators, and biasing circuitry turn off. The EXT output becomes high impedance
and the external pullup resistor connected to EXT pulls
VEXT to VIN, turning off the P-channel MOSFET. When in
shutdown mode, the converter’s output goes to 0.
Frequency Synchronization
(MAX1847 only)
The MAX1847 is capable of synchronizing its switching
frequency with an external clock source. Drive SYNC
with a logic-level clock input signal to synchronize the
MAX1847. A switching cycle starts on the rising edge
of the signal applied to SYNC. Note that the frequency
of the signal applied to SYNC must be higher than the
default frequency set by RFREQ. This is required so that
the internal clock does not start a switching cycle prematurely. If SYNC is inactive for an entire clock cycle of
the internal oscillator, the internal oscillator takes over
the switching operation. Choose RFREQ such that fOSC
= 0.9 ✕ fSYNC.
EXT Polarity (MAX1847 only)
The MAX1847 features an option to utilize an N-channel
MOSFET configuration, rather than the typical P-channel MOSFET configuration (Figure 1). In order to drive
the different polarities of these MOSFETs, the MAX1847
is capable of reversing the phase of EXT by 180
degrees. When driving a P-channel MOSFET, connect
POL to GND. When driving an N-Channel MOSFET,
connect POL to VL. These POL connections ensure the
proper polarity for EXT. For design guidance in regard
to this application, refer to the MAX1856 data sheet.
Design Procedure
Initial Specifications
In order to start the design procedure, a few parameters
must be identified: the minimum input voltage expected
(V IN(MIN) ), the maximum input voltage expected
(VIN(MAX)), the desired output voltage (VOUT), and the
expected maximum load current (ILOAD).
Calculate the Equivalent Load Resistance
This is a simple calculation used to shorten the verification equations:
RLOAD = VOUT / ILOAD
______________________________________________________________________________________
11
MAX1846/MAX1847
Detailed Description
The MAX1846/MAX1847 current-mode PWM controller
use an inverting topology that is ideal for generating
output voltages from -2V to -200V. Features include
shutdown, adjustable internal operating frequency or
synchronization to an external clock, soft-start,
adjustable current limit, and a wide (+3V to +16.5V)
input range.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
VIN
+12V
12µF VP1-0190
25V
12.2µH
1:4
CMR1U-02
0.47µF
8
1
2
VL POL
15
IN
SHDN
16
SYNC
MAX1847
0.033µF
270kΩ
150kΩ
3
5
COMP
14
470Ω
13
100pF
100V
12µF
100V
EXT
CS
4
VOUT
-48V AT 100mA
IRLL2705
N.C.
PGND
7, 9
0.05Ω
0.5W
383kΩ
1%
12
FREQ
REF
FB
6
GND
10, 11
1800pF
10.0kΩ
1%
0.1µF
Figure 1. Using an N-Channel MOSFET (MAX1847 only)
Calculate the Duty Cycle
The duty cycle is the ratio of the on-time of the MOSFET
switch to the oscillator period. This is determined by the
ratio of the input voltage to the output voltage. Since
the input voltage typically has a range of operation, a
minimum (DMIN) and maximum (DMAX) duty cycle is
calculated by:
DMIN =
DMAX =
− VOUT + VD
VIN(MAX) − VSW − VLIM − VOUT + VD
− VOUT + VD
VIN(MIN) − VSW − VLIM − VOUT + VD
where VD is the forward drop across the output diode,
VSW is the drop across the external FET when on, and
V LIM is the current-limit threshold. To begin with,
assume VD = 0.5V for a Schottky diode, VSW = 100mV,
and VLIM = 100mV. Remember that VOUT is negative
when using this formula.
Setting the Output Voltage
The output voltage is set using two external resistors to
form a resistive-divider to FB between the output and
REF (refer to R1 and R2 in Figure 1). VREF is nominally
12
1.25V and the regulation voltage for FB is nominally 0.
The load presented to the reference by the feedback
resistors must be less than 500µA. This is to guarantee
that VREF is in regulation (see Electrical Characteristics
Table). Conversely, the current through the feedback
resistors must be large enough so that the leakage current of the FB input (50nA) is insignificant. Therefore,
select R2 so that IR2 is between 50µA and 250µA.
IR2 = VREF / R2
where VREF = 1.25V. A typical value for R2 is 10kΩ.
Once R2 is selected, calculate R1 with the following
equation:
R1 = R2 x (-VOUT / VREF)
Setting the Operating Frequency
The MAX1846/MAX1847 are capable of operating at
switching frequencies from 100kHz to 500kHz. Choice
of operating frequency depends on a number of factors:
1) Noise considerations may dictate setting (or synchronizing) f OSC above or below a certain frequency or band of frequencies, particularly in RF
applications.
______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
3)
4)
5)
Higher frequencies allow the use of smaller value
(hence smaller size) inductors and capacitors.
Higher frequencies consume more operating
power both to operate the IC and to charge and
discharge the gate of the external FET. This tends
to reduce the efficiency at light loads.
Higher frequencies may exhibit lower overall efficiency due to more transition losses in the FET;
however, this shortcoming can often be nullified
by trading some of the inductor and capacitor size
benefits for lower-resistance components.
High-duty-cycle applications may require lower
frequencies to accommodate the controller minimum off-time of 0.4µs. Calculate the maximum
oscillator frequency with the following formula:
fOSC(MAX) =
VIN(MIN) − VSW − VLIM
VIN(MIN) − VSW − VLIM − VOUT + VD
1
×
t OFF(MIN)
tions, most circuits are more efficient and economical
operating in continuous mode, which refers to continuous current in the inductor. In continuous mode there is
a trade-off between efficiency and transient response.
Higher inductance means lower inductor ripple current,
lower peak current, lower switching losses, and, therefore, higher efficiency. Lower inductance means higher
inductor ripple current and faster transient response. A
reasonable compromise is to choose the ratio of inductor ripple current to average continuous current at minimum duty cycle to be 0.4. Calculate the inductor ripple
with the following formula:
IRIPPLE =
(
0.4 × ILOAD(MAX) × VIN(MAX) − VSW − VLIM − VOUT + VD
(VIN(MAX) − VSW − VLIM )
)
Then calculate an inductance value:
L = (VIN(MAX) / IRIPPLE) x (DMIN / fOSC)
Choose the closest standard value. Once again, remember that VOUT is negative when using this formula.
Remember that VOUT is negative when using this formula.
Determining Peak Inductor Current
The oscillator frequency is set by a resistor, RFREQ,
connected from FREQ to GND. The relationship
between fOSC (in Hz) and RFREQ (in Ω) is slightly nonlinear, as illustrated in the Typical Operating
Characteristics. Choose the resistor value from the
graph and check the oscillator frequency using the following formula:
The peak inductor current required for a particular output is:
fOSC =
(
) (
)
1
ILPEAK = ILDC + (ILPP / 2)
where ILDC is the average DC input current and ILPP is
the inductor peak-to-peak ripple current. The ILDC and
ILPP terms are determined as follows:
ILDC =
(
)
2
−19
 5.21 × 10 −7 + 1.92 × 10 −11 × R
× (RFREQ ) 
FREQ − 4.86 × 10


ILPP =
External Synchronization (MAX1847 only)
The SYNC input provides external-clock synchronization (if desired). When SYNC is driven with an external
clock, the frequency of the clock directly sets the
MAX1847’s switching frequency. A rising clock edge
on SYNC is interpreted as a synchronization input. If
the sync signal is lost, the internal oscillator takes over
at the end of the last cycle, and the frequency is
returned to the rate set by RFREQ. Choose RFREQ such
that fOSC = 0.9 x fSYNC.
Choosing Inductance Value
The inductance value determines the operation of the
current-mode regulator. Except for low-current applica-
ILOAD × ( − VOUT + VD )
VIN(MIN) − VSW − VLIM
(VIN(MIN) − VSW − VLIM ) × (− VOUT + VD )
L × fOSC × ( − VOUT + VD )
where L is the selected inductance value. The saturation rating of the selected inductor should meet or
exceed the calculated value for ILPEAK, although most
coil types can be operated up to 20% over their saturation rating without difficulty. In addition to the saturation
criteria, the inductor should have as low a series resistance as possible. For continuous inductor current, the
power loss in the inductor resistance (PLR) is approximated by:
PLR ~ (ILOAD x VOUT / VIN)2 x RL
where RL is the inductor series resistance.
______________________________________________________________________________________
13
MAX1846/MAX1847
2)
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
Once the peak inductor current is calculated, the current sense resistor, RCS, is determined by:
RCS = 85mV / ILPEAK
For high peak inductor currents (>1A), Kelvin-sensing
connections should be used to connect CS and PGND
to R CS . Connect PGND and GND together at the
ground side of RCS. A lowpass filter between RCS and
CS may be required to prevent switching noise from
tripping the current-sense comparator at heavy loads.
Connect a 100Ω resistor between CS and the high side
of RCS, and connect a 1000pF capacitor between CS
and GND.
Checking Slope-Compensation Stability
In a current-mode regulator, the cycle-by-cycle stability
is dependent on slope compensation to prevent subharmonic oscillation at duty cycles greater than 50%.
For the MAX1846/MAX1847, the internal slope compensation is optimized for a minimum inductor value (LMIN)
with respect to duty cycle. For duty cycles greater then
50%, check stability by calculating LMIN using the following equation:
[(
)
LMIN = VIN(MIN) × RCS / MS
[
]
]
× (2 × DMAX − 1) / (1 − DMAX )
where VIN(MIN) is the minimum expected input voltage,
Ms is the Slope Compensation Ramp (41 mV/µs) and
DMAX is the maximum expected duty cycle. If LMIN is
larger than L, increase the value of L to the next standard value that is larger than L MIN to ensure slope
compensation stability.
Power MOSFET Selection
The MAX1846/MAX1847 drive a wide variety of P-channel power MOSFETs (PFETs). The best performance,
especially with input voltages below 5V, is achieved
with low-threshold PFETs that specify on-resistance
with a gate-to-source voltage (VGS) of 2.7V or less.
When selecting a PFET, key parameters include:
1) Total gate charge (QG)
2) Reverse transfer capacitance (CRSS)
3) On-resistance (RDS(ON))
4) Maximum drain-to-source voltage (VDS(MAX))
5) Minimum threshold voltage (VTH(MIN))
At high switching rates, dynamic characteristics (parameters 1 and 2 above) that predict switching losses
may have more impact on efficiency than R DS(ON),
which predicts DC losses. QG includes all capacitance
14
associated with charging the gate. In addition, this
parameter helps predict the current needed to drive the
gate at the selected operating frequency. The power
MOSFET in an inverting converter must have a high
enough voltage rating to handle the input voltage plus
the magnitude of the output voltage and any spikes
induced by leakage inductance.
Choose RDS(ON)(MAX) specified at VGS < VIN(MIN) to be
one to two times RCS. Verify that VIN(MAX) < VGS(MAX)
and VDS(MAX) > VIN(MAX) - VOUT + VD. Choose the riseand fall-times (tR, tF) to be less than 50ns.
Output Capacitor Selection
The output capacitor (COUT) does all the filtering in an
inverting converter. The output ripple is created by the
variations in the charge stored in the output capacitor
with each pulse and the voltage drop across the
capacitor’s equivalent series resistance (ESR) caused
by the current into and out of the capacitor. There are
two properties of the output capacitor that affect ripple
voltage: the capacitance value, and the capacitor’s
ESR. The output ripple due to the output capacitor’s
value is given by:
VRIPPLE-C = (ILOAD ✕ DMAX ✕ TOSC ) / COUT
The output ripple due to the output capacitor’s ESR is
given by:
VRIPPLE-R = ILPP ✕ RESR
These two ripple voltages are additive and the total output ripple is:
VRIPPLE-T = VRIPPLE-C + VRIPPLE-R
The ESR-induced ripple usually dominates this last
equation, so typically output capacitor selection is
based mostly upon the capacitor’s ESR, voltage rating,
and ripple current rating. Use the following formula to
determine the maximum ESR for a desired output ripple
voltage (VRIPPLE-D):
RESR = VRIPPLE-D / ILPP
Select a capacitor with ESR rating less than RESR. The
value of this capacitor is highly dependent on dielectric
type, package size, and voltage rating. In general, when
choosing a capacitor, it is recommended to use low-ESR
capacitor types such as ceramic, organic, or tantalum
capacitors. Ensure that the selected capacitor has sufficient margin to safely handle the maximum ripple current
(ILPP) and the maximum output voltage.
Choosing Compensation Components
The MAX1846/MAX1847 are externally loop-compensated devices. This provides flexibility in designs to
accommodate a variety of applications. Proper com-
______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
Calculating Poles and Zeros
The MAX1846/MAX1847 current-mode architecture
takes the double pole caused by the inductor and output capacitor and shifts one of these poles to a much
higher frequency. This makes loop compensation easier. To compensate these devices, we must know the
center frequencies of the right-half plane zero (zRHP)
and the higher frequency pole (pOUT2). Calculate the
zRHP frequency with the following formula:
2


− (1 − D MAX ) × VIN(MIN) − VOUT × RLOAD


ZRHP =
(2π × VOUT × L)
(
)
The calculations for pOUT2 are very complex. For most
applications where VOUT does not exceed -48V (in a
negative sense), the pOUT2 will not be lower than 1/8th
of the oscillator frequency and is generally at a higher
frequency than zRHP. Therefore:
pOUT2 ≥ 0.125 ✕ fOSC
A pole is created by the output capacitor and the load
resistance. This pole must also be compensated and
its center frequency is given by the formula:
pOUT1 = 1 / (2π ✕ RLOAD ✕ COUT)
Finally, there is a zero introduced by the ESR of the output capacitor. This zero is determined from the following equation:
zESR = 1 / (2π ✕ COUT ✕ RESR)
Calculating the Required Pole Frequency
To ensure stability of the MAX1846/MAX1847, the introduced pole (PDOM) by the compensation network must
roll-off the error amplifier gain to 1 before z RHP or
POUT2 occurs. First calculate the DC open-loop gain to
determine the frequency of the pole to introduce.
(
)
2
G × R × 1 − D

O (
MAX ) × VIN(MIN) − VOUT 
 M
× R

LOAD

ADC = 
R

× VIN(MIN) + TOSC (1 − DMAX )
 CS

B× 
 VIN(MIN)
 
× RLOAD
× RCS + MS1  
 2L


 

(
)
where:
B is the feedback divider attenuation factor =
(-VOUT / VREF),
G M is the error amplifier transconductance =
400 µA/V,
RO is the error amplifier output resistance = 3 MΩ,
M S1 is the slope compensation factor =
[(1.636A / µs) ✕ RCS],
RCS is the selected current sense resistor,
L is the selected inductance value
If zRHP is at a lower frequency than pOUT2, the required
dominant pole frequency is given by:
pDOM = zRHP / ADC
Otherwise the required dominant pole frequency is:
pDOM = pOUT2 / ADC
Determining the Compensation Component Values
Using p DOM, calculate the compensation capacitor
required:
CCOMP = 1 / (2π ✕ RO ✕ pDOM)
Select the next largest standard value of capacitor and
then calculate the compensation resistor required to
cancel out the output-capacitor-induced pole (pOUT1)
determined previously. A zero is needed to cancel the
output-induced pole and the frequency of this zero
must equal pOUT1. Therefore:
zCOMP = pOUT1
RCOMP = RLOAD ✕ COUT / CCOMP
Choose the nearest lower standard value of the resistor. Now check the final values selected for the compensation components:
pCOMP = 1 / [2π ✕ CCOMP x (RO + RCOMP)]
In order for pCOMP to compensate the loop, the openloop gain must reach unity at a lower frequency than
the right-half-plane zero or the second output pole,
whichever is lower in frequency. If the second output
pole and the right-half-plane zero are close together in
frequency, the higher resulting phase shift at unity gain
______________________________________________________________________________________
15
MAX1846/MAX1847
pensation of the control loop is important to prevent
excessive output ripple and poor efficiency caused by
instability. The goal of compensation is to cancel
unwanted poles and zeros in the DC-DC converter’s
transfer function created by the power-switching and
filter elements. More precisely, the objective of compensation is to ensure stability by ensuring that the DCDC converter’s phase shift is less than 180° by a safe
margin, at the frequency where the loop gain falls
below unity. One method for ensuring adequate phase
margin is to introduce corresponding zeros and poles
in the feedback network to approximate a single-pole
response with a -20dB/decade slope all the way to
unity-gain crossover.
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
may require a larger compensation capacitor than calculated. It might take more than a couple of iterations to
obtain a suitable combination.
Finally, the zero introduced by the output capacitor’s
ESR must be compensated. This is accomplished by
placing a capacitor between REF and FB creating a
pole directly in the feedback loop. Calculate the value
of this capacitor using the frequency of zESR and the
selected feedback resistor values with the formula:
R + R2
CFB = RESR × COUT × 1
R1 × R2
Applications Information
Maximum Output Power
The maximum output power that the MAX1846/MAX1847
can provide depends on the maximum input power available and the circuit’s efficiency:
POUT(MAX) = Efficiency ✕ PIN(MAX)
Furthermore, the efficiency and input power are both
functions of component selection. Efficiency losses can
be divided into three categories: 1) resistive losses
across the inductor, MOSFET on-resistance, currentsense resistor, and the ESR of the input and output
capacitors; 2) switching losses due to the MOSFET’s
transition region, and charging the MOSFET’s gate
capacitance; and 3) inductor core losses. Typically
80% efficiency can be assumed for initial calculations.
The required input power depends on the inductor current limit, input voltage, output voltage, output current,
inductor value, and the switching frequency. The maximum output power is approximated by the following
formula:
PMAX = [VIN - (VLIM + ILIM x RDS(ON))] x ILIM x
[1 - (LIR / 2)] x [(-VOUT + VD) / (VIN - VSW - VLIM
- VOUT + VD)]
where I LIM is the peak current limit and LIR is the
inductor current-ripple ratio and is calculated by:
LIR = ILPP / ILDC
Again, remember that V OUT for the MAX1846/
MAX1847 is negative.
Diode Selection
The MAX1846/MAX1847’s high-switching frequency
demands a high-speed rectifier. Schottky diodes are
recommended for most applications because of their
fast recovery time and low forward voltage. Ensure that
the diode’s average current rating exceeds the peak
inductor current by using the diode manufacturer’s data.
Additionally, the diode’s reverse breakdown voltage must
16
exceed the potential difference between VOUT and the
input voltage plus the leakage inductance spikes. For
high output voltages (-50V or more), Schottky diodes may
not be practical because of this voltage requirement. In
these cases, use an ultrafast recovery diode with adequate reverse-breakdown voltage.
Input Filter Capacitor
The input capacitor (CIN) in inverting converter designs
reduces the current peaks drawn from the input supply
and reduces noise injection. The source impedance of
the input supply largely determines the value of CIN.
High source impedance requires high input capacitance, particularly as the input voltage falls. Since
inverting converters act as “constant-power” loads to
their input supply, input current rises as the input voltage falls. Consequently, in low-input-voltage designs,
increasing C IN and/or lowering its ESR can add as
much as 5% to the conversion efficiency.
Bypass Capacitor
In addition to CIN and COUT, other ceramic bypass
capacitors are required with the MAX1846/MAX1847.
Bypass REF to GND with a 0.1µF or larger capacitor.
Bypass VL to GND with a 0.47µF or larger capacitor. All
bypass capacitors should be located as close to their
respective pins as possible.
PC Board Layout Guidelines
Good PC board layout and routing are required in highfrequency-switching power supplies to achieve good
regulation, high efficiency, and stability. It is strongly
recommended that the evaluation kit PC board layouts
be followed as closely as possible. Place power components as close together as possible, keeping their
traces short, direct, and wide. Avoid interconnecting
the ground pins of the power components using vias
through an internal ground plane. Instead, keep the
power components close together and route them in a
“star” ground configuration using component-side copper, then connect the star ground to internal ground
using multiple vias.
Main Application Circuits
The MAX1846/MAX1847 are extremely versatile devices.
Figure 2 shows a generic schematic of the MAX1846.
Table 1 lists component values for several typical applications. These component values also apply to the
MAX1847. The first two applications are featured in the
MAX1846/MAX1847 EV Kit.
______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
MAX1846/MAX1847
VIN
APPLICATION B
ONLY
CIN
22k
P
D1
1
0.47µF
VOUT
10
IN
VL
EXT
CS
9
COUT
L1
8
MAX1846
RCS
3
CCOMP
RCOMP
RFREQ
2
4
COMP
PGND
R1
7
FREQ
REF
FB
5
R2
GND
CFB
6
0.1µF
NOTE: APPLICATIONS A & B USE POS CAPACITORS. APPLICATIONS C & D USE ALUMINUM ELECTROLYTIC CAPACITORS.
Figure 2. MAX1846 Main Application Circuit
Table 1. Component List for Main Application Circuits
A
B
C
D
Input (V)
CIRCUIT ID
12
3 to 5.5
12
12
Output (V)
-5
-12
-48
-72
Output (A)
2
0.4
0.1
0.1
0.047
0.22
0.068
0.1
CCOMP (µF)
CIN (µF)
3 x 10
3 x 22
10
10
COUT (µF)
2 x 100
2 x 47
47
33
CFB (pF)
390
1200
1800
1800
R1 (kΩ) (1%)
40.2
95.3
383
576
R2 (kΩ) (1%)
10
10
10
10
RCOMP (kΩ)
8.2
10
150
1800
RCS (Ω)
0.02
0.02
0.05
0.05
RFREQ (kΩ)
D1
L1 (µH)
P1
150
150
150
150
CMSH5-40
CMSH5-40
CMR1U-02
CMR1U-02
10
10
47
82
FDS6685
FDS6375
IRFR5410
IRFR5410
______________________________________________________________________________________
17
MAX1846/MAX1847
High-Efficiency, Current-Mode,
Inverting PWM Controller
Component Suppliers
SUPPLIER
COMPONENT
PHONE
WEBSITE
Capacitors
803-946-0690
www.avxcorp.com
Diodes
516-435-1110
www.centralsemi.com
Coilcraft
Inductors
847-639-6400
www.coilcraft.com
Dale
Resistors
402-564-3131
www.vishay.com/brands/dale/main.html
Fairchild
MOSFETs
408-721-2181
www.fairchildsemi.com
International
Rectifier
MOSFETs
310-322-3331
www.irf.com
IRC
Resistors
512-992-7900
www.irctt.com
AVX
Central Semiconductor
Kemet
Capacitors
864-963-6300
www.kemet.com
MOSFETs, Diodes
602-303-5454
www.onsemi.com
Capacitors, Resistors
201-348-7522
www.panasonic.com
On Semiconductor
Panasonic
Sanyo
Capacitors
619-661-6835
www.secc.co.jp
Siliconix
MOSFETs
408-988-8000
www.siliconix.com
Sprague
Capacitors
603-224-1961
www.vishay.com/brands/sprague/main.html
Sumida
Inductors
847-956-0666
www.remtechcorp.com
Vitramon
Resistors
203-268-6261
www.vishay.com/brands/vitramon/main.html
Note: Please indicate that you are using the MAX1846/MAX1847 when contacting these component suppliers.
Pin Configurations
TOP VIEW
VL 1
FREQ
2
COMP
3
REF
4
FB
Chip Information
TRANSISTOR COUNT: 2441
PROCESS TECHNOLOGY: BiCMOS
10 IN
MAX1846
5
10-PIN µMAX
POL 1
16 SYNC
VL 2
15 IN
9
EXT
8
CS
7
PGND COMP 4
6
REF 5
12 PGND
FB 6
11 GND
N.C. 7
10 GND
GND
14 EXT
FREQ 3
MAX1847
SHDN 8
13 CS
9
N.C.
16-PIN QSOP
18
______________________________________________________________________________________
High-Efficiency, Current-Mode,
Inverting PWM Controller
10LUMAX.EPS
______________________________________________________________________________________
19
MAX1846/MAX1847
Package Information
High-Efficiency, Current-Mode,
Inverting PWM Controller
QSOP.EPS
MAX1846/MAX1847
Package Information (continued)
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
20 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2001 Maxim Integrated Products
Printed USA
is a registered trademark of Maxim Integrated Products.
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