TI1 LM3405 Constant current buck regulator Datasheet

LM3405
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LM3405 1.6MHz, 1A Constant Current Buck Regulator for Powering LEDs
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FEATURES
DESCRIPTION
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•
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Integrated with a 1A power switch, the LM3405 is a
current-mode control switching buck regulator
designed to provide a simple, high efficiency solution
for driving high power LEDs. With a 0.205V reference
voltage feedback control to minimize power
dissipation, an external resistor sets the current as
needed for driving various types of LEDs. Switching
frequency is internally set to 1.6MHz, allowing small
surface mount inductors and capacitors to be used.
The LM3405 utilizes current-mode control and
internal compensation offering ease of use and
predictable, high performance regulation over a wide
range of operating conditions. Additional features
include user accessible EN/DIM pin for enabling and
PWM dimming of LEDs, thermal shutdown, cycle-bycycle current limit and over-current protection.
1
2
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VIN Operating Range of 3V to 15V
Thin SOT-6 Package
1.6MHz Switching Frequency
300mΩ NMOS Switch
40nA Shutdown Current at VIN = 5V
EN/DIM Input for Enabling and PWM Dimming
of LEDs
Internally Compensated Current-mode Control
Cycle-by-cycle Current Limit
Input Voltage UVLO
Over-current Protection
Thermal Shutdown
APPLICATIONS
•
•
•
•
LED Driver
Constant Current Source
Industrial Lighting
LED Flashlights
TYPICAL APPLICATION CIRCUIT
Efficiency vs LED Current (VIN = 5V)
D2
VIN
BOOST
VIN
C3
C1
L1
VOUT
SW
ON
LM3405
D1
C2
OFF
IF
C4
EN/DIM
FB
GND
R1
1
2
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
Connection Diagram
BOOST
1
6
SW
1
6
GND
2
5
VIN
2
5
FB
3
4
EN/DIM
3
4
Figure 1. 6-Lead SOT
See Package Number DDC (R-PDSO-G6)
Figure 2. Pin 1 Identification
PIN DESCRIPTIONS
Pin(s)
Name
1
BOOST
2
GND
3
FB
4
EN/DIM
Application Information
Voltage at this pin drives the internal NMOS power switch. A bootstrap capacitor is connected between the
BOOST and SW pins.
Signal and Power ground pin. Place the LED current-setting resistor as close as possible to this pin for accurate
current regulation.
Feedback pin. Connect an external resistor from FB to GND to set the LED Current.
Enable control input. Logic high enables operation. Toggling this pin with a periodic logic square wave of
varying duty cycle at different frequencies controls the brightness of LEDs. Do not allow this pin to float or be
greater than VIN + 0.3V.
5
VIN
Input supply voltage. Connect a bypass capacitor locally from this pin to GND.
6
SW
Switch pin. Connect this pin to the inductor, catch diode, and bootstrap capacitor.
ABSOLUTE MAXIMUM RATINGS
(1)
If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors
for availability and specifications.
VALUE / UNIT
VIN
–0.5V to 20V
SW Voltage
–0.5V to 20V
Boost Voltage
–0.5V to 26V
Boost to SW Voltage
–0.5V to 6.0V
FB Voltage
–0.5V to 3.0V
EN/DIM Voltage
–0.5V to (VIN + 0.3V)
Junction Temperature
ESD Susceptibility
150°C
(2)
2kV
Storage Temperature
–65°C to +150°C
Soldering Information Infrared/Convection Reflow (15sec)
(1)
(2)
2
220°C
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under
which the device is intended to be functional. For specific specifications and test conditions, see the Electrical Characteristics.
Human body model, 1.5kΩ in series with 100pF.
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OPERATING RATINGS
(1)
VALUE / UNIT
VIN
3V to 15V
EN/DIM voltage
–0.5V to (VIN + 0.3V)
Boost to SW Voltage
2.5V to 5.5V
Junction Temperature Range
–40°C to +125°C
Thermal Resistance θJA (2)
(1)
(2)
118°C/W
Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings define the conditions under
which the device is intended to be functional. For specific specifications and test conditions, see the Electrical Characteristics.
Thermal shutdown will occur if the junction temperature (TJ) exceeds 165°C. The maximum allowable power dissipation (PD) at any
ambient temperature (TA) is PD = (TJ(MAX) – TA)/θJA . This number applies to packages soldered directly onto a 3" x 3" PC board with
2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still air, θJA = 204°C/W.
ELECTRICAL CHARACTERISTICS
Unless otherwise specified, VIN = 12V. Limits in standard type are for TJ = 25°C only; limits in boldface type apply over the
junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits are specified through test, design, or
statistical correlation. Typical values represent the most likely parametric norm, and are provided for reference purposes only.
Symbol
VFB
ΔVFB/(ΔVINxVFB)
IFB
UVLO
Parameter
Conditions
Feedback Voltage
Min
Typ
Max
0.188
0.205
0.220
Feedback Voltage Line Regulation VIN = 3V to 15V
Feedback Input Bias Current
Under-voltage Lockout
V
0.01
Sink/Source
VIN Rising
VIN Falling
Units
1.9
UVLO Hysteresis
%/V
10
250
2.74
2.95
nA
V
2.3
0.44
fSW
Switching Frequency
DMAX
Maximum Duty Cycle
VFB = 0V
RDS(ON)
Switch ON Resistance
VBOOST - VSW = 3V
Switch Current Limit
VBOOST - VSW = 3V, VIN = 3V
Quiescent Current
Quiescent Current (Shutdown)
Enable Threshold Voltage
VEN/DIM Rising
Shutdown Threshold Voltage
VEN/DIM Falling
EN/DIM Pin Current
Sink/Source
0.01
µA
Switch Leakage
VIN = 15V
0.1
µA
ICL
IQ
VEN/DIM_TH
IEN/DIM
ISW
1.2
1.6
85
94
V
1.9
MHz
300
600
mΩ
2.0
2.8
A
Switching, VFB = 0.195V
1.8
2.8
mA
VEN/DIM = 0V
0.3
1.2
%
µA
1.8
V
0.4
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TYPICAL PERFORMANCE CHARACTERISTICS
Unless otherwise specified, VIN = 12V, VBOOST - VSW = 5V and TA = 25°C.
4
Efficiency vs LED Current
Efficiency vs Input Voltage (IF = 1A)
Figure 3.
Figure 4.
Efficiency vs Input Voltage (IF = 0.7A)
Efficiency vs Input Voltage (IF = 0.35A)
Figure 5.
Figure 6.
VFB vs Temperature
Oscillator Frequency vs Temperature
Figure 7.
Figure 8.
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TYPICAL PERFORMANCE CHARACTERISTICS (continued)
Unless otherwise specified, VIN = 12V, VBOOST - VSW = 5V and TA = 25°C.
Current Limit vs Temperature
RDS(ON) vs Temperature (VBOOST - VSW = 3V)
Figure 9.
Figure 10.
Quiescent Current vs Temperature
Startup Response to EN/DIM Signal
(VIN = 15V, IF = 0.2A)
Figure 11.
Figure 12.
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Block Diagram
VIN
VIN
Current Sense Amplifier
OFF
Thermal
Shutdown
Undervoltage
Lockout
Oscillator
RSENSE
+
-
Current
Limit
+
+
+
Corrective Ramp
Error
Signal
C1
BOOST
Output
Control
Logic
Reset
Pulse
ISENSE
D2
0.3:
Switch
Driver
C3
SW
Over-current
Comparator
-
EN/DIM
ON
Internal
Regulator
and
Enable
Circuit
D1
+
PWM
Comparator
L1
0.328V
VOUT
IL
C2
C4
+
-
IF
LED1
FB
Internal
Compensation
+
Error Amplifier
+
-
VREF
R1
0.205V
GND
Figure 13. Simplified Block Diagram
6
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APPLICATION INFORMATION
THEORY OF OPERATION
The LM3405 is a PWM, current-mode control switching buck regulator designed to provide a simple, high
efficiency solution for driving LEDs with a preset switching frequency of 1.6MHz. This high frequency allows the
LM3405 to operate with small surface mount capacitors and inductors, resulting in LED drivers that need only a
minimum amount of board space. The LM3405 is internally compensated, simple to use, and requires few
external components.
The following description of operation of the LM3405 will refer to the Simplified Block Diagram (Figure 13) and to
the waveforms in Figure 14. The LM3405 supplies a regulated output current by switching the internal NMOS
power switch at constant frequency and variable duty cycle. A switching cycle begins at the falling edge of the
reset pulse generated by the internal oscillator. When this pulse goes low, the output control logic turns on the
internal NMOS power switch. During this on-time, the SW pin voltage (VSW) swings up to approximately VIN, and
the inductor current (IL) increases with a linear slope. IL is measured by the current sense amplifier, which
generates an output proportional to the switch current. The sense signal is summed with the regulator’s
corrective ramp and compared to the error amplifier’s output, which is proportional to the difference between the
feedback voltage and VREF. When the PWM comparator output goes high, the internal power switch turns off until
the next switching cycle begins. During the switch off-time, inductor current discharges through the catch diode
D1, which forces the SW pin to swing below ground by the forward voltage (VD1) of the catch diode. The
regulator loop adjusts the duty cycle (D) to maintain a constant output current (IF) through the LED, by forcing FB
pin voltage to be equal to VREF (0.205V).
VSW
D = TON/TSW
VIN
SW
Voltage
TOFF
TON
0
-VD1
t
IL
TSW
ILPK
IF
'iL
Inductor
Current
t
0
Figure 14. SW Pin Voltage and Inductor Current Waveforms of LM3405
BOOST FUNCTION
Capacitor C3 and diode D2 in Figure 13 are used to generate a voltage VBOOST. The voltage across C3, VBOOST VSW, is the gate drive voltage to the internal NMOS power switch. To properly drive the internal NMOS switch
during its on-time, VBOOST needs to be at least 2.5V greater than VSW. Large value of VBOOST - VSW is
recommended to achieve better efficiency by minimizing both the internal switch ON resistance (RDS(ON)), and the
switch rise and fall times. However, VBOOST - VSW should not exceed the maximum operating limit of 5.5V.
When the LM3405 starts up, internal circuitry from VIN supplies a 20mA current to the BOOST pin, flowing out of
the BOOST pin into C3. This current charges C3 to a voltage sufficient to turn the switch on. The BOOST pin will
continue to source current to C3 until the voltage at the feedback pin is greater than 123mV.
There are various methods to derive VBOOST:
1. From the input voltage (VIN)
2. From the output voltage (VOUT)
3. From a shunt or series zener diode
4. From an external distributed voltage rail (VEXT)
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The first method is shown in the Simplified Block Diagram of Figure 13. Capacitor C3 is charged via diode D2 by
VIN. During a normal switching cycle, when the internal NMOS power switch is off (TOFF) (refer to Figure 14),
VBOOST equals VIN minus the forward voltage of D2 (VD2), during which the current in the inductor (L1) forward
biases the catch diode D1 (VD1). Therefore the gate drive voltage stored across C3 is:
VBOOST - VSW = VIN - VD2 + VD1
(1)
When the NMOS switch turns on (TON), the switch pin rises to:
VSW = VIN – (RDS(ON) x IL)
(2)
Since the voltage across C3 remains unchanged, VBOOST is forced to rise thus reverse biasing D2. The voltage at
VBOOST is then:
VBOOST = 2VIN – (RDS(ON) x IL) – VD2 + VD1
(3)
Depending on the quality of the diodes D1 and D2, the gate drive voltage in this method can be slightly less or
larger than the input voltage VIN. For best performance, ensure that the variation of the input supply does not
cause the gate drive voltage to fall outside the recommended range:
2.5V < VIN - VD2 + VD1 < 5.5V
(4)
The second method for deriving the boost voltage is to connect D2 to the output as shown in Figure 15. The gate
drive voltage in this configuration is:
VBOOST - VSW = VOUT – VD2 + VD1
(5)
Since the gate drive voltage needs to be in the range of 2.5V to 5.5V, the output voltage VOUT should be limited
to a certain range. For the calculation of VOUT, see OUTPUT VOLTAGE section.
Figure 15. VBOOST derived from VOUT
The third method can be used in the applications where both VIN and VOUT are greater than 5.5V. In these cases,
C3 cannot be charged directly from these voltages; instead C3 can be charged from VIN or VOUT minus a zener
voltage (VD3) by placing a zener diode D3 in series with D2 as shown in Figure 16. When using a series zener
diode from the input, the gate drive voltage is VIN - VD3 - VD2 + VD1.
Figure 16. VBOOST derived from VIN through a Series Zener
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An alternate method is to place the zener diode D3 in a shunt configuration as shown in Figure 17. A small
350mW to 500mW, 5.1V zener in a SOT or SOD package can be used for this purpose. A small ceramic
capacitor such as a 6.3V, 0.1µF capacitor (C5) should be placed in parallel with the zener diode. When the
internal NMOS switch turns on, a pulse of current is drawn to charge the internal NMOS gate capacitance. The
0.1µF parallel shunt capacitor ensures that the VBOOST voltage is maintained during this time. Resistor R2 should
be chosen to provide enough RMS current to the zener diode and to the BOOST pin. A recommended choice for
the zener current (IZENER) is 1mA. The current IBOOST into the BOOST pin supplies the gate current of the NMOS
power switch. It reaches a maximum of around 3.6mA at the highest gate drive voltage of 5.5V over the LM3405
operating range.
For the worst case IBOOST, increase the current by 50%. In that case, the maximum boost current will be:
IBOOST-MAX = 1.5 x 3.6mA = 5.4mA
(6)
R2 will then be given by:
R2 = (VIN - VZENER) / (IBOOST_MAX + IZENER)
(7)
For example, let VIN = 12V, VZENER = 5V, IZENER = 1mA, then:
R2 = (12V - 5V) / (5.4mA + 1mA) = 1.09kΩ
(8)
Figure 17. VBOOST derived from VIN through a Shunt Zener
The fourth method can be used in an application which has an external low voltage rail, VEXT. C3 can be charged
through D2 from VEXT, independent of VIN and VOUT voltage levels. Again for best performance, ensure that the
gate drive voltage, VEXT - VD2 + VD1, falls in the range of 2.5V to 5.5V.
SETTING THE LED CURRENT
LM3405 is a constant current buck regulator. The LEDs are connected between VOUT and FB pin as shown in the
Typical Application Circuit. The FB pin is at 0.205V in regulation and therefore the LED current IF is set by VFB
and the resistor R1 from FB to ground by the following equation:
IF = VFB / R1
(9)
IF should not exceed the 1A current capability of LM3405 and therefore R1 minimum must be approximately
0.2Ω. IF should also be kept above 200mA for stable operation, and therefore R1 maximum must be
approximately 1Ω. If average LED currents less than 200mA are desired, the EN/DIM pin can be used for PWM
dimming. See LED PWM DIMMING section.
OUTPUT VOLTAGE
The output voltage is primarily determined by the number of LEDs (n) connected from VOUT to FB pin and
therefore VOUT can be written as :
VOUT = ((n x VF) + VFB)
(10)
where VF is the forward voltage of one LED at the set LED current level (see LED manufacturer datasheet for
forward characteristics curve).
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ENABLE MODE / SHUTDOWN MODE
The LM3405 has both enable and shutdown modes that are controlled by the EN/DIM pin. Connecting a voltage
source greater than 1.8V to the EN/DIM pin enables the operation of LM3405, while reducing this voltage below
0.4V places the part in a low quiescent current (0.3µA typical) shutdown mode. There is no internal pull-up on
EN/DIM pin, therefore an external signal is required to initiate switching. Do not allow this pin to float or rise to
0.3V above VIN. It should be noted that when the EN/DIM pin voltage rises above 1.8V while the input voltage is
greater than UVLO, there is a finite delay before switching starts. During this delay the LM3405 will go through a
power on reset state after which the internal soft-start process commences. The soft-start process limits the
inrush current and brings up the LED current (IF) in a smooth and controlled fashion. The total combined duration
of the power on reset delay, soft-start delay and the delay to fully establish the LED current is in the order of
100µs (refer to Figure 23).
The simplest way to enable the operation of LM3405 is to connect the EN/DIM pin to VIN which allows self startup of LM3405 whenever the input voltage is applied. However, when an input voltage of slow rise time is used to
power the application and if both the input voltage and the output voltage are not fully established before the softstart time elapses, the control circuit will command maximum duty cycle operation of the internal power switch to
bring up the output voltage rapidly. When the feedback pin voltage exceeds 0.205V, the duty cycle will have to
reduce from the maximum value accordingly, to maintain regulation. It takes a finite amount of time for this
reduction of duty cycle and this will result in a spike in LED current for a short duration as shown in Figure 18. In
applications where this LED current overshoot is undesirable, EN/DIM pin voltage can be delayed with respect to
VIN such that VIN is fully established before the EN/DIM pin voltage reaches the enable threshold. This delay can
be implemented by a simple Ra-Ca network as shown in Figure 19. The effect of adding this Ra-Ca network on
the LED current is shown in Figure 20. For a fast rising input voltage (200µs for example), there is no need to
delay the EN/DIM signal since soft-start can smoothly bring up the LED current as shown in Figure 21.
Figure 18. Startup Response to VIN with 5ms rise time
Figure 19. EN/DIM delayed with respect to VIN
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Figure 20. Startup Response to VIN with EN/DIM delayed
Figure 21. Startup Response to VIN with 200µs rise time
LED PWM DIMMING
The LED brightness can be controlled by applying a periodic pulse signal to the EN/DIM pin and varying its
frequency and/or duty cycle. This so-called PWM dimming method controls the average light output by pulsing
the LED current between the set value and zero. A logic high level at the EN/DIM pin turns on the LED current
whereas a logic low level turns off the LED current. Figure 22 shows a typical LED current waveform in PWM
dimming mode. As explained in the previous section, there is approximately a 100µs delay from the EN/DIM
signal going high to fully establishing the LED current as shown in Figure 23. This 100µs delay sets a maximum
frequency limit for the driving signal that can be applied to the EN/DIM pin for PWM dimming. Figure 24 shows
the average LED current versus duty cycle of PWM dimming signal for various frequencies. The applicable
frequency range to drive LM3405 for PWM dimming is from 100Hz to 5kHz. The dimming ratio reduces
drastically when the applied PWM dimming frequency is greater than 5kHz.
Figure 22. PWM Dimming of LEDs using the EN/DIM Pin
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Figure 23. Startup Response to EN/DIM with IF = 1A
Figure 24. Average LED Current versus Duty Cycle of PWM Dimming Signal at EN/DIM Pin
UNDER-VOLTAGE LOCKOUT
Under-voltage lockout (UVLO) prevents the LM3405 from operating until the input voltage exceeds 2.74V
(typical). The UVLO threshold has approximately 440mV of hysteresis, so the part will operate until VIN drops
below 2.3V (typical). Hysteresis prevents the part from turning off during power up if VIN is non-monotonic.
CURRENT LIMIT
The LM3405 uses cycle-by-cycle current limit to protect the internal power switch. During each switching cycle, a
current limit comparator detects if the power switch current exceeds 2.0A (typical), and turns off the switch until
the next switching cycle begins.
OVER-CURRENT PROTECTION
The LM3405 has a built in over-current comparator that compares the FB pin voltage to a threshold voltage that
is 60% higher than the internal reference VREF. Once the FB pin voltage exceeds this threshold level (typically
328mV), the internal NMOS power switch is turned off, which allows the feedback voltage to decrease towards
regulation. This threshold provides an upper limit for the LED current. LED current overshoot is limited to
328mV/R1 by this comparator during transients.
THERMAL SHUTDOWN
Thermal shutdown limits total power dissipation by turning off the internal power switch when the IC junction
temperature exceeds 165°C. After thermal shutdown occurs, the power switch does not turn on until the junction
temperature drops below approximately 150°C.
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DESIGN GUIDE
INDUCTOR (L1)
The Duty Cycle (D) can be approximated quickly using the ratio of output voltage (VOUT) to input voltage (VIN):
D=
VOUT
VIN
(11)
The catch diode (D1) forward voltage drop and the voltage drop across the internal NMOS must be included to
calculate a more accurate duty cycle. Calculate D by using the following formula:
VOUT + VD1
D=
VIN + VD1 - VSW
(12)
VSW can be approximated by:
VSW = IF x RDS(ON)
(13)
The diode forward drop (VD1) can range from 0.3V to 0.7V depending on the quality of the diode. The lower VD1
is, the higher the operating efficiency of the converter.
The inductor value determines the output ripple current (ΔiL, as defined in Figure 14). Lower inductor values
decrease the size of the inductor, but increases the output ripple current. An increase in the inductor value will
decrease the output ripple current. The ratio of ripple current to LED current is optimized when it is set between
0.3 and 0.4 at 1A LED current. This ratio r is defined as:
r=
'iL
lF
(14)
One must also ensure that the minimum current limit (1.2A) is not exceeded, so the peak current in the inductor
must be calculated. The peak current (ILPK) in the inductor is calculated as:
ILPK = IF + ΔiL/2
(15)
When the designed maximum output current is reduced, the ratio r can be increased. At a current of 0.2A, r can
be made as high as 0.7. The ripple ratio can be increased at lighter loads because the net ripple is actually quite
low, and if r remains constant the inductor value can be made quite large. An equation empirically developed for
the maximum ripple ratio at any current below 2A is:
r = 0.387 x IOUT-0.3667
(16)
Note that this is just a guideline.
The LM3405 operates at a high frequency allowing the use of ceramic output capacitors without compromising
transient response. Ceramic capacitors allow higher inductor ripple without significantly increasing LED current
ripple. See the output capacitor and feed-forward capacitor sections for more details on LED current ripple.
Now that the ripple current or ripple ratio is determined, the inductance is calculated by:
L=
VOUT + VD1
IF x r x fSW
x (1-D)
(17)
where fSW is the switching frequency and IF is the LED current. When selecting an inductor, make sure that it is
capable of supporting the peak output current without saturating. Inductor saturation will result in a sudden
reduction in inductance and prevent the regulator from operating correctly. Because of the operating frequency of
LM3405, ferrite based inductors are preferred to minimize core losses. This presents little restriction since the
variety of ferrite based inductors is huge. Lastly, inductors with lower series resistance (DCR) will provide better
operating efficiency. For recommended inductor selection, refer to Circuit Examples and Recommended
Inductance Range in Table 1. Note that it is a good practice to use small inductance value at light load (for
example, IF = 0.2A) to increase inductor current ramp signal, such that noise immunity is improved.
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Table 1. Recommended Inductance Range
IF
Inductance Range and Inductor Current Ripple
4.7µH-10µH
1.0A
Inductance
4.7µH
6.8µH
10µH
ΔiL / IF*
51%
35%
24%
6.8µH-15µH
0.6A
Inductance
6.8µH
10µH
15µH
ΔiL / IF*
58%
40%
26%
4.7µH**-22µH
0.2A
Inductance
10µH
15µH
22µH
ΔiL / IF*
119%
79%
54%
INPUT CAPACITOR (C1)
An input capacitor is necessary to ensure that VIN does not drop excessively during switching transients. The
primary specifications of the input capacitor are capacitance, voltage rating, RMS current rating, and ESL
(Equivalent Series Inductance). The input voltage rating is specifically stated by the capacitor manufacturer.
Make sure to check any recommended deratings and also verify if there is any significant change in capacitance
at the operating input voltage and the operating temperature. The input capacitor maximum RMS input current
rating (IRMS-IN) must be greater than:
IRMS-IN = IF x
r2
Dx 1-D+
12
(18)
It can be shown from the above equation that maximum RMS capacitor current occurs when D = 0.5. Always
calculate the RMS at the point where the duty cycle D, is closest to 0.5. The ESL of an input capacitor is usually
determined by the effective cross sectional area of the current path. A large leaded capacitor will have high ESL
and a 0805 ceramic chip capacitor will have very low ESL. At the operating frequency of the LM3405, certain
capacitors may have an ESL so large that the resulting inductive impedance (2πfL) will be higher than that
required to provide stable operation. It is strongly recommended to use ceramic capacitors due to their low ESR
and low ESL. A 10µF multilayer ceramic capacitor (MLCC) is a good choice for most applications. In cases
where large capacitance is required, use surface mount capacitors such as Tantalum capacitors and place at
least a 1µF ceramic capacitor close to the VIN pin. For MLCCs it is recommended to use X7R or X5R dielectrics.
Consult capacitor manufacturer datasheet to see how rated capacitance varies over operating conditions.
OUTPUT CAPACITOR (C2)
The output capacitor is selected based upon the desired reduction in LED current ripple. A 1µF ceramic capacitor
results in very low LED current ripple for most applications. Due to the high switching frequency, the 1µF
capacitor alone (without feed-forward capacitor C4) can filter more than 90% of the inductor current ripple for
most applications where the sum of LED dynamic resistance and R1 is larger than 1Ω. Since the internal
compensation is tailored for small output capacitance with very low ESR, it is strongly recommended to use a
ceramic capacitor with capacitance less than 3.3µF.
Given the availability and quality of MLCCs and the expected output voltage of designs using the LM3405, there
is really no need to review other capacitor technologies. A benefit of ceramic capacitors is their ability to bypass
high frequency noise. A certain amount of switching edge noise will couple through the parasitic capacitances in
the inductor to the output. A ceramic capacitor will bypass this noise. In cases where large capacitance is
required, use Electrolytic or Tantalum capacitors with large ESR, and verify the loop performance on bench. Like
the input capacitor, recommended multilayer ceramic capacitors are X7R or X5R. Again, verify actual
capacitance at the desired operating voltage and temperature.
Check the RMS current rating of the capacitor. The maximum RMS current rating of the capacitor is:
IRMS-OUT = IF x
r
12
(19)
One may select a 1206 size ceramic capacitor for C2, since its current rating is typically higher than 1A, more
than enough for the requirement.
14
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FEED-FORWARD CAPACITOR (C4)
The feed-forward capacitor (designated as C4) connected in parallel with the LED string is required to provide
multiple benefits to the LED driver design. It greatly improves the large signal transient response and suppresses
LED current overshoot that may otherwise occur during PWM dimming; it also helps to shape the rise and fall
times of the LED current pulse during PWM dimming thus reducing EMI emission; it reduces LED current ripple
by bypassing some of inductor ripple from flowing through the LED. For most applications, a 1µF ceramic
capacitor is sufficient. In fact, the combination of a 1µF feed-forward ceramic capacitor and a 1µF output ceramic
capacitor leads to less than 1% current ripple flowing through the LED. Lower and higher C4 values can be used,
but bench validation is required to ensure the performance meets the application requirement.
Figure 25 shows a typical LED current waveform during PWM dimming without feed-forward capacitor. At the
beginning of each PWM cycle, overshoot can be seen in the LED current. Adding a 1µF feed-forward capacitor
can totally remove the overshoot as shown in Figure 26.
Figure 25. PWM Dimming without Feed-Forward Capacitor
Figure 26. PWM Dimming with a 1µF Feed-Forward Capacitor
CATCH DIODE (D1)
The catch diode (D1) conducts during the switch off-time. A Schottky diode is required for its fast switching time
and low forward voltage drop. The catch diode should be chosen such that its current rating is greater than:
ID1 = IF x (1-D)
(20)
The reverse breakdown rating of the diode must be at least the maximum input voltage plus appropriate margin.
To improve efficiency, choose a Schottky diode with a low forward voltage drop.
BOOST DIODE (D2)
A standard diode such as the 1N4148 type is recommended. For VBOOST circuits derived from voltages less than
3.3V, a small-signal Schottky diode is recommended for better efficiency. A good choice is the BAT54 small
signal diode.
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BOOST CAPACITOR (C3)
A 0.01µF ceramic capacitor with a voltage rating of at least 6.3V is sufficient. The X7R and X5R MLCCs provide
the best performance.
POWER LOSS ESTIMATION
The main power loss in LM3405 includes three basic types of loss in the internal power switch: conduction loss,
switching loss, and gate charge loss. In addition, there is loss associated with the power required for the internal
circuitry of IC.
The conduction loss is calculated as:
(21)
If the inductor ripple current is fairly small (for example, less than 40%) , the conduction loss can be simplified to:
PCOND = IF2 x RDS(ON) x D
(22)
The switching loss occurs during the switch on and off transition periods, where voltage and current overlap
resulting in power loss. The simplest means to determine this loss is to empirically measure the rise and fall
times (10% to 90%) of the voltage at the switch pin.
Switching power loss is calculated as follows:
PSW = 0.5 x VIN x IF x fSW x ( TRISE + TFALL )
(23)
The gate charge loss is associated with the gate charge QG required to drive the switch:
PG = fSW x VIN x QG
(24)
The power loss required for operation of the internal circuitry:
PQ = IQ x VIN
(25)
IQ is the quiescent operating current, and is typically around 1.8mA for the LM3405.
The total power loss in the IC is:
PINTERNAL = PCOND + PSW + PG + PQ
(26)
An example of power losses for a typical application is shown in Table 2:
Table 2. Power Loss Tabulation
Conditions
Power loss
VIN
12V
VOUT
4.1V
IOUT
1.0A
VD1
0.45V
RDS(ON)
300mΩ
PCOND
111mW
PSW
288mW
fSW
1.6MHz
TRISE
18ns
TFALL
12ns
IQ
1.8mA
PQ
22mW
QG
1.4nC
PG
27mW
D is calculated to be 0.37
spacer
spacer
Σ ( PCOND + PSW + PQ + PG ) = PINTERNAL
(27)
spacer
PINTERNAL = 448mW
16
(28)
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PCB Layout Considerations
When planning layout there are a few things to consider when trying to achieve a clean, regulated output. The
most important consideration when completing the layout is the close coupling of the GND connections of the
input capacitor C1 and the catch diode D1. These ground ends should be close to one another and be connected
to the GND plane with at least two through-holes. Place these components as close to the IC as possible. The
next consideration is the location of the GND connection of the output capacitor C2, which should be near the
GND connections of C1 and D1.
There should be a continuous ground plane on the bottom layer of a two-layer board except under the switching
node island.
The FB pin is a high impedance node and care should be taken to make the FB trace short to avoid noise pickup
that causes inaccurate regulation. The LED current setting resistor R1 should be placed as close as possible to
the IC, with the GND of R1 placed as close as possible to the GND of the IC. The VOUT trace to LED anode
should be routed away from the inductor and any other traces that are switching.
High AC currents flow through the VIN, SW and VOUT traces, so they should be as short and wide as possible.
Radiated noise can be decreased by choosing a shielded inductor.
The remaining components should also be placed as close as possible to the IC. Please see Application Note
AN-1229 for further considerations and the LM3405 demo board as an example of a four-layer layout.
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LM3405 Circuit Examples
D2
VIN
BOOST
VIN
C3
C1
L1
VOUT
SW
LM3405
D1
DC or
PWM
C2
C4
EN/DIM
IF
LED1
FB
GND
R1
Figure 27. VBOOST derived from VIN
( VIN = 5V, IF = 1A )
Table 3. Bill of Materials for Figure 27
Part ID
Part Value
Part Number
Manufacturer
U1
1A LED Driver
LM3405
National Semiconductor
C1, Input Cap
10µF, 6.3V, X5R
C3216X5R0J106M
TDK
C2, Output Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
C3, Boost Cap
0.01µF, 16V, X7R
0805YC103KAT2A
AVX
C4, Feedforward Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
D1, Catch Diode
Schottky, 0.37V at 1A, VR = 10V
MBRM110LT1G
ON Semiconductor
D2, Boost Diode
Schottky, 0.36V at 15mA
CMDSH-3
Central Semiconductor
L1
4.7µH, 1.6A
SLF6028T-4R7M1R6
TDK
R1
0.2Ω, 0.5W, 1%
WSL2010R2000FEA
Vishay
LED1
1A, White LED
LXHL-PW09
Lumileds
18
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D2
VIN
BOOST
VIN
C3
C1
L1
VOUT
SW
LM3405
D1
DC or
PWM
C2
EN/DIM
C4
IF
LED1
FB
GND
R1
Figure 28. VBOOST derived from VOUT
( VIN = 12V, IF = 1A )
Table 4. Bill of Materials for Figure 28
Part ID
Part Value
Part Number
Manufacturer
U1
1A LED Driver
LM3405
National Semiconductor
C1, Input Cap
10µF, 25V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
C3, Boost Cap
0.01µF, 16V, X7R
0805YC103KAT2A
AVX
C4, Feedforward Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
D1, Catch Diode
Schottky, 0.5V at 1A, VR = 30V
SS13
Vishay
D2, Boost Diode
Schottky, 0.36V at 15mA
CMDSH-3
Central Semiconductor
L1
4.7µH, 1.6A
SLF6028T-4R7M1R6
TDK
R1
0.2Ω, 0.5W, 1%
WSL2010R2000FEA
Vishay
LED1
1A, White LED
LXHL-PW09
Lumileds
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LM3405
SNVS429B – OCTOBER 2006 – REVISED MAY 2013
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C5
D3
R2
D2
VIN
BOOST
VIN
C3
C1
L1
VOUT
SW
LM3405
D1
C2
C4
DC or
PWM
IF
LED1
EN/DIM
FB
GND
R1
Figure 29. VBOOST derived from VIN through a Shunt Zener Diode (D3)
( VIN = 15V, IF = 1A )
Table 5. Bill of Materials for Figure 29
Part ID
Part Value
Part Number
Manufacturer
U1
1A LED Driver
LM3405
National Semiconductor
C1, Input Cap
10µF, 25V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
C3, Boost Cap
0.01µF, 16V, X7R
0805YC103KAT2A
AVX
C4, Feedforward Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
C5, Shunt Cap
0.1µF, 16V, X7R
GRM219R71C104KA01D
Murata
D1, Catch Diode
Schottky, 0.5V at 1A, VR = 30V
SS13
Vishay
D2, Boost Diode
Schottky, 0.36V at 15mA
CMDSH-3
Central Semiconductor
D3, Zener Diode
4.7V, 350mW, SOT
BZX84C4V7
Fairchild
L1
6.8µH, 1.5A
SLF6028T-6R8M1R5
TDK
R1
0.2Ω, 0.5W, 1%
WSL2010R2000FEA
Vishay
R2
1.91kΩ, 1%
CRCW08051K91FKEA
Vishay
LED1
1A, White LED
LXHL-PW09
Lumileds
20
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SNVS429B – OCTOBER 2006 – REVISED MAY 2013
D3
D2
BOOST
VIN
VIN
C3
C1
L1
VOUT
SW
LM3405
DC or
PWM
D1
C2
C4
EN/DIM
IF
LED1
FB
GND
R1
Figure 30. VBOOST derived from VIN through a Series Zener Diode (D3)
( VIN = 15V, IF = 1A )
Table 6. Bill of Materials for Figure 30
Part ID
Part Value
Part Number
Manufacturer
U1
1A LED Driver
LM3405
National Semiconductor
C1, Input Cap
10µF, 25V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
C3, Boost Cap
0.01µF, 16V, X7R
0805YC103KAT2A
AVX
C4, Feedforward Cap
1µF, 10V, X7R
GRM319R71A105KC01D
Murata
D1, Catch Diode
Schottky, 0.5V at 1A, VR = 30V
SS13
Vishay
D2, Boost Diode
Schottky, 0.36V at 15mA
CMDSH-3
Central Semiconductor
D3, Zener Diode
11V, 350mW, SOT
BZX84C11
Fairchild
L1
6.8µH, 1.5A
SLF6028T-6R8M1R5
TDK
R1
0.2Ω, 0.5W, 1%
WSL2010R2000FEA
Vishay
LED1
1A, White LED
LXHL-PW09
Lumileds
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D3
D2
VIN
BOOST
VIN
C3
C1
L1
VOUT
SW
LM3405
D1
C2
LED1
IF
C4
DC or
PWM
EN/DIM
LED2
FB
GND
R1
Figure 31. VBOOST derived from VOUT through a Series Zener Diode (D3)
( VIN = 15V, IF = 1A )
Table 7. Bill of Materials for Figure 31
Part ID
Part Value
Part Number
Manufacturer
U1
1A LED Driver
LM3405
National Semiconductor
C1, Input Cap
10µF, 25V, X5R
ECJ-3YB1E106K
Panasonic
C2, Output Cap
1µF, 16V, X7R
GRM319R71A105KC01D
Murata
C3, Boost Cap
0.01µF, 16V, X7R
0805YC103KAT2A
AVX
C4, Feedforward Cap
1µF, 16V, X7R
GRM319R71A105KC01D
Murata
D1, Catch Diode
Schottky, 0.5V at 1A, VR = 30V
SS13
Vishay
D2, Boost Diode
Schottky, 0.36V at 15mA
CMDSH-3
Central Semiconductor
D3, Zener Diode
3.9V, 350mW, SOT
BZX84C3V9
Fairchild
L1
6.8µH, 1.5A
SLF6028T-6R8M1R5
TDK
R1
0.2Ω, 0.5W, 1%
WSL2010R2000FEA
Vishay
LED1
1A, White LED
LXHL-PW09
Lumileds
LED2
1A, White LED
LXHL-PW09
Lumileds
22
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REVISION HISTORY
Changes from Revision A (May 2013) to Revision B
•
Page
Changed layout of National Data Sheet to TI format .......................................................................................................... 22
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PACKAGE OPTION ADDENDUM
www.ti.com
2-May-2013
PACKAGING INFORMATION
Orderable Device
Status
(1)
Package Type Package Pins Package
Drawing
Qty
Eco Plan
Lead/Ball Finish
(2)
MSL Peak Temp
Op Temp (°C)
Top-Side Markings
(3)
(4)
LM3405XMK/NOPB
ACTIVE
SOT
DDC
6
1000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SPNB
LM3405XMKX/NOPB
ACTIVE
SOT
DDC
6
3000
Green (RoHS
& no Sb/Br)
CU SN
Level-1-260C-UNLIM
-40 to 125
SPNB
(1)
The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2)
Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability
information and additional product content details.
TBD: The Pb-Free/Green conversion plan has not been defined.
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that
lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between
the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.
Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight
in homogeneous material)
(3)
MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4)
Multiple Top-Side Markings will be inside parentheses. Only one Top-Side Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a
continuation of the previous line and the two combined represent the entire Top-Side Marking for that device.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
Addendum-Page 1
Samples
PACKAGE MATERIALS INFORMATION
www.ti.com
8-May-2013
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device
Package Package Pins
Type Drawing
SPQ
Reel
Reel
A0
Diameter Width (mm)
(mm) W1 (mm)
B0
(mm)
K0
(mm)
P1
(mm)
W
Pin1
(mm) Quadrant
LM3405XMK/NOPB
SOT
DDC
6
1000
178.0
8.4
3.2
3.2
1.4
4.0
8.0
Q3
LM3405XMKX/NOPB
SOT
DDC
6
3000
178.0
8.4
3.2
3.2
1.4
4.0
8.0
Q3
Pack Materials-Page 1
PACKAGE MATERIALS INFORMATION
www.ti.com
8-May-2013
*All dimensions are nominal
Device
Package Type
Package Drawing
Pins
SPQ
Length (mm)
Width (mm)
Height (mm)
LM3405XMK/NOPB
SOT
DDC
6
1000
210.0
185.0
35.0
LM3405XMKX/NOPB
SOT
DDC
6
3000
210.0
185.0
35.0
Pack Materials-Page 2
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