LINER LT1512IS8 Sepic constant-current/ constant-voltage battery charger Datasheet

LT1512
SEPIC Constant-Current/
Constant-Voltage
Battery Charger
U
DESCRIPTION
FEATURES
■
■
■
■
■
■
■
The LT ®1512 is a 500kHz current mode switching regulator specially configured to create a constant-current/
constant-voltage battery charger. In addition to the usual
voltage feedback node, it has a current sense feedback
circuit for accurately controlling output current of a flyback
or SEPIC (Single-Ended Primary Inductance Converter)
topology charger. These topologies allow the current
sense circuit to be ground referred and completely separated from the battery itself, simplifying battery switching
and system grounding problems. In addition, these topologies allow charging even when the input voltage is
lower than the battery voltage.
Charger Input Voltage May Be Higher, Equal to or
Lower Than Battery Voltage
Charges Any Number of Cells Up to 30V*
1% Voltage Accuracy for Rechargeable Lithium
Batteries
100mV Current Sense Voltage for High Efficiency
Battery Can Be Directly Grounded
500kHz Switching Frequency Minimizes
Inductor Size
Charging Current Easily Programmable or Shut Down
U
APPLICATIONS
■
■
■
■
Maximum switch current on the LT1512 is 1.5A. This
allows battery charging currents up to 1A for a single
lithium-ion cell. Accuracy of 1% in constant-voltage mode
is perfect for lithium battery applications. Charging current can be easily programmed for all battery types.
Battery Charging of NiCd, NiMH, Lead-Acid
or Lithium Rechargeable Cells
Precision Current Limited Power Supply
Constant-Voltage/Constant-Current Supply
Transducer Excitation
, LTC and LT are registered trademarks of Linear Technology Corporation.
*Maximum Input Voltage = 40V – VBAT
U
TYPICAL APPLICATION
Maximum Charging Current
1.0
CHARGE
+
C3
22µF
25V
SYNC
AND/OR
SHUTDOWN
SHUTDOWN
SINGLE
LITHIUM
CELL (4.1V)
L1 A*
•
C2**
D1
2.2µF MBRS130LT3
VIN
VSW
0.5A
L1 B*
LT1512
S/S
FB
IFB
GND GND S VC
R1
•
R4
24Ω
R2
C5
0.1µF
R5
1k
0.8
CURRENT (A)
WALL
ADAPTER
INPUT
C4
0.22µF
+
C1
22µF
25V
Figure 1. SEPIC Charger with 0.5A Output Current
DOUBLE
LITHIUM
CELL (8.2V)
0.4
6V BATTERY
12V BATTERY
0.2
INDUCTOR = 33µH
R3
0.2Ω
*L1 A, L1 B ARE TWO 33µH WINDINGS ON A
SINGLE INDUCTOR: COILTRONICS CTX33-3
**TOKIN CERAMIC 1E225ZY5U-C203-F
0.6
0
0
5
15
10
INPUT VOLTAGE (V)
20
25
1512 TA02
1512 F01
ACTUAL PROGRAMMED CHARGING CURRENT WILL BE INDEPENDENT OF
INPUT VOLTAGE AND BATTERY VOLTAGE IF IT DOES NOT EXCEED THE
VALUES SHOWN. THESE ARE ELECTRICAL LIMITATIONS BASED ON MAXIMUM
SWITCH CURRENT. PACKAGE THERMAL LIMITATIONS MAY REDUCE
MAXIMUM CHARGING CURRENT. SEE APPLICATIONS INFORMATION.
1
LT1512
U
W W
W
Input Voltage .......................................................... 30V
Switch Voltage ........................................................ 40V
S/S Pin Voltage ....................................................... 30V
FB Pin Voltage (Transient, 10ms) ......................... ±10V
VFB Pin Current .................................................... 10mA
IFB Pin Voltage (Transient, 10ms) ......................... ±10V
Storage Temperature Range ................ – 65°C to 150°C
Ambient Temperature Range
LT1512C (Note 3) .................................... 0°C to 70°C
LT1512I .............................................. – 40°C to 85°C
Operating Junction Temperature Range
LT1512C (Note 3) ............................ – 20°C to 125°C
LT1512I ............................................ – 40°C to 125°C
Short Circuit ......................................... 0°C to 150°C
Lead Temperature (Soldering, 10 sec)................. 300°C
U
RATI GS
W
AXI U
U
ABSOLUTE
PACKAGE/ORDER I FOR ATIO
ORDER PART
NUMBER
TOP VIEW
VC 1
8
VSW
FB 2
7
GND
IFB 3
6
GND S
S/S 4
5
VIN
N8 PACKAGE
8-LEAD PDIP
LT1512CN8
LT1512CS8
LT1512IN8
LT1512IS8
S8 PACKAGE
8-LEAD PLASTIC SO
S8 PART
MARKING
TJMAX = 125°C, θJA = 100°C/ W (N)
TJMAX = 125°C, θJA = 130°C/ W (S)
1512
1512I
NOTE: CONTACT FACTORY CONCERNING 16-LEAD
FUSED-LEAD GN PACKAGE WITH LOWER THERMAL
RESISTANCE
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
VREF
VFB Reference Voltage
Measured at FB Pin
VC = 0.8V
1.233
1.228
1.245
1.245
1.257
1.262
V
V
300
550
600
nA
nA
0.01
0.03
%/V
FB Input Current
●
VFB = VREF
●
VIREF
FB Reference Voltage Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
IFB Reference Voltage
Measured at IFB Pin
VFB = 0V, VC = 0.8V
●
– 107
–110
– 100
– 100
– 93
– 90
mV
mV
VIFB = VIREF (Note 2)
●
10
25
35
µA
IFB Reference Voltage Line Regulation
2.7V ≤ VIN ≤ 25V, VC = 0.8V
●
0.01
0.05
%/V
Error Amplifier Transconductance
∆IC = ±25µA
1500
●
1100
700
1900
2300
µmho
µmho
120
200
350
µA
1400
2400
µA
1.95
0.40
2.30
0.52
V
V
IFB Input Current
gm
AV
f
Error Amplifier Source Current
VFB = VREF – 150mV, VC = 1.5V
●
Error Amplifier Sink Current
VFB = VREF + 150mV, VC = 1.5V
●
Error Amplifier Clamp Voltage
High Clamp, VFB = 1V
Low Clamp, VFB = 1.5V
1.70
0.25
VC Pin Threshold
Duty Cycle = 0%
0.8
1
1.25
V
Switching Frequency
2.7V ≤ VIN ≤ 25V
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ < 0°C (LT1512I)
●
450
430
400
500
500
550
580
580
kHz
kHz
kHz
●
88
95
●
40
35
Error Amplifier Voltage Gain
500
Maximum Switch Duty Cycle
Switch Current Limit Blanking Time
BV
2
Output Switch Breakdown Voltage
130
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ < 20°C (LT1512I)
47
V/ V
%
260
ns
V
V
LT1512
ELECTRICAL CHARACTERISTICS
VIN = 5V, VC = 0.6V, VFB = VREF, IFB = 0V, VSW and S/S pins open, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
VSAT
Output Switch ON Resistance
ISW = 2A
●
MIN
TYP
MAX
0.5
0.8
ILIM
Switch Current Limit
Duty Cycle = 50%
Duty Cycle = 80% (Note 1)
●
●
Ω
1.9
1.7
2.7
2.5
A
A
∆IIN
∆ISW
Supply Current Increase During Switch ON Time
15
25
mA/A
Control Voltage to Switch Current
Transconductance
2
Minimum Input Voltage
IQ
1.5
1.3
UNITS
A/V
●
2.4
2.7
V
Supply Current
2.7V ≤ VIN ≤ 25V
●
4
5.5
mA
Shutdown Supply Current
2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V
0°C ≤ TJ ≤ 125°C
– 40°C ≤ TJ ≤ 0°C (LT1512I)
●
12
30
50
µA
µA
2.7V ≤ VIN ≤ 25V
●
0.6
1.3
2
V
●
5
12
25
µs
●
– 10
15
µA
●
600
800
kHz
Shutdown Threshold
Shutdown Delay
0V ≤ VS/S ≤ 5V
S/S Pin Input Current
Synchronization Frequency Range
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: For duty cycles (DC) between 50% and 85%, minimum
guaranteed switch current is given by ILIM = 0.667 (2.75 – DC).
Note 2: The IFB pin is servoed to its regulating state with VC = 0.8V.
Note 3: Commercial devices are guaranteed over 0°C to 125°C junction
temperature range and 0°C to 70°C ambient temperature range. These
parts are also designed, characterized and expected to operate over the
– 20°C to 85°C extended ambient temperature range, but are not tested at
– 20°C or 85°C. Devices with full guaranteed electrical specifications over
the ambient temperature range – 40°C to 85°C are available as industrial
parts with an “I” suffix.
Maximum allowable ambient temperature may be limited by power
dissipation. Parts may not necessarily be operated simultaneously at
maximum power dissipation and maximum ambient temperature.
Temperature rise calculations must be done as shown in the Applications
Information section to ensure that maximum junction temperature does
not exceed 125°C limit. With high power dissipation, maximum ambient
temperature may be less than 70°C.
U W
TYPICAL PERFORMANCE CHARACTERISTICS
Switch Saturation Voltage
vs Switch Current
Switch Current Limit
vs Duty Cycle
3.0
150°C
100°C
0.9
25°C
SWITCH CURRENT LIMIT (A)
0.8
0.7
0.6
0.5
–55°C
0.4
0.3
0.2
3.0
2.5
2.8
25°C AND
125°C
2.0
–55°C
1.5
1.0
INPUT VOLTAGE (V)
1.0
SWITCH SATURATION VOLTAGE (V)
Minimum Input Voltage
vs Temperature
2.6
2.4
2.2
2.0
0.5
0.1
0
0
0
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
SWITCH CURRENT (A)
1512 G01
0
10 20 30 40 50 60 70 80 90 100
DUTY CYCLE (%)
1512 G02
1.8
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
1512 G03
3
LT1512
U W
TYPICAL PERFORMANCE CHARACTERISTICS
Feedback Input Current
vs Temperature
0
800
fSYNC = 700kHz
2.5
2.0
1.5
1.0
0.5
0
–50 –25
Negative Feedback Input Current
vs Temperature
700
NEGATIVE FEEDBACK INPUT CURRENT (µA)
3.0
FEEDBACK INPUT CURRENT (nA)
MINIMUM SYNCHRONIZATION VOLTAGE (VP-P)
Minimum Peak-to-Peak
Synchronization Voltage vs Temp
VFB = VREF
600
500
400
300
200
100
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
1512 G04
1512 G05
–10
–20
–30
–40
–50
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
1512 G06
U
U
U
PIN FUNCTIONS
VC: The compensation pin is primarily used for frequency
compensation, but it can also be used for soft starting and
current limiting. It is the output of the error amplifier and
the input of the current comparator. Peak switch current
increases from 0A to 1.8A as the VC voltage varies from 1V
to 1.9V. Current out of the VC pin is about 200µA when the
pin is externally clamped below the internal 1.9V clamp
level. Loop frequency compensation is performed with a
capacitor or series RC network from the VC pin directly to
the ground pin (avoid ground loops).
FB: The feedback pin is used for positive output voltage
sensing. This pin is the inverting input to the voltage error
amplifier. The R1/R2 voltage divider connected to FB
defines Li-Ion float voltage at full charge, or acts as a
voltage limiter for NiCd or NiMH applications. Input bias
current is typically 300nA, so divider current is normally
set to 100µA to swamp out any output voltage errors due
to bias current. The noninverting input of this amplifier is
tied internally to a 1.245V reference. The grounded end of
the output voltage divider should be connected directly to
the LT1512 ground pin (avoid ground loops).
IFB: The current feedback pin is used to sense charging
current. It is the input to a current sense amplifier that
controls charging current when the battery voltage is
below the programmed voltage. During constant-current
4
operation, the IFB pin regulates at – 100mV. Input resistance of this pin is 5kΩ, so filter resistance (R4, Figure 1)
should be less than 50Ω. The 24Ω, 0.22µF filter shown in
Figure 1 is used to convert the pulsating current in the
sense resistor to a smooth DC current feedback signal.
S/S: This pin can be used for shutdown and/or synchronization. It is logic level compatible, but can be tied to VIN if
desired. It defaults to a high ON state when floated. A logic
low state will shut down the charger to a micropower state.
Driving the S/S pin with a continuous logic signal of
600kHz to 800kHz will synchronize switching frequency to
the external signal. Shutdown is avoided in this mode with
an internal timer.
VIN: The input supply pin should be bypassed with a low
ESR capacitor located right next to the IC chip. The
grounded end of the capacitor must be connected directly
to the ground plane to which the GND pin is connected.
GND S, GND: The LT1512 uses separate ground pins for
switch current (GND) and the control circuitry (GND S).
This isolates the control ground from any induced voltage
created by fast switch currents. Both pins should be tied
directly to the ground plane, but the external control
circuit components such as the voltage divider, frequency
compensation network and IFB bypass capacitor should
LT1512
U
U
U
PIN FUNCTIONS
radiation and voltage spikes. In particular, the path in
Figure 1 which includes SW to C2, D1, C1 and around to
the LT1512 ground pin should be as short as possible to
minimize voltage spikes at switch turn-off.
be connected directly to the GND S pin or to the ground
plane close to the point where the GND S pin is connected.
VSW: The switch pin is the collector of the power switch,
carrying up to 1.5A of current with fast rise and fall times.
Keep the traces on this pin as short as possible to minimize
W
BLOCK DIAGRAM
VIN
SHUTDOWN
DELAY AND RESET
S/S
SYNC
SW
LOW DROPOUT
2.3V REG
500kHz
OSC
ANTI-SAT
LOGIC
DRIVER
SWITCH
+
IFBA
5k
–
IFB
COMP
62k
–
–
+
FB
1.245V
REF
+
EA
IA
AV ≈ 6
VC
0.08Ω
–
GND
1512 F02
GND S
Figure 2
U
OPERATION
The LT1512 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage or current. Referring to the
Block Diagram, the switch is turned “on” at the start of each
oscillator cycle. It is turned “off” when switch current
reaches a predetermined level. Control of output voltage
and current is obtained by using the output of a dual
feedback voltage sensing error amplifier to set switch
current trip level. This technique has the advantage of
simplified loop frequency compensation. A low dropout
internal regulator provides a 2.3V supply for all internal
circuitry on the LT1512. This low dropout design allows
input voltage to vary from 2.7V to 25V. A 500kHz oscillator
is the basic clock for all internal timing. It turns “on” the
output switch via the logic and driver circuitry. Special
adaptive antisat circuitry detects onset of saturation in the
power switch and adjusts driver current instantaneously to
limit switch saturation. This minimizes driver dissipation
and provides very rapid turn-off of the switch.
A unique error amplifier design has two inverting inputs
which allow for sensing both output voltage and current. A
1.245V bandgap reference biases the noninverting input.
The first inverting input of the error amplifier is brought out
for positive output voltage sensing. The second inverting
input is driven by a “current” amplifier which is sensing
output current via an external current sense resistor. The
5
LT1512
U
OPERATION
current amplifier is set to a fixed gain of – 12.5 which
provides a – 100mV current limit sense voltage.
The error signal developed at the amplifier output is
brought out externally and is used for frequency compensation. During normal regulator operation this pin sits at a
voltage between 1V (low output current) and 1.9V (high
output current). Switch duty cycle goes to zero if the VC pin
is pulled below the VC pin threshold, placing the LT1512 in
an idle mode.
U
W
U
U
APPLICATIONS INFORMATION
The LT1512 is an IC battery charger chip specifically optimized to use the SEPIC converter topology. The SEPIC
topology has unique advantages for battery charging. It will
operate with input voltages above, equal to or below the
battery voltage, has no path for battery discharge when
turned off and eliminates the snubber losses of flyback
designs. It also has a current sense point that is ground
referred and need not be connected directly to the battery.
The two inductors shown are actually just two identical
windings on one inductor core, although two separate
inductors can be used.
A current sense voltage is generated with respect to ground
across R3 in Figure 1. The average current through R3 is
always identical to the current delivered to the battery. The
LT1512 current limit loop will servo the voltage across R3 to
– 100mV when the battery voltage is below the voltage limit
set by the output divider R1/R2. Constant current charging
is therefore set at 100mV/R3. R4 and C4 filter the current
signal to deliver a smooth feedback voltage to the IFB pin. R1
and R2 form a divider for battery voltage sensing and set the
battery float voltage. The suggested value for R2 is 12.4k. R1
is calculated from:
R1 =
R2(VBAT – 1.245)
1.245 + R2(0.3µA)
VBAT = battery float voltage
0.3µA = typical FB pin bias current
A value of 12.4k for R2 sets divider current at 100µA. This is
a constant drain on the battery when power to the charger is
off. If this drain is too high, R2 can be increased to 41.2k,
reducing divider current to 30µA. This introduces an addi-
6
tional uncorrectable error to the constant voltage float mode
of about ±0.5% as calculated by:
VBAT Error =
±0.15µA(R1)(R2)
1.245(R1+ R2)
±0.15µA = expected variation in FB bias current around
the nominal 0.3µA typical value.
With R2 = 41.2k and R1 = 228k, (VBAT = 8.2V), the error due
to variations in bias current would be ±0.42%.
A second option is to disconnect the voltage divider with a
small NMOS transistor as shown in Figure 3. To ensure
adequate drive to the transistor (even when the VIN voltage is
at its lowest operating point of 2.4V), the FET gate is driven
wth a peak detected voltage via D2. Note that there are two
connections for D2. The L1 A connection must be used if the
voltage divider is set for less than 3.5V (fully charged
battery). Gate drive is equal to battery voltage plus input
voltage. The disadvantage of this connection is that Q1 will
still be “on” if the VIN voltage is active and the charger is shut
down via the S/S pin. The L1 B connection allows Q1 to turn
off when VIN is off or when shutdown is initiated, but the
reduced gate drive (=VBAT) is not adequate to ensure a Q1
on-state for fully charged battery voltages less than 3.5V. Do
not substitute for Q1 unless the new device has adequate
VGS maximum rating, especially if D2 is connected to L1A.
C6 filters the gate drive and R5 pulls the gate low when
switching stops.
Disconnecting the divider leaves only D1 diode leakage as a
battery drain. See Diode Selection for a discussion of diode
leakage.
LT1512
U
W
U
U
APPLICATIONS INFORMATION
CONNECT D2 ANODE HERE FOR FULLY
CHARGED BATTERY VOLTAGE LESS
THAN 3.5V. Q1 WILL NOT BE TURNED OFF
IN SHUTDOWN IF VIN IS PRESENT
•
CONNECT D2 ANODE HERE IF FULLY CHARGED
BATTERY VOLTAGE IS GREATER THAN 3.5V AND
Q1 MUST BE TURNED OFF IN SHUTDOWN WITH
VIN STILL ACTIVE
L1 A
D2
1N4148
C2
VIN
D1
VSW
R1
SHUTDOWN
S/S
R5
470k
C6
470pF
LT1512
L1 B
+
Q1
2N7002
GND
FB
R3
R2
1512 F03
Figure 3. Eliminating Divider Current
Maximum Input Voltage
Maximum input voltage for the circuit in Figure 1 is partly
determined by battery voltage. A SEPIC converter has a
maximum switch voltage equal to input voltage plus output
voltage. The LT1512 has a maximum input voltage of 30V
and a maximum switch voltage of 40V, so this limits
maximum input voltage to 30V, or 40V – VBAT, whichever
is less. Maximum VBAT = 40V – VIN.
Shutdown and Synchronization
The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulled high or left floating for normal operation. A logic low
on the S/S pin activates shutdown, reducing input supply
current to 12µA. To synchronize switching, drive the S/S pin
between 600kHz and 800kHz.
Inductor Selection
L1A and L1B are normally just two identical windings on one
core, although two separate inductors can be used. A typical
value is 33µH, which gives about 0.25A peak-to-peak inductor current. Lower values will give higher ripple current,
which reduces maximum charging current. 15µH can be
used if charging currents are at least 20% lower than the
values shown in the maximum charging current graph.
Higher inductance values give slightly higher maximum
charging current, but are larger and more expensive. A low
loss toroid core such as KoolMµ®, Molypermalloy or Metglas®
is recommended. Series resistance should be less than
0.1Ω for each winding. “Open core” inductors, such as rods
or barrels are not recommended because they generate
large magnetic fields which may interfere with other electronics close to the charger.
Input Capacitor
The SEPIC topology has relatively low input ripple current
compared to other topologies and higher harmonics are
especially low. RMS ripple current in the input capacitor is
less than 0.1A with L = 33µH and less than 0.2A with
L = 15µH. A low ESR 22µF, 25V solid tantalum capacitor
(AVX type TPS or Sprague type 593D) is adequate for most
applications with the following caveat. Solid tantalum capacitors can be destroyed with a very high turn-on surge
current such as would be generated if a low impedance input
source were “hot switched” to the charger input. If this
condition can occur, the input capacitor should have the
highest possible voltage rating, at least twice the surge input
voltage if possible. Consult with the capacitor manufacturer
before a final choice is made. A 2.2µF ceramic capacitor such
as the one used for the coupling capacitor can also be used.
These capacitors do not have a turn-on surge limitation. The
input capacitor must be connected directly to the VIN pin and
the ground plane close to the LT1512.
KoolMµ is a registered trademark of Magnetics, Inc.
Metglas is a registered trademark of AlliedSignal Inc.
7
LT1512
U
U
W
U
APPLICATIONS INFORMATION
Output Capacitor
It is assumed as a worst case that all the switching output
ripple current from the battery charger could flow in the
output capacitor. This is a desirable situation if it is necessary to have very low switching ripple current in the battery
itself. Ferrite beads or line chokes are often inserted in series
with the battery leads to eliminate high frequency currents
that could create EMI problems. This forces all the ripple
current into the output capacitor. Total RMS current into the
capacitor has a maximum value of about 0.5A, and this is
handled with a 22µF, 25V capacitor shown in Figure 1. This is
an AVX type TPS or Sprague type 593D surface mount solid
tantalum unit intended for switching applications. Do not
substitute other types without ensuring that they have
adequate ripple current ratings. See Input Capacitor section
for details of surge limitation on solid tantalum capacitors if
the battery may be “hot switched” to the output of the
charger.
Coupling Capacitor
C2 in Figure 1 is the coupling capacitor that allows a SEPIC
converter topology to work with input voltages either higher
or lower than the battery voltage. DC bias on the capacitor is
equal to input voltage. RMS ripple current in the coupling
capacitor has a maximum value of about 0.5A at full charging current. A conservative formula to calculate this is:
I
(V + V )(1.1)
ICOUP(RMS) = CHRG IN BAT
2(VIN )
(1.1 is a fudge factor to account for inductor ripple current
and other losses)
With ICHRG = 0.5A, VIN = 15V and VBAT = 8.2V, ICOUP = 0.43A
The recommended capacitor is a 2.2µF ceramic type from
Marcon or Tokin. These capacitors have extremely low ESR
and high ripple current ratings in a small package. Solid
tantalum units can be substituted if their ripple current rating
is adequate, but typical values will increase to 22µF or more
to meet the ripple current requirements.
Diode Selection
The switching diode should be a Schottky type to minimize
both forward and reverse recovery losses. Average diode
current is the same as output charging current , so this will
be under 1A. A 1A diode is recommended for most applications, although smaller devices could be used at reduced
GND
VIN
1
4
+VIN
R4
R1
L1A
2 WINDING
INDUCTOR
4
L1B
3
1
L1A
2
L1B
R3
C4 R2
2
3
D1
D1
C3
VBATT
C2B C2A
S/S
+
a. Double-Sided (Vias Connect to the Backside of Ground Plane.
Dash Lines Indicate Interconnects on Backside. Demo Board
Uses This Layout, Except that R5 Has Been Added to Increase
Phase Margin)
R5
R3
VIN
S/S
GND
C5
R1 R2
IFB GND S
VBATT
VC VSW
GND
C1
1512 F04a
R4
C4
S/S
1512 F04b
b. Single-Sided Altenative Layout
Figure 4. LT1512 Suggested Layouts for Critical Thermal and Electrical Paths
8
+
C3
C5
U1
GND
C1
FB
R5
C2
LT1512
U
W
U
U
APPLICATIONS INFORMATION
charging current. Maximum diode reverse voltage will be
equal to input voltage plus battery voltage.
Diode reverse leakage current will be of some concern
during charger shutdown. This leakage current is a direct
drain on the battery when the charger is not powered. High
current Schottky diodes have relatively high leakage currents (2µA to 200µA) even at room temperature. The latest
very-low-forward devices have especially high leakage currents. It has been noted that surface mount versions of some
Schottky diodes have as much as ten times the leakage of
their through-hole counterparts. This may be because a low
forward voltage process is used to reduce power dissipation
in the surface mount package. In any case, check leakage
specifications carefully before making a final choice for the
switching diode. Be aware that diode manufacturers want to
specify a maximum leakage current that is ten times higher
than the typical leakage. It is very difficult to get them to
specify a low leakage current in high volume production.
This is an on going problem for all battery charger circuits
and most customers have to settle for a diode whose typical
leakage is adequate, but theoretically has a worst-case
condition of higher than desired battery drain.
Thermal Considerations
Care should be taken to ensure that worst-case conditions
do not cause excessive die temperatures. Typical thermal
resistance is 130°C/W for the S8 package but this number
will vary depending on the mounting technique (copper
area, air flow, etc).
Average supply current (including driver current) is:
IIN = 4mA +
(VBAT )(ICHRG )(0.024)
VIN
For VIN = 10V, VBAT = 8.2V, ICHRG = 0.5A, RSW = 0.65Ω
IIN = 4mA + 10mA = 14mA
PSW = 0.24W
PD = (0.014)(10) + 0.24 = 0.38W
The S8 package has a thermal resistance of 130°C/W.
(Contact factory concerning 16-lead fused-lead package
with footprint approximately same as S8 package and with
lower thermal resistance.) Die temperature rise will be
(0.38W)(130°C/W) = 49°C. A maximum ambient temperature of 60°C will give a die temperature of 60°C + 49°C =
109°C. This is only slightly less than the maximum junction
temperature of 125°C, illustrating the importance of doing
these calculations!
Programmed Charging Current
LT1512 charging current can be programmed with a PWM
signal from a processor as shown in Figure 5. C6 and D2
form a peak detector that converts a positive logic signal to a
negative signal. The average negative signal at the input to
R5 is equal to the processor VCC level multiplied by the
inverse PWM ratio. This assumes that the PWM signal is a
CMOS output that swings rail-to-rail with a source resistance less than a few hundred ohms. The negative voltage is
converted to a current by R5 and R6 and filtered by C7. This
current multiplied by R4 generates a voltage that subtracts
from the 100mV sense voltage of the LT1512. This is not a
high precision technique because of the errors in VCC and
the diode voltage, but it can typically be used to adjust
charging current over a 20% to 100% range with good
repeatability (full charging current accuracy is not affected).
To reduce the load on the logic signal, R4 has been increased
Switch power dissipation is given by:
(I
)2 (RSW )(VBAT + VIN )(VBAT)
PSW = CHRG
(VIN )2
RSW = output switch ON resistance
Total power dissipation of the die is equal to supply current
times supply voltage, plus switch power:
PD(TOTAL) = (IIN)(VIN) + PSW
LT1512
IFB
PWM
INPUT
≥1kHz
+
C6
1µF
R5
4.02k
D2
R6
4.02k
+
C7
10µF
L1B
R4
200Ω
C4
0.22µF
R3
1512 F05
Figure 5. Programming Charge Current
9
LT1512
U
U
W
U
APPLICATIONS INFORMATION
from 24Ω to 200Ω. This causes a known increase in fullscale charging current (PWM = 0) of 3% due to the 5k input
resistance of the IFB pin. Note that 100% duty cycle gives full
charging current and that very low duty cycles (especially
zero!) will not operate correctly. Very low duty cycle (<10%)
is a problem because the peak detector requires a finite
up-time to reset C6.
U
PACKAGE DESCRIPTION
More Help
Linear Technology Field Application Engineers have a CAD
spreadsheet program for detailed calculations of circuit
operating conditions, and our Applications Department is
always ready to lend a helping hand. For additional information refer to the LT1372 data sheet. This part is identical to
the LT1512 except for the current amplifier circuitry.
Dimensions in inches (millimeters) unless otherwise noted.
N8 Package
8-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.400*
(10.160)
MAX
8
7
6
5
1
2
3
4
0.255 ± 0.015*
(6.477 ± 0.381)
0.300 – 0.325
(7.620 – 8.255)
0.009 – 0.015
(0.229 – 0.381)
(
+0.025
0.325 –0.015
8.255
+0.635
–0.381
)
0.045 – 0.065
(1.143 – 1.651)
0.065
(1.651)
TYP
0.005
(0.127)
MIN
0.100 ± 0.010
(2.540 ± 0.254)
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
10
0.130 ± 0.005
(3.302 ± 0.127)
0.125
(3.175)
MIN
0.018 ± 0.003
(0.457 ± 0.076)
0.015
(0.380)
MIN
N8 0695
LT1512
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
S8 Package
8-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.189 – 0.197*
(4.801 – 5.004)
8
7
6
5
0.150 – 0.157**
(3.810 – 3.988)
0.228 – 0.244
(5.791 – 6.197)
1
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
2
3
4
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
0°– 8° TYP
0.016 – 0.050
0.406 – 1.270
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
0.014 – 0.019
(0.355 – 0.483)
0.050
(1.270)
BSC
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
SO8 0695
11
LT1512
U
TYPICAL APPLICATION
The circuit in Figure 6 will provide adapter current limiting
to ensure that the battery charger never overloads the
adapter. In addition, it adjusts charging current to a lower
value if other system power increases to the point where
the adapter would be overloaded. This allows the LT1512
WALL
ADAPTER
INPUT
SYSTEM
POWER
R6
0.2Ω
+
2
3
–
C2**
D1
2.2µF MBRS130LT3
5
1
LM301
LT1512
SYNC
4
AND/OR
S/S
SHUTDOWN
GND GND S VC
VSW
8
0.5A
L1 B*
30pF
8
6
C3
22µF
25V
VIN
+
4
Q1
2N3904
L1 A*
•
R5
1k
7
to charge the battery at the maximum possible rate without
concern about varying system power levels. The LM301 op
amp used here is unusual in that it can operate with its
inputs at a voltage equal to the positive supply voltage.
D2
1N4148
7
6
FB
IFB
1
C5
0.1µF
R1
•
R4
24Ω
3
TO FB PIN
R7
12k
2
C4
0.22µF
R2
+
C1
22µF
25V
R3
0.2Ω
*L1 A, L1 B ARE TWO 33µH WINDINGS ON A
COMMON CORE: COILTRONICS CTX33-3
**TOKIN CERAMIC 1E225ZY5U-C203-F
1512 F06
Figure 6. Adding Adapter Current Limiting
RELATED PARTS
PART NUMBER DESCRIPTION
COMMENTS
LT1239
Backup Battery Management System
Charges Backup Battery and Regulates Backup Battery Output when
Main Battery Removed
LTC®1325
Microprocessor Controlled Battery Management System
Can Charge, Discharge and Gas Gauge NiCd, NiMH and Pb-Acid Batteries
with Software Charging Profiles
LT1510
1.5A Constant-Current/Constant-Voltage Battery Charger
Step-Down Charger for Li-Ion, NiCd and NiMH
LT1511
3.0A Constant-Current/Constant-Voltage Battery Charger
with Input Current Limiting
Step-Down Charger that Allows Charging During Computer Operation and
Prevents Wall-Adapter Overload
LT1513
SEPIC Constant-Current/Constant-Voltage Battery Charger Step-Up/Step-Down Charger for Up to 2A Current
12
Linear Technology Corporation
LT/GP 1096 7K • PRINTED IN THE USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● TELEX: 499-3977
 LINEAR TECHNOLOGY CORPORATION 1996
Similar pages