LINER LT3652HV High voltage high current controller for battery charging and power management Datasheet

LTC4000
High Voltage High Current
Controller for Battery Charging
and Power Management
Description
Features
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Complete High Performance Battery Charger When
Paired with a DC/DC Converter
Wide Input and Output Voltage Range: 3V to 60V
Input Ideal Diode for Low Loss Reverse Blocking
and Load Sharing
Output Ideal Diode for Low Loss PowerPath™ and
Load Sharing with the Battery
Instant-On Operation with Heavily Discharged
Battery
Programmable Input and Charge Current:
±1% Accuracy
±0.25% Accurate Programmable Float Voltage
Programmable C/X or Timer Based Charge
Termination
NTC Input for Temperature Qualified Charging
28-Lead 4mm × 5mm QFN or SSOP Packages
The LTC®4000 is a high voltage, high performance controller
that converts many externally compensated DC/DC power
supplies into full-featured battery chargers.
Features of the LTC4000’s battery charger include: accurate
(±0.25%) programmable float voltage, selectable timer or
current termination, temperature qualified charging using
an NTC thermistor, automatic recharge, C/10 trickle charge
for deeply discharged cells, bad battery detection and
status indicator outputs. The battery charger also includes
precision current sensing that allows lower sense voltages
for high current applications.
The LTC4000 supports intelligent PowerPath control. An
external PFET provides low loss reverse current protection. Another external PFET provides low loss charging
or discharging of the battery. This second PFET also
facilitates an instant-on feature that provides immediate
downstream system power even when connected to a
heavily discharged or short faulted battery.
Applications
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High Power Battery Charger Systems
High Performance Portable Instruments
Industrial Battery Equipped Devices
Notebook/Subnotebook Computers
The LTC4000 is available in a low profile 28-lead 4mm ×
5mm QFN and SSOP packages.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks and
PowerPath is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
Typical Application
48V to 10.8V at 10A Buck Converter Charger for Three LiFePO4 Cells
IN
5mΩ
LT3845
OUT
VC
SHDN
14.7k
ITH
47nF
CC
IID IGATE CSP
BAT
127k
LTC4000
ENC
CHRG
FLT
FBG
133k
BFB
IIMON
10nF
1.13M
NTC
IBMON
10nF
TMR
CL
0.1µF
24.9k
10k
GND BIAS
CX
22.1k
12
11
ICHARGE
10
8
Si7135DP
OFB
VM
3.0V
5mΩ
CSN
BGATE
1µF
1.10M
100k
1.15M
10k
1µF
10.8V FLOAT
10A MAX CHARGE
CURRENT
3-CELL LiFePO4
BATTERY PACK
VOUT
10.5
10
VOUT
6
9.5
4
9
2
0
8.5
ICHARGE
6
7
VOUT (V)
RST
CLN
IN
12V, 15A
100µF
Charge Current and VOUT Profile
vs VBAT During a Charge Cycle
ICHARGE (A)
15V TO 60V
Si7135DP
8
10
9
VBAT (V)
11
8
12
4000 TA01b
4000 TA01a
NTHS0603
N02N1002J
4000f
1
LTC4000
Absolute Maximum Ratings
(Note 1)
IN, CLN, IID, CSP, CSN, BAT........................ –0.3V to 62V
IN-CLN, CSP-CSN.............................................–1V to 1V
OFB, BFB, FBG............................................ –0.3V to 62V
FBG.............................................................–1mA to 2mA
IGATE............Max (VIID, VCSP) – 10V to Max (VIID, VCSP)
BGATE........Max (VBAT, VCSN) – 10V to Max (VBAT, VCSN)
ENC, CX, NTC, VM....................................–0.3V to VBIAS
IL, CL, TMR, IIMON, CC............................–0.3V to VBIAS
BIAS..............................................–0.3V to Min (6V, VIN)
IBMON...................................–0.3V to Min (VBIAS, VCSP)
ITH................................................................ –0.3V to 6V
CHRG, FLT, RST........................................... –0.3V to 62V
CHRG, FLT, RST...........................................–1mA to 2mA
Operating Junction Temperature Range
(Note 2).................................................................. 125°C
Storage Temperature Range................... –65°C to 150°C
Pin Configuration
TOP VIEW
3
26 RST
CL
4
25 VM
TMR
5
24 GND
GND
6
23 IN
ITH
CX
CC
27 IIMON
28 27 26 25 24 23
CLN
28 IL
2
IN
1
IBMON
GND
ENC
IID
TOP VIEW
22 IGATE
VM 1
RST 2
21 OFB
IIMON 3
20 CSP
IL 4
19 CSN
29
GND
ENC 5
18 BGATE
IBMON 6
17 BAT
CX 7
16 BFB
CL 8
15 FBG
NTC
BIAS
CHRG
FLT
GND
TMR
9 10 11 12 13 14
UFD PACKAGE
28-LEAD (4mm × 5mm) PLASTIC QFN
TJMAX = 125°C, θJA = 43°C/W, θJC = 4°C/W
EXPOSED PAD (PIN 29) IS GND, MUST BE SOLDERED TO PCB
FLT
7
22 CLN
CHRG
8
21 CC
BIAS
9
20 ITH
NTC 10
19 IID
FBG 11
18 IGATE
BFB 12
17 OFB
BAT 13
16 CSP
BGATE 14
15 CSN
GN PACKAGE
28-LEAD PLASTIC SSOP
TJMAX = 125°C, θJA = 80°C/W, θJC = 25°C/W
Order Information
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC4000EUFD#PBF
LTC4000EUFD#TRPBF
4000
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 125°C
LTC4000IUFD#PBF
LTC4000IUFD#TRPBF
4000
28-Lead (4mm × 5mm) Plastic QFN
–40°C to 125°C
LTC4000EGN#PBF
LTC4000EGN#TRPBF
LTC4000GN
28-Lead Plastic SSOP
–40°C to 125°C
LTC4000IGN#PBF
LTC4000IGN#TRPBF
LTC4000GN
28-Lead Plastic SSOP
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
4000f
2
LTC4000
Electrical Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = VCLN = 3V to 60V unless otherwise noted (Notes 2, 3).
SYMBOL
PARAMETER
VIN
Input Supply Operating Range
IIN
Input Quiescent Operating Current
IBAT
Battery Pin Operating Current
Battery Only Quiescent Current
CONDITIONS
MIN
l
TYP
3
MAX
60
0.4
UNITS
V
mA
VIN ≥ 3V, VCSN = VCSP ≥ VBAT
l
50
100
µA
VIN ≥ 0V, VCSN = VCSP ≥ VBAT
l
10
20
μA
0.4
V
Shutdown
ENC Input Voltage Low
l
ENC Input Voltage High
l
1.5
–4
–2
l
1.5
2.5
l
1.133
1.125
1.136
1.136
l
1.181
1.193
ENC Pull-Up Current
VENC = 0V
ENC Open Circuit Voltage
VENC = Open
V
–0.5
µA
V
Voltage Regulation
VBFB_REG
Battery Feedback Voltage
BFB Input Current
VOFB_REG
VBFB = 1.2V
Output Feedback Voltage
OFB Input Current
± 0.1
VOFB = 1.2V
RFBG
Ground Return Feedback Resistance
VRECHRG(RISE)
Rising Recharge Battery Threshold Voltage
VRECHRG(HYS)
Recharge Battery Threshold Voltage Hysteresis % of VBFB_REG
VOUT(INST_ON)
Instant-On Battery Voltage Threshold
VLOBAT
VLOBAT(HYS)
V
V
µA
1.204
± 0.1
l
% of VBFB_REG
1.139
1.147
V
µA
100
400
Ω
97.6
98.3
%
l
96.9
% of VBFB_REG
l
82
86
90
%
Falling Low Battery Threshold Voltage
% of VBFB_REG
l
65
68
71
%
Low Battery Threshold Voltage Hysteresis
% of VBFB_REG
Ratio of Monitored-Current Voltage to Sense
Voltage
VIN,CLN = 50mV, VIIMON/VIN,CLN
VCSP,CSN = 50mV, VIBMON/VCSP,CSN
Sense Voltage Offset
VCSP,CSN = 50mV, VCSP = 60V or
VIN,CLN = 50mV, VIN = 60V (Note 4)
0.5
%
3
%
Current Regulation
VOS
l
19
20
–300
CLN Pin Current
21
V/V
300
µV
±1
µA
CSP Pin Current
VIGATE = Open, VIID = 0V
90
μA
CSN Pin Current
VBGATE = Open, VBAT = 0V
45
μA
IIL
Pull-Up Current for the Input Current Limit
Programming Pin
l
–55
–50
–45
μA
ICL
Pull-Up Current for the Charge Current Limit
Programming Pin
l
–55
–50
–45
μA
ICL_TRKL
Pull-Up Current for the Charge Current Limit
Programming Pin in Trickle Charge Mode
l
–5.5
–5.0
–4.5
μA
40
90
140
kΩ
VBFB < VLOBAT
Input Current Monitor Resistance to GND
40
90
140
kΩ
A4, A5 Error Amp Offset for the Current Loops VCL = 0.8V, VIL = 0.8V
(See Figure 1)
Charge Current Monitor Resistance to GND
l
–10
0
10
mV
Maximum Programmable Current Limit
Voltage Range
l
0.985
1.0
1.015
V
4000f
3
LTC4000
Electrical Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = VCLN = 3V to 60V unless otherwise noted (Notes 2, 3).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Charge Termination
CX Pin Pull-Up Current
VCX = 0.1V
l
–5.5
–5.0
–4.5
µA
VCX,IBMON(OS)
CX Comparator Offset Voltage, IBMON Falling
VCX = 0.1V
l
0.5
10
25
mV
VCX,IBMON(HYS)
CX Comparator Hysteresis Voltage
5
mV
TMR Pull-Up Current
VTMR = 0V
–5.0
μA
TMR Pull-Down Current
VTMR = 1.5V
5.0
μA
TMR Pin Frequency
CTMR = 0.01μF
400
500
600
Hz
2.1
2.5
V
tT
Charge Termination Time
CTMR = 0.1μF
l
2.3
2.9
3.5
h
tT/tBB
Ratio of Charge Terminate Time to Bad Battery CTMR = 0.1μF
Indicator Time
l
3.95
4
4.05
h/h
VNTC(COLD)
NTC Cold Threshold
VNTC Rising, % of VBIAS
l
73
75
77
%
VNTC(HOT)
NTC Hot Threshold
VNTC Falling, % of VBIAS
l
33
35
37
%
VNTC(HYS)
NTC Thresholds Hysteresis
% of VBIAS
VNTC(OPEN)
NTC Open Circuit Voltage
% of VBIAS
RNTC(OPEN)
NTC Open Circuit Input Resistance
TMR Threshold for CX Termination
l
5
l
45
50
%
55
300
%
kΩ
Voltage Monitoring and Open Drain Status Pins
VVM(TH)
VM Input Falling Threshold
VVM(HYS)
VM Input Hysteresis
VM Input Current
l
1.181
VVM = 1.2V
1.193
1.204
V
40
mV
±0.1
µA
IRST,CHRG,FLT(LKG) Open Drain Status Pins Leakage Current
VPIN = 60V
VRST,CHRG,FLT(VOL) Open Drain Status Pins Voltage Output Low
IPIN = 1mA
l
±1
µA
Input PowerPath Forward Regulation Voltage
VIID,CSP, 3V ≤ VCSP ≤ 60V
l
0.1
Input PowerPath Fast Reverse Turn-Off
Threshold Voltage
VIID,CSP, 3V ≤ VCSP ≤ 60V,
VIGATE = VCSP – 2.5V,
∆IIGATE/∆ VIID,CSP ≥ 100μA/mV
l
Input PowerPath Fast Forward Turn-On
Threshold Voltage
VIID,CSP, 3V ≤ VCSP ≤ 60V,
VIGATE = VIID – 1.5V,
∆IIGATE/∆ VIID,CSP ≥ 100μA/mV
l
Input Gate Turn-Off Current
VIID = VCSP, VIGATE = VCSP – 1.5V
–0.3
μA
Input Gate Turn-On Current
VCSP = VIID – 20mV,
VIGATE = VIID – 1.5V
0.3
μA
IIGATE(FASTOFF)
Input Gate Fast Turn-Off Current
VCSP = VIID + 0.1V,
VIGATE = VCSP – 5V
–0.5
mA
IIGATE(FASTON)
Input Gate Fast Turn-On Current
VCSP = VIID – 0.1V,
VIGATE = VIID – 1.5V
0.7
mA
VIGATE(ON)
Input Gate Clamp Voltage
IIGATE = 2µA, VIID = 12V to 60V,
VCSP = VIID – 0.5V, Measure
VIID – VIGATE
l
13
15
V
Input Gate Off Voltage
IIGATE = – 2μA, VIID = 3V to 59.9V,
VCSP = VIID + 0.5V, Measure
VCSP – VIGATE
l
0.45
0.7
V
0.4
V
8
20
mV
–90
–50
–20
mV
40
80
130
mV
Input PowerPath Control
4000f
4
LTC4000
Electrical Characteristics
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C. VIN = VCLN = 3V to 60V unless otherwise noted (Notes 2, 3).
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Battery PowerPath Control
Battery Discharge PowerPath Forward
Regulation Voltage
VBAT,CSN, 2.8V ≤ VBAT ≤ 60V
l
0.1
8
20
mV
Battery PowerPath Fast Reverse Turn-Off
Threshold Voltage
VBAT,CSN, 2.8V ≤ VBAT ≤ 60V, Not
Charging, VBGATE = VCSN – 2.5V,
∆IBGATE/∆VBAT,CSN ≥ 100μA/mV
l
–90
–50
–20
mV
Battery PowerPath Fast Forward Turn-On
Threshold Voltage
VBAT,CSN, 2.8V ≤ VCSN ≤ 60V,
VBGATE = VBAT – 1.5V,
∆IBGATE/∆ VBAT,CSN ≥ 100μA/mV
l
40
80
130
mV
Battery Gate Turn-Off Current
VBGATE = VCSN – 1.5V, VCSN ≥ VBAT,
VOFB < VOUT(INST_ON) and Charging
in Progress, or VCSN = VBAT and Not
Charging
–0.3
μA
Battery Gate Turn-On Current
VBGATE = VBAT – 1.5V, VCSN ≥ VBAT,
VOFB > VOUT(INST_ON) and Charging in
Progress, or VCSN = VBAT – 20mV
0.3
μA
IBGATE(FASTOFF)
Battery Gate Fast Turn-Off Current
VCSN = VBAT + 0.1V and Not
Charging, VBGATE = VCSN – 5V
–0.5
mA
IBGATE(FASTON)
Battery Gate Fast Turn-On Current
VCSN = VBAT – 0.1V,
VBGATE = VBAT – 1.5V
0.7
mA
VBGATE(ON)
Battery Gate Clamp Voltage
IBGATE = 2μA, VBAT = 12V to 60V,
VCSN = VBAT – 0.5V, Measure
VBAT – VBGATE
Battery Gate Off Voltage
l
IBGATE = – 2μA, VBAT = 2.8V to 60V,
VCSN = VBAT + 0.5V and not Charging,
Measure VCSN – VBGATE
l
13
15
V
0.45
0.7
V
2.9
3.5
V
–10
BIAS Regulator Output and Control Pins
VBIAS
BIAS Output Voltage
No Load
ΔVBIAS
BIAS Output Voltage Load Regulation
IBIAS = – 0.5mA
–0.5
BIAS Output Short-Circuit Current
VBIAS = 0V
–12
mA
Transconductance of Error Amp
CC = 1V
0.5
mA/V
l
Open Loop DC Voltage Gain of Error Amp
CC = Open
IITH(PULL_UP)
Pull-Up Current on the ITH Pin
VITH = 0V, VCC = 0V
IITH(PULL_DOWN)
Pull-Down Current on the ITH Pin
VITH = 0.4V, CC = Open
Open Loop DC Voltage Gain of ITH Driver
ITH = Open
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC4000 is tested under conditions such that TJ ≈ TA. The
LTC4000E is guaranteed to meet specifications from 0°C to 85°C junction
temperature. Specifications over the –40°C to 125°C operating junction
temperature range are assured by design, characterization and correlation
with statistical process controls. The LTC4000I is guaranteed over the
full –40°C to 125°C operating junction temperature range. Note that the
maximum ambient temperature consistent with these specifications is
2.4
80
–6
l
0.5
–5
%
dB
–4
μA
1
mA
60
dB
determined by specific operating conditions in conjunction with board
layout, the rated package thermal impedance and other environmental
factors. The junction temperature (TJ, in °C) is calculated from the ambient
temperature (TA, in °C) and power dissipation (PD, in Watts) according to
the following formula:
TJ = TA + (PD • θJA), where θJA (in °C/W) is the package thermal
impedance.
Note 3: All currents into pins are positive; all voltages are referenced to
GND unless otherwise noted.
Note 4: These parameters are guaranteed by design and are not 100%
tested.
4000f
5
LTC4000
Typical Performance Characteristics
Input Quiescent Current and
Battery Quiescent Current Over
Temperature
1.0
Battery Float Voltage Feedback, Output
Voltage Regulation Feedback and VM
Falling Threshold Over Temperature
Battery Only Quiescent Current
Over Temperature
100
VIN = VBAT = 15V
VCSN = 15.5V
IIN
1.20
1.19
VBAT = 60V
10
1.18
IBAT
PIN VOLTAGE (V)
0.1
1
IBAT (µA)
IIN/IBAT (mA)
VBAT = 15V
VBAT = 3V
0.1
0.01
1.17
VVM(TH)
VOFB_REG
1.16
1.15
1.14
VBFB_REG
1.13
1.12
1.11
0
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
0.001
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G01
4000 G02
Battery Thresholds: Rising Recharge,
Instant-On Regulation and Falling Low
Battery As a Percentage of Battery
Float Feedback Over Temperature
100
–45.0
1.015
1.010
–47.5
IIL/ICL (µA)
VOUT(INST_ON)
80
75
70
Maximum Programmable Current
Limit Voltage Over Temperature
IL and CL Pull-Up Current Over
Temperature
VRECHRG(RISE)
90
85
–50.0
VLOBAT
–55.0
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G04
0.985
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G06
CX Comparator Offset Voltage with
VIBMON Falling Over Temperature
Current Sense Offset Voltage Over
Common Mode Voltage Range
300
VMAX(IN,CLN) = VMAX(CSP, CSN) = 15V
200
100
VOS (µV)
VOS(CSP, CSN)
0
VOS(IN, CSN)
VOS(CSP, CSN)
0
VOS(IN, CSN)
–100
–200
–200
–300
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
–300
4000 G07
VCX,IBMON (mV)
200
VOS (µV)
0.995
4000 G05
Current Sense Offset Voltage
Over Temperature
–100
1.000
0.990
60
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
100
1.005
–52.5
65
300
4000 G03
VIIMON /VIBMON (V)
PERCENT OF VBFB_REG (%)
95
1.10
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
0
10
20
30
40
50
VMAX(IN, CLN) /VMAX(CSP, CSN) (V)
60
4000 G08
20
19
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G09
4000f
6
LTC4000
Typical Performance Characteristics
Charge Termination Time with 0.1µF
Timer Capacitor Over Temperature
3.5
80
VNTC(COLD)
75
3.3
14
12
2.9
2.7
65
60
55
VNTC(OPEN)
50
45
VNTC(HOT)
40
2.5
VIID,CSP / VBAT,CSN (mV)
PERCENT OF VBIAS (%)
70
3.1
TT (h)
PowerPath Forward Voltage
Regulation Over Temperature
NTC Thresholds Over
Temperature
VIGATE (ON)/VBGATE(ON) (V)
0
–30
14.0
13.5
13.0
12.5
12.0
–60
11.5
–90
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
11.0
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G13
4000 G14
BIAS Voltage at 0.5mA Load Over
Temperature
1.5
1.4
3.1
VIN = 3V
2.7
450
400
350
300
250
200
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
ITH Pull-Down Current
vs VITH
2.0
1.2
1.1
1.0
0.9
0.8
0.7
2.6
500
2.5
VITH = 0.4V
1.3
IITH(PULL-DOWN) (mA)
VBIAS (V)
VIN = 15V
VCSP = VCSN = 15V
550
ITH Pull-Down Current Over
Temperature
3.2
VIN = 60V
600
4000 G15
IITH(PULL-DOWN) (mA)
VIID,CSP /VBAT,CSN (mV)
VIID = VBAT = 15V
14.5
30
2.8
PowerPath Turn-Off Gate Voltage
Over Temperature
VMAX(IID,CSP), IGATE/VMAX(BAT,CSN),BGATE (mV)
15.0
VIID = VBAT = 15V
2.9
4000 G12
PowerPath Turn-On Gate Clamp
Voltage Over Temperature
90
3.0
4
4000 G11
PowerPath Fast Off, Fast On
and Forward Regulation Over
Temperature
60
VIID = VBAT = 60V
6
0
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
30
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G10
120
VIID = VBAT = 3V
8
2
35
2.3
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
VIID = VBAT = 15V
10
1.5
1.0
0.5
0.6
2.5
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G16
0.5
–60 –40 –20 0 20 40 60 80 100 120 140
TEMPERATURE (°C)
4000 G17
0
0
0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
VITH (V)
1
4000 G18
4000f
7
LTC4000
Pin Functions
(QFN/SSOP)
VM (Pin 1/Pin 25): Voltage Monitor Input. High impedance
input to an accurate comparator with a 1.193V threshold
(typical). This pin controls the state of the RST output
pin. Connect a resistor divider (RVM1, RVM2) between the
monitored voltage and GND, with the center tap point connected to this pin. The falling threshold of the monitored
voltage is calculated as follows:
VVM _ RST =
RVM1 + RVM2
• 1.193V
RVM2
where RVM2 is the bottom resistor between the VM pin
and GND. Tie to the BIAS pin if voltage monitoring function is not used.
RST (Pin 2/Pin 26): High Voltage Open Drain Reset Output.
When the voltage at the VM pin is below 1.193V, this status
pin is pulled low. When driven low, this pin can disable a
DC/DC converter when connected to the converter’s enable pin. This pin can also drive an LED to provide a visual
status indicator of a monitored voltage. Short this pin to
GND when not used.
IIMON (Pin 3/Pin 27): Input Current Monitor. The voltage
on this pin is 20 times (typical) the sense voltage (VIN,CLN)
across the input current sense resistor(RIS), therefore
providing a voltage proportional to the input current.
Connect an appropriate capacitor to this pin to obtain a
voltage representation of the time-average input current.
Short this pin to GND to disable input current limit feature.
IL (Pin 4/Pin 28): Input Current Limit Programming. Connect the input current programming resistor (RIL) to this
pin. This pin sources 50µA of current. The regulation loop
compares the voltage on this pin with the input current
monitor voltage (VIIMON), and drives the ITH pin accordingly to ensure that the programmed input current limit is
not exceeded. The input current limit is determined using
the following formula:
ENC (Pin 5/Pin 1): Enable Charging Pin. High impedance
digital input pin. Pull this pin above 1.5V to enable charging and below 0.5V to disable charging. Leaving this pin
open causes the internal 1µA pull-up current to pull the
pin to 2.5V (typical).
IBMON (Pin 6/Pin 2): Battery Charge Current Monitor. The
voltage on this pin is 20 times (typical) the sense voltage
(VCSP,CSN) across the battery current sense resistor (RCS),
therefore providing a voltage proportional to the battery
charge current. Connect an appropriate capacitor to this
pin to obtain a voltage representation of the time-average
battery charge current. Short this pin to GND to disable
charge current limit feature.
CX (Pin 7/Pin 3): Charge Current Termination Programming. Connect the charge current termination programming resistor (RCX) to this pin. This pin is a high
impedance input to a comparator and sources 5μA of
current. When the voltage on this pin is greater than the
charge current monitor voltage (VIBMON), the CHRG pin
turns high impedance indicating that the CX threshold is
reached. When this occurs, the charge current is immediately terminated if the TMR pin is shorted to the BIAS
pin, otherwise charging continues until the charge termination timer expires. The charge current termination value
is determined using the following formula:
IC / X =
(0.25µA • R CX ) − 0.5mV
R CS
Where RCS is the sense resistor connected to the CSP
and the CSN pins. Note that if RCX = RCL ≤ 19.1kΩ, where
RCL is the charge current programming resistor, then the
charge current termination value is one tenth the full charge
current, more familiarly known as C/10. Short this pin to
GND to disable CX termination.
⎛R ⎞
IILIM = 2.5µA • ⎜ IL ⎟
⎝ RIS ⎠
where RIS is the sense resistor connected to the IN and
the CLN pins. Leave the pin open for the maximum input
current limit of 50mV/RIS.
4000f
8
LTC4000
Pin Functions
(QFN/SSOP)
CL (Pin 8/Pin 4): Charge Current Limit Programming. Connect the charge current programming resistor (RCL) to this
pin. This pin sources 50µA of current. The regulation loop
compares the voltage on this pin with the charge current
monitor voltage (VIBMON), and drives the ITH pin accordingly to ensure that the programmed charge current limit
is not exceeded. The charge current limit is determined
using the following formula:
ICLIM
⎛R ⎞
= 2.5µA • ⎜ CL ⎟
⎝ RCS ⎠
Where RCS is the sense resistor connected to the CSP
and the CSN pins. Leave the pin open for the maximum
charge current limit of 50mV/RCS.
TMR (Pin 9/Pin 5): Charge Timer. Attach 1nF of external
capacitance (CTMR) to GND for each 104 seconds of charge
termination time and 26 seconds of bad battery indicator
time. Short to GND to prevent bad battery indicator time
and charge termination time from expiring – allowing a
continuous trickle charge and top off float voltage regulation charge. Short to BIAS to disable bad battery detect
and enable C/X charging termination.
GND (Pins 10, 28, 29/Pins 6, 24): Device Ground Pins.
Connect the ground pins to a suitable PCB copper ground
plane for proper electrical operation. The QFN package
exposed pad must be soldered to PCB ground for rated
thermal performance.
FLT, CHRG (Pin 11, Pin 12/Pin 7, Pin 8): Charge Status
Indicator Pins. These pins are high voltage open drain pull
down pins. The FLT pin pulls down when there is an under
or over temperature condition during charging or when
the voltage on the BFB pin stays below the low battery
threshold during charging for a period longer than the bad
battery indicator time. The CHRG pin pulls down during
a charging cycle. Please refer to the application information section for details on specific modes indicated by the
combination of the states of these two pins. Pull up each
of these pins with an LED in series with a resistor to a
voltage source to provide a visual status indicator. Short
these pins to GND when not used.
BIAS (Pin 13/Pin 9): 2.9V Regulator Output. Connect a
capacitor of at least 470nF to bypass this 2.9V regulated
voltage output. Use this pin to bias the resistor divider to
set up the voltage at the NTC pin.
NTC (Pin 14/Pin 10): Thermistor Input. Connect a thermistor from NTC to GND, and a corresponding resistor
from BIAS to NTC. The voltage level on this pin determines
if the battery temperature is safe for charging. The charge
current and charge timer are suspended if the thermistor
indicates a temperature that is unsafe for charging. Once the
temperature returns to the safe region, charging resumes.
Leave the pin open or connected to a capacitor to disable
the temperature qualified charging function.
FBG (Pin 15/Pin 11): Feedback Ground Pin. This is the
ground return pin for the resistor dividers connected to
the BFB and OFB pins. As soon as the voltage at IN is valid
(>3V typical), this pin has a 100Ω resistance to GND. When
the voltage at IN is not valid, this pin is disconnected from
GND to ensure that the resistor dividers connected to the
BFB and OFB pins do not continue to drain the battery
when the battery is the only available power source.
BFB (Pin 16/Pin 12): Battery Feedback Voltage Pin. This
pin is a high impedance input pin used to sense the battery
voltage level. In regulation, the battery float voltage loop
sets the voltage on this pin to 1.136V (typical). Connect
this pin to the center node of a resistor divider between
the BAT pin and the FBG pin to set the battery float voltage.
The battery float voltage can then be obtained as follows:
VFLOAT =
RBFB2 + RBFB1
• 1.136V
RBFB2
BAT (Pin 17/Pin 13): Battery Pack Connection. Connect
the battery to this pin. This pin is the anode of the battery
ideal diode driver (the cathode is the CSN pin).
4000f
9
LTC4000
Pin Functions
(QFN/SSOP)
BGATE (Pin 18/Pin 14): External Battery PMOS Gate Drive
Output. When not charging, the BGATE pin drives the
external PMOS to behave as an ideal diode from the BAT
pin (anode) to the CSN pin (cathode). This allows efficient
delivery of any required additional power from the battery
to the downstream system connected to the CSN pin.
When charging a heavily discharged battery, the BGATE
pin is regulated to set the output feedback voltage (OFB
pin) to 86% of the battery float voltage (0.974V typical).
This allows the instant-on feature, providing an immediate valid voltage level at the output when the LTC4000 is
charging a heavily discharged battery. Once the voltage
on the OFB pin is above the 0.974V typical value, then the
BGATE pin is driven low to ensure an efficient charging
path from the CSN pin to the BAT pin.
CSN (Pin 19/Pin 15): Charge Current Sense Negative Input
and Battery Ideal Diode Cathode. Connect a sense resistor
between this pin and the CSP pin. The LTC4000 senses
the voltage across this sense resistor and regulates it to
a voltage equal to 1/20th (typical) of the voltage set at the
CL pin. The maximum regulated sense voltage is 50mV.
The CSN pin is also the cathode input of the battery ideal
diode driver (the anode input is the BAT pin). Tie this pin
to the CSP pin if no charge current limit is desired. Refer to
the Applications Information section for complete details.
CSP (Pin 20/Pin 16): Charge Current Sense Positive Input
and Input Ideal Diode Cathode. Connect a sense resistor between this pin and the CSN pin for charge current
sensing and regulation. This input should be tied to CSN
to disable the charge current regulation function. This
pin is also the cathode of the input ideal diode driver (the
anode is the IID pin).
OFB (Pin 21/Pin 17): Output Feedback Voltage Pin. This
pin is a high impedance input pin used to sense the output
voltage level. In regulation, the output voltage loop sets the
voltage on this feedback pin to 1.193V. Connect this pin
to the center node of a resistor divider between the CSP
pin and the FBG pin to set the output voltage when battery
charging is terminated and all the output load current is
provided from the input. The output voltage can then be
obtained as follows:
VOUT =
ROFB2 + ROFB1
• 1.193V
ROFB2
When charging a heavily discharged battery (such that VOFB
< VOUT(INST_ON)), the battery PowerPath PMOS connected
to BGATE is regulated to set the voltage on this feedback
pin to 0.974V (approximately 86% of the battery float
voltage). The instant-on output voltage is then as follows:
VOUT(INST _ ON) =
ROFB2 + ROFB1
• 0.974V
ROFB2
IGATE (Pin 22/Pin 18): Input PMOS Gate Drive Output. The
IGATE pin drives the external PMOS to behave as an ideal
diode from the IID pin (anode) to the CSP pin (cathode).
IID (Pin 23/Pin 19): Input Ideal Diode Anode. This pin is
the anode of the input ideal diode driver (the cathode is
the CSP pin).
ITH (Pin 24/Pin 20): High Impedance Control Voltage Pin.
When any of the regulation loops (input current, charge
current, battery float voltage or the output voltage) indicate
that its limit is reached, the ITH pin will sink current (up to
1mA) to regulate that particular loop at the limit. In many
applications, this ITH pin is connected to the control/compensation node of a DC/DC converter. Without any external
pull-up, the operating voltage range on this pin is GND to
2.5V. With an external pull-up, the voltage on this pin can
be pulled up to 6V. Note that the impedance connected to
this pin affects the overall loop gain. For details, refer to
the Applications Information section.
4000f
10
LTC4000
Pin Functions
(QFN/SSOP)
CC (Pin 25/Pin 21): Converter Compensation Pin. Connect
an R-C network from this pin to the ITH pin to provide a
suitable loop compensation for the converter used. Refer
to the Applications Information section for discussion and
procedure on choosing an appropriate R-C network for a
particular DC/DC converter.
CLN (Pin 26/Pin 22): Input Current Sense Negative Input.
Connect a sense resistor between this pin and the IN pin.
The LTC4000 senses the voltage across this sense resistor and regulates it to a voltage equal to 1/20th (typical)
of the voltage set at the IL pin. Tie this pin to the IN pin if
no input current limit is desired. Refer to the Applications
Information section for complete details.
IN (Pin 27/Pin 23): Input Supply Voltage: 3V to 60V.
Supplies power to the internal circuitry and the BIAS pin.
Connect the power source to the downstream system
and the battery charger to this pin. This pin is also the
positive sense pin for the input current limit. Connect a
sense resistor between this pin and the CLN pin. Tie this
pin to CLN if no input current limit is desired. A local 0.1µF
bypass capacitor to ground is recommended on this pin.
4000f
11
LTC4000
Block Diagram
RIS
CIN
RVM1
OUT
DC/DC CONVERTER
CCLN
RC
IN
CLN
RST
VM
–
CIBMON
CC
ITH
CC
IID
IBMON
IGATE
CSP
CP1
RVM2
1.193V
8mV
+
+
–
–
+
A8
gm = 0.33m
A9
gm = 0.33m
60k
A1
+–
IIMON
8mV
gm
A11
50µA
RIL
A4
IL
1V
+
–gm
–
gm–
BIAS
A5
–
CBIAS
ROFB1
5µA/
50µA
1V
gm+
CL
–
A7
+
–
IN
LDO,
BG,
REF
BATTERY IDEAL DIODE
AND INSTANT-ON DRIVER
ITH AND CC DRIVER
A10
BGATE
ENABLE
CHARGING
0.974V
60k
RCS
CSN
LINEAR
GATE
DRIVER
AND
VOLTAGE
CLAMP
A2
INPUT IDEAL
DIODE DRIVER
BIAS
A6
+
–
REF
gm
RCL
OFB
gm
BIAS
OFB
1.193V
CP6
BFB
–
0.771V
ROFB2
1.136V
+
BFB
+
CP4
TOO COLD
–
–
RBFB2
1.109V
FBG
NTC FAULT
+
RBFB1
+
CP5
NTC
SYSTEM
LOAD
CL
CIID
–
+
IN
LOGIC
CP3
–
CP2
BAT
–
BIAS
TOO HOT
+
CBAT
5µA
10mV
–+
BIAS
+
–
RNTC
CX
BATTERY PACK
2µA
TMR
CTMR
RCX
OSCILLATOR
GND
ENC
CHRG
FLT
4000 BD
R3
Figure 1. LTC4000 Functional Block Diagram
4000f
12
LTC4000
Operation
Overview
The LTC4000 is designed to simplify the conversion of
any externally compensated DC/DC converter into a high
performance battery charger with PowerPath control. It
only requires the DC/DC converter to have a control or
external-compensation pin (usually named VC or ITH)
whose voltage level varies in a positive monotonic way
with its output. The output variable can be either output
voltage or output current. For the following discussion,
refer to the Block Diagram in Figure 1.
The LTC4000 includes four different regulation loops: input
current, charge current, battery float voltage and output
voltage (A4-A7). Whichever loop requires the lowest voltage on the ITH pin for its regulation controls the external
DC/DC converter.
The input current regulation loop ensures that the programmed input current limit (using a resistor at IL) is not
exceeded at steady state. The charge current regulation
loop ensures that the programmed battery charge current
limit (using a resistor at CL) is not exceeded. The float voltage regulation loop ensures that the programmed battery
stack voltage (using a resistor divider from BAT to FBG
via BFB) is not exceeded. The output voltage regulation
loop ensures that the programmed system output voltage
(using a resistor divider from CSP to FBG via OFB) is not
exceeded. The LTC4000 also provides monitoring pins
for the input current and charge current at the IIMON and
IBMON pins respectively.
The LTC4000 features an ideal diode controller at the input
from the IID pin to the CSP pin and a PowerPath controller
at the output from the BAT pin to the CSN pin. The output
PowerPath controller behaves as an ideal diode controller
when not charging. When charging, the output PowerPath
controller has two modes of operation. If VOFB is greater
than VOUT(INST_ON), BGATE is driven low. When VOFB is
less than VOUT(INST_ON), a linear regulator implements
the instant-on feature. This feature provides regulation of
the BGATE pin so that a valid voltage level is immediately
available at the output when the LTC4000 is charging an
over-discharged, dead or short faulted battery.
The state of the ENC pin determines whether charging
is enabled. When ENC is grounded, charging is disabled
and the battery float voltage loop is disabled. Charging is
enabled when the ENC pin is left floating or pulled high
(≥1.5V)
The LTC4000 offers several user configurable battery
charge termination schemes. The TMR pin can be configured for either C/X termination, charge timer termination or
no termination. After a particular charge cycle terminates,
the LTC4000 features an automatic recharge cycle if the
battery voltage drops below 97.6% of the programmed
float voltage.
Trickle charge mode drops the charge current to one
tenth of the normal charge current (programmed using a
resistor from the CL pin to GND) when charging into an
over discharged or dead battery. When trickle charging,
a capacitor on the TMR pin can be used to program a
time out period. When this bad battery timer expires and
the battery voltage fails to charge above the low battery
threshold (VLOBAT), the LTC4000 will terminate charging
and indicate a bad battery condition through the status
pins (FLT and CHRG).
The LTC4000 also includes an NTC pin, which provides
temperature qualified charging when connected to an NTC
thermistor thermally coupled to the battery pack. To enable
this feature, connect the thermistor between the NTC and
the GND pins, and a corresponding resistor from the BIAS
pin to the NTC pin. The LTC4000 also provides a charging
status indicator through the FLT and the CHRG pins.
Aside from biasing the thermistor-resistor network, the
BIAS pin can also be used for a convenient pull up voltage.
This pin is the output of a low dropout voltage regulator
that is capable of providing up to 0.5mA of current. The
regulated voltage on the BIAS pin is available as soon as
the IN pin is within its operating range (≥3V).
Input Ideal Diode
The input ideal diode feature provides low loss conduction
and reverse blocking from the IID pin to the CSP pin. This
reverse blocking prevents reverse current from the output
(CSP pin) to the input (IID pin) which causes unnecessary drain on the battery and in some cases may result
in unexpected DC/DC converter behavior.
4000f
13
LTC4000
Operation
The ideal diode behavior is achieved by controlling an
external PMOS connected to the IID pin (drain) and the
CSP pin (source). The controller (A1) regulates the external
PMOS by driving the gate of the PMOS device such that the
voltage drop across IID and CSP is 8mV (typical). When
the external PMOS ability to deliver a particular current
with an 8mV drop across its source and drain is exceeded,
the voltage at the gate clamps at VIGATE(ON) and the PMOS
behaves like a fixed value resistor (RDS(ON)).
Input Current Regulation and Monitoring
One of the loops driving the ITH and CC pins is the input
current regulation loop (Figure 2). This loop prevents
the input current sensed through the input current sense
resistor (RIS) from exceeding the programmed input
current limit.
RIS
IN
LOAD
CCLN
(OPTIONAL)
A8
IN
CLN
+
–
CIN
A8
gm = 0.33m
IIMON
BIAS
CIIMON
(OPTIONAL)
LTC4000
60k
50µA
RIL
1V
IL
CC
+
–
–
A4
CC
–
+
RC
ITH
TO DC/DC
4000 FO2
Figure 2. Input Current Regulation Loop
Battery Charger Overview
In addition to the input current regulation loop, the
LTC4000 regulates charge current, battery voltage and
output voltage.
When a battery charge cycle begins, the battery charger
first determines if the battery is over-discharged. If the battery feedback voltage is below VLOBAT, an automatic trickle
charge feature uses the charge current regulation loop to
set the battery charge current to 10% of the programmed
full scale value. If the TMR pin is connected to a capacitor
or open, the bad battery detection timer is enabled. When
this bad battery detection timer expires and the battery voltage is still below VLOBAT, the battery charger automatically
terminates and indicates, via the FLT and CHRG pins, that
the battery was unresponsive to charge current.
Once the battery voltage is above VLOBAT, the charge current
regulation loop begins charging in full power constantcurrent mode. In this case, the programmed full charge
current is set with a resistor on the CL pin.
Depending on available input power and external load
conditions, the battery charger may not be able to charge
at the full programmed rate. The external load is always
prioritized over the battery charge current. The input
current limit programming is always observed, and only
additional power is available to charge the battery. When
system loads are light, battery charge current is maximized.
Once the float voltage is achieved, the battery float voltage regulation loop takes over from the charge current
regulation loop and initiates constant voltage charging. In
constant voltage charging, charge current slowly declines.
Charge termination can be configured with the TMR pin
in several ways. If the TMR pin is tied to the BIAS pin,
C/X termination is selected. In this case, charging is
terminated when constant voltage charging reduces the
charge current to the C/X level programmed at the CX
pin. Connecting a capacitor to the TMR pin selects the
charge timer termination and a charge termination timer
is started at the beginning of constant voltage charging.
Charging terminates when the termination timer expires.
When continuous charging at the float voltage is desired,
tie the TMR pin to GND to disable termination.
Upon charge termination, the PMOS connected to BGATE
behaves as an ideal diode from BAT to CSN. The diode
function prevents charge current but provides current
to the system load as needed. If the system load can be
completely supplied from the input, the battery PMOS turns
off. While terminated, if the input current limit is not in
regulation, the output voltage regulation loop takes over to
ensure that the output voltage at CSP remains in control.
The output voltage regulation loop regulates the voltage
at the CSP pin such that the output feedback voltage at
the OFB pin is 1.193V.
If the system load requires more power than is available
from the input, the battery ideal diode controller provides
supplemental power from the battery. When the battery
voltage discharges below 97.1% of the float voltage
(VBFB < VRECHRG(FALL)), the automatic recharge feature
initiates a new charge cycle.
4000f
14
LTC4000
Operation
Charge Current Regulation
The first loop involved in a normal charging cycle is the
charge current regulation loop (Figure 3). As with the input
current regulation loop, this loop also drives the ITH and
CC pins. This loop ensures that the charge current sensed
through the charge current sense resistor (RCS) does not
exceed the programmed full charge current.
+
–
IBMON
CSN
LTC4000
A9
gm = 0.33m
60k
BIAS
RCL
BFB
+
–
FBG
1.136V
A6
CC
CC
–
+
RC
ITH
TO DC/DC
Figure 4. Battery Float Voltage Regulation Loop with FBG
ROFB1
CSP
LTC4000
4000 FO4
BAT PMOS
CCSP
CIBMON
(OPTIONAL)
RBFB2
BAT
TO SYSTEM
RIS
CSP
RBFB1
50µA AT NORMAL 1V
5µA AT TRICKLE
ROFB2
CSP
LTC4000
OFB
+
–
FBG
1.193V
A7
CC
CC
–
+
RC
ITH
TO DC/DC
CC
+
–
–
A5
CC
–
+
4000 FO5
RC
ITH
TO DC/DC
CL
4000 FO3
Figure 3. Charge Current Regulation Loop
Battery Voltage Regulation
Once the float voltage is reached, the battery voltage regulation loop takes over from the charge current regulation
loop (Figure 4).
The float voltage level is programmed using the feedback
resistor divider between the BAT pin and the FBG pin with
the center node connected to the BFB pin. Note that the
ground return of the resistor divider is connected to the
FBG pin. The FBG pin disconnects the resistor divider
load when VIN < 3V to ensure that the float voltage resistor divider does not consume battery current when the
battery is the only available power source. For VIN ≥ 3V,
the typical resistance from the FBG pin to GND is 100Ω.
Output Voltage Regulation
When charging terminates and the system load is completely supplied from the input, the PMOS connected to
BGATE is turned off. In this scenario, the output voltage
regulation loop takes over from the battery float voltage
regulation loop (Figure 5). The output voltage regulation
loop regulates the voltage at the CSP pin such that the
output feedback voltage at the OFB pin is 1.193V.
Figure 5. Output Voltage Regulation Loop with FBG
Battery Instant-On and Ideal Diode
The LTC4000 controls the external PMOS connected to the
BGATE pin with a controller similar to the input ideal diode
controller driving the IGATE pin. When not charging, the
PMOS behaves as an ideal diode between the BAT (anode)
and the CSN (cathode) pins. The controller (A2) regulates
the external PMOS to achieve low loss conduction by driving the gate of the PMOS device such that the voltage drop
from the BAT pin to the CSN pin is 8mV. When the ability
to deliver a particular current with an 8mV drop across
the PMOS source and drain is exceeded, the voltage at
the gate clamps at VBGATE(ON) and the PMOS behaves like
a fixed value resistor (RDS(ON)).
The ideal diode behavior allows the battery to provide current to the load when the input supply is in current limit
or the DC/DC converter is slow to react to an immediate
load increase at the output. In addition to the ideal diode
behavior, BGATE also allows current to flow from the CSN
pin to the BAT pin during charging.
There are two regions of operation when current is flowing
from the CSN pin to the BAT pin. The first is when charging into a battery whose voltage is below the instant-on
threshold (VOFB < VOUT(INST_ON)). In this region of operation, the controller regulates the voltage at the CSP pin
4000f
15
LTC4000
Operation
to be approximately 86% of the final float voltage level
(VOUT(INST_ON)). This feature provides a CSP voltage significantly higher than the battery voltage when charging
into a heavily discharged battery. This instant-on feature
allows the LTC4000 to provide sufficient voltage at the
output (CSP pin), independent of the battery voltage.
The second region of operation is when the battery feedback
voltage is greater than or equal to the instant-on threshold
(VOUT(INST_ON)). In this region, the BGATE pin is driven
low and clamped at VBGATE(ON) to allow the PMOS to turn
completely on, reducing any power dissipation due to the
charge current.
Battery Temperature Qualified Charging
The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the
battery pack. The comparators CP3 and CP4 implement
the temperature detection as shown in the Block Diagram
in Figure 1. The rising threshold of CP4 is set at 75% of
VBIAS (cold threshold) and the falling threshold of CP3 is
set at 35% of VBIAS (hot threshold). When the voltage at
the NTC pin is above 75% of VBIAS or below 35% of VBIAS
then the LTC4000 pauses any charge cycle in progress.
When the voltage at the NTC pin returns to the range of
40% to 70% of VBIAS, charging resumes.
When charging is paused, the external charging PMOS
turns off and charge current drops to zero. If the LTC4000
is charging in the constant voltage mode and the charge
termination timer is enabled, the timer pauses until the
thermistor indicates a return to a valid temperature. If the
battery charger is in the trickle charge mode and the bad
battery detection timer is enabled, the bad battery timer
pauses until the thermistor indicates a return to a valid
temperature.
Input UVLO and Voltage Monitoring
The regulated voltage on the BIAS pin is available as soon
as VIN ≥ 3V. When VIN ≥ 3V, the FBG pin is pulled low to
GND with a typical resistance of 100Ω and the rest of the
chip functionality is enabled.
When the IN pin is high impedance and a battery is connected to the BAT pin, the BGATE pin is pulled down with
a 2μA (typical) current source to hold the battery PMOS
gate voltage at VBGATE(ON) below VBAT. This allows the
battery to power the output. The total quiescent current
consumed by LTC4000 from the battery when IN is not
valid is typically ≤ 10µA.
Besides the internal input UVLO, the LTC4000 also provides
voltage monitoring through the VM pin. The RST pin is
pulled low when the voltage on the VM pin falls below
1.193V (typical). On the other hand, when the voltage on
the VM pin rises above 1.233V (typical), the RST pin is
high impedance.
One common use of this voltage monitoring feature is to
ensure that the converter is turned off when the voltage
at the input is below a certain level. In this case, connect
the RST pin to the DC/DC converter chip select or enable
pin (see Figure 6).
RIS
IN
RVM1
IN
VM
IN
CLN
–
DC/DC
CONVERTER
EN
RST
CP1
RVM2
1.193V
+
LTC4000
4000 FO6
Figure 6. Input Voltage Monitoring with RST Connected to
the EN Pin of the DC/DC Converter
4000f
16
LTC4000
Applications Information
Input Ideal Diode PMOS Selection
The input external PMOS is selected based on the expected
maximum current, power dissipation and reverse voltage drop. The PMOS must be able to withstand a gate to
source voltage greater than VIGATE(ON) (15V maximum) or
the maximum regulated voltage at the IID pin, whichever
is less. A few appropriate external PMOS for a number of
different requirements are shown at Table 1.
Table 1. PMOS
RDS(ON) AT
VGS = 10V
(Ω)
MAX ID
(A)
SiA923EDJ
0.054
4.5
–20
Vishay
Si9407BDY
0.120
4.7
–60
Vishay
Si4401BDY
0.014
10.5
–40
Vishay
Si4435DDY
0.024
11.4
–30
Vishay
SUD19P06-60
0.060
18.3
–60
Vishay
Si7135DP
0.004
60
–30
Vishay
PART NUMBER
MAX VDS
(V)
MANUFACTURER
Note that in general the larger the capacitance seen on
the IGATE pin, the slower the response of the ideal diode
driver. The fast turn off and turn on current is limited to
–0.5mA and 0.7mA typical respectively (IIGATE(FASTOFF) and
IIGATE(FASTON)). If the driver can not react fast enough to a
sudden increase in load current, most of the extra current
is delivered through the body diode of the external PMOS.
This increases the power dissipation momentarily. It is
important to ensure that the PMOS is able to withstand
this momentary increase in power dissipation.
Input Current Limit Setting and Monitoring
The regulated input current limit is set using a resistor at
the IL pin according to the following formula:
RIS =
VIL
20 • IILIM
The input current through the sense resistor is available
for monitoring through the IIMON pin. The voltage on
the IIMON pin varies with the current through the sense
resistor as follows:
VIIMON = 20 • IRIS • RIS = 20 • ( VIN – VCLN )
The regulation voltage level at the IIMON pin is clamped
at 1V with an accurate internal reference. At 1V on the
IIMON pin, the input current limit is regulated at the following value:
IILIM(MAX)(A) =
0.050V
RIS (Ω)
When this maximum current limit is desired, leave the IL
pin open or set it to a voltage >1.05V such that amplifier
A4 can regulate the IIMON voltage accurately to the internal
reference of 1V.
If the input current is noisy, add a filter capacitor to the
CLN pin to reduce the AC content. For example, when using a buck DC/DC converter, the use of a CCLN capacitor
is strongly recommended. Where the highest accuracy is
important, pick the value of CCLN such that the AC content
is less than or equal to 50% of the average voltage across
the sense resistor.
The voltage on the IIMON pin can be filtered further by
putting a capacitor on the pin (CIIMON). The voltage on the
IIMON pin is also the feedback input to the input current
regulation error amplifier. Any capacitor connected to
this pin places a pole in the input current regulation loop.
Therefore, this filter capacitor should NOT be arbitrarily
large as it will slow down the overall compensated loop.
For details on loop compensation please refer to the
Compensation section.
where VIL is the voltage on the IL pin. The IL pin is internally
pulled up with an accurate current source of 50µA. Therefore
an equivalent formula to obtain the input current limit is:
RIL =
ILIM • RIS
R
⇒ IILIM = IL • 2.5µA
2.5µA
RIS
4000f
17
LTC4000
Applications Information
Charge Current Limit Setting and Monitoring
The regulated full charge current is set according to the
following formula:
RCS
VCL
=
20 • ICLIM
where VCL is the voltage on the CL pin. The CL pin is
internally pulled up with an accurate current source of
50µA. Therefore, an equivalent formula to obtain the input
current limit is:
RCL
•R
I
R
= CLIM CS ⇒ ICLIM = CL • 2.5µA
2.5µA
RCS
The charge current through the sense resistor is available
for monitoring through the IBMON pin. The voltage on
the IBMON pin varies with the current through the sense
resistor as follows:
VIBMON = 20 • IRCS • RCS = 20 • ( VCSP – VCSN )
Similar to the IIMON pin, the regulation voltage level at
the IBMON pin is clamped at 1V with an accurate internal
reference. At 1V on the IBMON pin, the charge current
limit is regulated to the following value:
0.050V
ICLIM(MAX)(A) =
RCS (Ω)
When this maximum charge current limit is desired, leave
the CL pin open or set it to a voltage >1.05V such that
amplifier A5 can regulate the IBMON pin voltage accurately
to the internal reference of 1V.
When the output current waveform of the DC/DC converter
or the system load current is noisy, it is recommended that
a capacitor is connected to the CSP pin (CCSP). This is to
reduce the AC content of the current through the sense
resistor (RCS). Where the highest accuracy is important,
pick the value of CCSP such that the AC content is less
than or equal to 50% of the average voltage across the
sense resistor. Similar to the IIMON pin, the voltage on the
IBMON pin is filtered further by putting a capacitor on the
pin (CIBMON). This filter capacitor should not be arbitrarily
large as it will slow down the overall compensated charge
current regulation loop. For details on the loop compensation, refer to the Compensation section.
Battery Float Voltage Programming
When the value of RBFB1 is much larger than 100Ω, the final
float voltage is determined using the following formula:
⎛V
⎞
RBFB1 = ⎜ FLOAT – 1⎟RBFB2
⎝ 1.136V ⎠
When higher accuracy is important, a slightly more accurate final float voltage can be determined using the
following formula:
⎛R
⎞ ⎛R
⎞
+R
VFLOAT = ⎜ BFB1 BFB2 • 1.136V ⎟ – ⎜ BFB1 • VFBG ⎟
RBFB2
⎝
⎠ ⎝ RBFB2
⎠
where VFBG is the voltage at the FBG pin during float
voltage regulation, which accounts for all the current
from all resistor dividers that are connected to this pin
(RFBG = 100Ω typical).
Low Battery Trickle Charge Programming and Bad
Battery Detection
When charging into an over-discharged or dead battery
(VBFB < VLOBAT), the pull-up current at the CL pin is reduced
to 10% of the normal pull-up current. Therefore, the trickle
charge current is set using the following formula:
RCL =
ICLIM(TRKL) • RCS
0.25µA
⇒ ICLIM(TRKL) = 0.25µA •
RCL
RCS
Therefore, when 50µA•RCL is less than 1V, the following
relation is true:
ICLIM(TRKL) =
ICLIM
10
Once the battery voltage rises above the low battery voltage
threshold, the charge current level rises from the trickle
charge current level to the full charge current level.
The LTC4000 also features bad battery detection. This
detection is disabled if the TMR pin is grounded or tied
to BIAS. However, when a capacitor is connected to the
4000f
18
LTC4000
Applications Information
TMR pin, a bad battery detection timer is started as soon
as trickle charging starts. If at the end of the bad battery
detection time the battery voltage is still lower than the
low battery threshold, charging is terminated and the part
indicates a bad battery condition by pulling the FLT pin low
and leaving the CHRG pin high impedance.
The bad battery detection time can be programmed according to the following formula:
CTMR (nF) = tBADBAT (h) • 138.5
Note that once a bad battery condition is detected, the
condition is latched. In order to re-enable charging, remove the battery and connect a new battery whose voltage
causes BFB to rise above the recharge battery threshold
(VRECHRG(RISE)). Alternatively toggle the ENC pin or remove
and reapply power to IN.
C/X Detection, Charge Termination and Automatic
Recharge
Once the constant voltage charging is reached, there are
two ways in which charging can terminate. If the TMR pin
is tied to BIAS, the battery charger terminates as soon as
the charge current drops to the level programmed by the
CX pin. The C/X current termination level is programmed
according to the following formula:
RCX =
IC/ X • RCS
(0.25µA • RCX ) − 0.5mV
+ 0.5mV ⇒ IC/X =
0.25µA
RCS
where RCS is the charge current sense resistor connected
between the CSP and the CSN pins.
When the voltage at BFB is higher than the recharge
threshold (97.6% of float), the C/X comparator is enabled.
In order to ensure proper C/X termination coming out of
a paused charging condition, connect a capacitor on the
CX pin according to the following formula:
CCX = 100CBGATE
where CBGATE is the total capacitance connected to the
BGATE pin.
For example, a typical capacitance of 1nF requires a capacitor greater than 100nF connected to the CX pin to ensure
proper C/X termination behavior.
If a capacitor is connected to the TMR pin, as soon as the
constant voltage charging is achieved, a charge termination timer is started. When the charge termination timer
expires, the charge cycle terminates. The total charge
termination time can be programmed according to the
following formula:
CTMR (nF) = t TERMINATE (h) • 34.6
If the TMR pin is grounded, charging never terminates and
the battery voltage is held at the float voltage. Note that
regardless of which termination behavior is selected, the
CHRG and FLT pins will both assume a high impedance
state as soon as the charge current falls below the programmed C/X level.
After the charger terminates, the LTC4000 automatically
restarts another charge cycle if the battery feedback voltage
drops below 97.1% of the programmed final float voltage
(VRECHRG(FALL)). When charging restarts, the CHRG pin
pulls low and the FLT pin remains high impedance.
Output Voltage Regulation Programming
The output voltage regulation level is determined using
the following formula:
⎛V
⎞
ROFB1 = ⎜ OUT − 1⎟ • ROFB2
⎝ 1.193 ⎠
As in the battery float voltage calculation, when higher
accuracy is important, a slightly more accurate output is
determined using the following formula:
⎛R
⎞
⎞ ⎛R
+R
VOUT = ⎜ OFB1 OFB2 • 1.193V ⎟ – ⎜ OFB1 • VFBG ⎟
ROFB2
⎝
⎠
⎠ ⎝ ROFB2
where VFBG is the voltage at the FBG pin during output
voltage regulation, which accounts for all the current from
all resistor dividers that are connected to this pin.
4000f
19
LTC4000
Applications Information
Battery Instant-On and Ideal Diode External PMOS
Consideration
On the other hand, when the battery voltage is above the
low battery threshold but still below the instant-on threshold, the power dissipation can be calculated as follows:
The instant-on voltage level is determined using the following formula:
PINST _ ON = [0.86 • VFLOAT – VBAT ] • ICLIM
ROFB1 + ROFB2
• 0.974V
ROFB2
Note that ROFB1 and ROFB2 are the same resistors that
program the output voltage regulation level. Therefore,
the output voltage regulation level is always 122.5% of
the instant-on voltage level.
where ICLIM is the full scale charge current limit.
VOUT(INST _ ON) =
For example, when charging a 3-cell Lithium Ion battery
with a programmed full charged current of 1A, the float
voltage is 12.6V, the bad battery voltage level is 8.55V and
the instant-on voltage level is 10.8V. During instant-on
operation and in the trickle charge mode, the worst case
maximum power dissipation in the PMOS is 1.08W. When
the battery voltage is above the bad battery voltage level,
then the worst case maximum power dissipation is 2.25W.
During instant-on operation, it is critical to consider the
charging PMOS power dissipation. When the battery voltage is below the low battery threshold (VLOBAT), the power
dissipation in the PMOS can be calculated as follows:
When overheating of the charging PMOS is a concern, it is
recommended that the user add a temperature detection
circuit that pulls down on the NTC pin. This pauses charging whenever the external PMOS temperature is too high.
A sample circuit that performs this temperature detection
function is shown in Figure 7.
PTRKL = [0.86 • VFLOAT – VBAT ] • ICLIM(TRKL)
where ICLIM(TRKL) is the trickle charge current limit.
TO SYSTEM
CSP
RCS
LTC4000
VISHAY CURVE 2
NTC RESISTOR
THERMALLY COUPLED
WITH CHARGING PMOS
RNTC2
CSN
BGATE
M2
R4 = RNTC2
AT 25°C
BAT
BIAS
R3
CBIAS
162k
2N7002L
NTC
Li-Ion
BATTERY PACK
RNTC1
LTC1540
–
+
RISING
TEMPERATURE
THRESHOLD
SET AT 90°C
20k
VOLTAGE HYSTERESIS CAN
BE PROGRAMMED FOR
TEMPERATURE HYSTERESIS
86mV ≈ 10°C
4000 F07
Figure 7. Charging PMOS Overtemperature Detection Circuit
Protecting PMOS from Overheating
4000f
20
LTC4000
Applications Information
Similar to the input external PMOS, the charging external
PMOS must be able to withstand a gate to source voltage
greater than VBGATE(ON) (15V maximum) or the maximum
regulated voltage at the CSP pin, whichever is less. Consider
the expected maximum current, power dissipation and
instant-on voltage drop when selecting this PMOS. The
PMOS suggestions in Table 1 are an appropriate starting
point depending on the application.
Float Voltage, Output Voltage and Instant-On Voltage
Dependencies
The formulas for setting the float voltage, output voltage
and instant-on voltage are repeated here:
RBFB1 + RBFB2
• 1.136V
RBFB2
+R
R
= OFB1 OFB2 • 1.193V
ROFB2
VFLOAT =
VOUT
VOUT(INST _ ON) =
ROFB1 + ROFB2
• 0.974V
ROFB2
In the typical application, VOUT is set higher than VFLOAT
to ensure that the battery is charged fully to its intended
float voltage. On the other hand, VOUT should not be
programmed too high since VOUT(INST_ON), the minimum
voltage on CSP, depends on the same resistors ROFB1 and
ROFB2 that set VOUT. As noted before, this means that the
output voltage regulation level is always 122.5% of the
instant-on voltage. The higher the programmed value of
VOUT(INST_ON), the larger the operating region when the
charger PMOS is driven in the linear region where it is
less efficient.
If ROFB1 and ROFB2 are set to be equal to RBFB1 and RBFB2
respectively, then the output voltage is set at 105% of
the float voltage and the instant-on voltage is set at 86%
of the float voltage. Figure 8 shows the range of possible
output voltages that can be set for VOUT(INST_ON) and VOUT
with respect to VFLOAT to ensure the battery can be fully
charged in an ideal scenario.
Taking into account possible mismatches between the
resistor dividers as well as mismatches in the various
regulation loops, VOUT should not be programmed to
be less than 105% of VFLOAT to ensure that the battery
can be fully charged. This automatically means that the
instant-on voltage level should not be programmed to be
less than 86% of VFLOAT.
POSSIBLE
OUTPUT
VOLTAGE RANGE
105%
NOMINAL OUTPUT VOLTAGE
NOMINAL FLOAT VOLTAGE
POSSIBLE
INSTANT-ON
VOLTAGE RANGE
100%
MINIMUM PRACTICAL
OUTPUT VOLTAGE
100%
100%
86%
NOMINAL INSTANT-ON VOLTAGE
75%
MINIMUM PRACTICAL
INSTANT-ON VOLTAGE
81.6%
4000 F08
Figure 8. Possible Voltage Ranges for VOUT and
VOUT(INST_ON) in Ideal Scenario
4000f
21
LTC4000
Applications Information
Battery Temperature Qualified Charging
To use the battery temperature qualified charging feature,
connect an NTC thermistor, RNTC, between the NTC pin
and the GND pin, and a bias resistor, R3, from the BIAS
pin to the NTC pin (Figure 9). Thermistor manufacturer
datasheets usually include either a temperature lookup
table or a formula relating temperature to the resistor
value at that corresponding temperature.
BIAS
CBIAS
R3
LTC4000
NTC
BAT
NTC RESISTOR
THERMALLY COUPLED
WITH BATTERY PACK
RNTC
4000 F09
Figure 9. NTC Thermistor Connection
In a simple application, R3 is a 1% resistor with a value
equal to the value of the chosen NTC thermistor at 25°C
(R25). In this simple setup, the LTC4000 will pause charging when the resistance of the NTC thermistor drops to
0.54 times the value of R25. For a Vishay Curve 2 thermistor, this corresponds to approximately 41.5°C. As the
temperature drops, the resistance of the NTC thermistor
rises. The LTC4000 is also designed to pause charging
when the value of the NTC thermistor increases to three
times the value of R25. For a Vishay Curve 2 thermistor,
this corresponds to approximately –1.5°C. With Vishay
Curve 2 thermistor, the hot and cold comparators each
have approximately 5°C of hysteresis to prevent oscillation
about the trip point.
The hot and cold threshold can be adjusted by changing
the value of R3. Instead of simply setting R3 to be equal to
R25, R3 is set according to one of the following formulas:
R3 =
Notice that with only one degree of freedom (i.e. adjusting
the value of R3), the user can only use one of the formulas above to set either the cold or hot threshold but not
both. If the value of R3 is set to adjust the cold threshold,
the value of the NTC resistor at the hot threshold is then
equal to 0.179 • RNTC at cold_threshold. Similarly, if the
value of R3 is set to adjust the hot threshold, the value
of the NTC resistor at the cold threshold is then equal to
5.571 • RNTC at cold_threshold.
Note that changing the value of R3 to be larger than R25
will move both the hot and cold threshold lower and vice
versa. For example, using a Vishay Curve 2 thermistor
whose nominal value at 25°C is 100k, the user can set
the cold temperature to be at 5°C by setting the value of
R3 = 75k, which automatically then sets the hot threshold
at approximately 50°C.
It is possible to adjust the hot and cold threshold independently by introducing another resistor as a second degree
of freedom (Figure 10). The resistor RD in effect reduces
the sensitivity of the resistance between the NTC pin and
ground. Therefore, intuitively this resistor will move the hot
threshold to a hotter temperature and the cold threshold
to a colder temperature.
BIAS
R3
LTC4000
CBIAS
NTC
RD
BAT
NTC RESISTOR
THERMALLY COUPLED
WITH BATTERY PACK
RNTC
4000 F10
Figure 10. NTC Thermistor Connection with
Desensitizing Resistor RD
RNTC at cold_ threshold
3
or
R3 = 1.857 • RNTC at hot _ threshold
4000f
22
LTC4000
Applications Information
The value of R3 and RD can now be set according to the
following formula:
R
at cold_ threshold – RNTC at hot _ threshold
R3 = NTC
2.461
RD = 0.219 • RNTC at cold_ threshold –
1.219 • RNTC at hot _ threshold
Note the important caveat that this method can only be
used to desensitize the thermal effect on the thermistor
and hence push the hot and cold temperature thresholds
apart from each other. When using the formulas above,
if the user finds that a negative value is needed for RD,
the two temperature thresholds selected are too close to
each other and a higher sensitivity thermistor is needed.
For example, this method can be used to set the hot
and cold thresholds independently to 60°C and –5°C.
Using a Vishay Curve 2 thermistor whose nominal value
at 25°C is 100k, the formula results in R3 = 130k and
RD = 41.2k for the closest 1% resistors values.
To increase thermal sensitivity such that the valid charging
temperature band is much smaller than 40°C, it is possible to put a PTC (positive thermal coefficient) resistor
in series with R3 between the BIAS pin and the NTC pin.
This PTC resistor also needs to be thermally coupled with
the battery. Note that this method increases the number of
thermal sensing connections to the battery pack from one
wire to three wires. The exact value of the nominal PTC
resistor required can be calculated using a similar method
as described above, keeping in mind that the threshold at
the NTC pin is always 75% and 35% of VBIAS.
Leaving the NTC pin floating or connecting it to a capacitor
disables all NTC functionality.
The FLT and CHRG Indicator Pins
The FLT and CHRG pins in the LTC4000 provide status
indicators. Table 2 summarizes the mapping of the pin
states to the part status.
Table 2. FLT and CHRG Status Indicator
FLT
CHRG
0
0
NTC Over Ranged – Charging Paused
STATUS
1
0
Charging Normally
0
1
Charging Terminated and Bad Battery Detected
1
1
VIBMON < (VC/X – 10mV)
where 1 indicates a high impedance state and 0 indicates
a low impedance pull-down state.
Note that VIBMON < (VCX – 10mV) corresponds to charge
termination only if the C/X termination is selected. If the
charger timer termination is selected, constant voltage
charging may continue for the remaining charger timer
period even after the indicator pins indicate that VIBMON
< (VCX – 10mV). This is also true when no termination is
selected, constant voltage charging will continue even after
the indicator pins indicate that VIBMON < (VCX – 10mV).
The BIAS Pin
For ease of use the LTC4000 provides a low dropout voltage regulator output on the BIAS pin. Designed to provide
up to 0.5mA of current at 2.9V, this pin requires at least
470nF of low ESR bypass capacitance for stability.
Use the BIAS pin as the pull-up source for the NTC resistor networks, since the internal reference for the NTC
circuitry is based on a ratio of the voltage on the BIAS
pin. Furthermore, various 100k pull-up resistors can be
conveniently connected to the BIAS pin.
4000f
23
LTC4000
Applications Information
Setting the Input Voltage Monitoring Resistor Divider
The falling threshold voltage level for this monitoring
function can be calculated as follows:
⎛V
⎞
RVM1 = ⎜ VM _ RST – 1⎟ • R VM2
⎝ 1.193V
⎠
where RVM1 and RVM2 form a resistor divider connected
between the monitored voltage and GND, with the center
tap point connected to the VM pin as shown in Figure 6. The
rising threshold voltage level can be calculated similarly.
Compensation
In order for the LTC4000 to control the external DC/DC
converter, it has to be able to overcome the sourcing bias
current of the ITH or VC pin of the DC/DC converter. The
typical sinking capability of the LTC4000 at the ITH pin is
1mA at 0.4V with a maximum voltage range of 0V to 6V.
It is imperative that the local feedback of the DC/DC
converter be set up such that during regulation of any of
the LTC4000 loops this local loop is out of regulation and
sources as much current as possible from its ITH/VC pin.
For example for a DC/DC converter regulating its output
voltage, it is recommended that the converter feedback
divider is programmed to be greater than 110% of the
output voltage regulation level programmed at the OFB pin.
There are four feedback loops to consider when setting up
the compensation for the LTC4000. As mentioned before
these loops are: the input current loop, the charge current
loop, the float voltage loop and the output voltage loop.
All of these loops have an error amp (A4-A7) followed by
another amplifier (A10) with the intermediate node driving the CC pin and the output of A10 driving the ITH pin
as shown in Figure 11. The most common compensation
network of a series capacitor (CC) and resistor (RC) between
the CC pin and the ITH pin is shown here.
Each of the loops has slightly different dynamics due to
differences in the feedback signal path. The analytic description of each of the loops is included in the Appendix section. In most situations, an alternative empirical approach
to compensation, as described here, is more practical.
A4-A7
gm4-7 = 0.2m
+
–
LTC4000
RO4-7
CC
A10
gm10 = 0.1m
–
+
CC
RC
ITH
RO10
4000 F11
Figure 11. Error Amplifier Followed by Output Amplifier Driving
CC and ITH Pins
Empirical Loop Compensation
Based on the five analytical expressions given in the Appendix section, and the transfer function from the ITH
pin to the input and output current of the external DC/DC
converter, the user can analytically determine the complete
loop transfer function of each of the loops. Once these are
obtained, it is a matter of analyzing the gain and phase
bode plots to ensure that there is enough phase and gain
margin at unity crossover with the selected values of RC
and CC for all operating conditions.
Even though it is clear that an analytical compensation
method is possible, sometimes certain complications
render this method difficult to tackle. These complications include the lack of easy availability of the switching
converter transfer function from the ITH or VC control
node to its input or output current, and the variability of
parameter values of the components such as the ESR of
the output capacitor or the RDS(ON) of the external PFETs.
Therefore a simpler and more practical way to compensate the LTC4000 is provided here. This empirical method
involves injecting an AC signal into the loop, observing
the loop transient response and adjusting the CC and RC
values to quickly iterate towards the final values. Much
of the detail of this method is derived from Application
Note 19 which can be found at www.linear.com using AN19
in the search box.
Figure 12 shows the recommended setup to inject an
AC-coupled output load variation into the loop. A function
generator with 50Ω output impedance is coupled through
a 50Ω/1000µF series RC network to the regulator output.
4000f
24
LTC4000
Applications Information
SWITCHING
CONVERTER
GND
ITH
1k
0.015µF
RC
CC
CLN
ITH
CC
IOUT
CSP
LTC4000
IN
A
10k
B
1500pF
50Ω
1W
1000µF
(OBSERVE
POLARITY)
SCOPE
GROUND
CLIP
CSN
GND BAT BGATE
50Ω
GENERATOR
f = 50Hz
VIN
4000 F12
Figure 12. Empirical Loop Compensation Setup
Generator frequency is set at 50Hz. Lower frequencies
may cause a blinking scope display and higher frequencies may not allow sufficient settling time for the output
transient. Amplitude of the generator output is typically
set at 5VP-P to generate a 100mAP-P load variation. For
lightly loaded outputs (IOUT < 100mA), this level may be
too high for small signal response. If the positive and
negative transition settling waveforms are significantly
different, amplitude should be reduced. Actual amplitude
is not particularly important because it is the shape of
the resulting regulator output waveform which indicates
loop stability.
A 2-pole oscilloscope filter with f = 10kHz is used to
block switching frequencies. Regulators without added
LC output filters have switching frequency signals at their
outputs which may be much higher amplitude than the
low frequency settling waveform to be studied. The filter
frequency is high enough for most applications to pass
the settling waveform with no distortion.
Oscilloscope and generator connections should be made
exactly as shown in Figure 12 to prevent ground loop errors. The oscilloscope is synced by connecting the channel B probe to the generator output, with the ground clip
of the second probe connected to exactly the same place
as channel A ground. The standard 50Ω BNC sync output
of the generator should not be used because of ground
loop errors. It may also be necessary to isolate either
the generator or oscilloscope from its third wire (earth
ground) connection in the power plug to prevent ground
loop errors in the scope display. These ground loop errors
are checked by connecting channel A probe tip to exactly
the same point as the probe ground clip. Any reading on
channel A indicates a ground loop problem.
Once the proper setup is made, finding the optimum
values for the frequency compensation network is fairly
straightforward. Initially, CC is made large (≥1μF) and RC
is made small (≈10k). This nearly always ensures that the
regulator will be stable enough to start iteration. Now, if
the regulator output waveform is single-pole over damped
(see the waveforms in Figure 13), the value of CC is reduced in steps of about 2:1 until the response becomes
slightly under damped. Next, RC is increased in steps of
2:1 to introduce a loop zero. This will normally improve
damping and allow the value of CC to be further reduced.
Shifting back and forth between RC and CC variations will
allow one to quickly find optimum values.
4000f
25
LTC4000
Applications Information
GENERATOR OUTPUT
REGULATOR OUTPUT
WITH LARGE CC, SMALL RC
WITH REDUCED CC, SMALL RC
EFFECT OF INCREASED RC
FURTHER REDUCTION IN CC
MAY BE POSSIBLE
IMPROPER VALUES WILL
CAUSE OSCILLATIONS
4000 F13
Figure 13. Typical Output Transient Response at Various
Stability Level
If the regulator response is under damped with the initial
large value of CC, RC should be increased immediately before
larger values of CC are tried. This will normally bring about
the over damped starting condition for further iteration.
The optimum values for RC and CC normally means the
smallest value for CC and the largest value for RC which
still guarantee well damped response, and which result in
the largest loop bandwidth and hence loop settling that is
as rapid as possible. The reason for this approach is that
it minimizes the variations in output voltage caused by
input ripple voltage and output load transients.
A switching regulator which is grossly over damped will
never oscillate, but it may have unacceptably large output
transients following sudden changes in input voltage or
output loading. It may also suffer from excessive overshoot
problems on startup or short circuit recovery. To guarantee
acceptable loop stability under all conditions, the initial
values chosen for RC and CC should be checked under all
conditions of input voltage and load current. The simplest
way of accomplishing this is to apply load currents of
minimum, maximum and several points in between. At
each load current, input voltage is varied from minimum
to maximum while observing the settling waveform.
If large temperature variations are expected for the system,
stability checks should also be done at the temperature
extremes. There can be significant temperature variations in several key component parameters which affect
stability; in particular, input and output capacitor value
and their ESR, and inductor permeability. The external
converter parametric variations also need some consideration especially the transfer function from the ITH/VC
pin voltage to the output variable (voltage or current). The
LTC4000 parameters that vary with temperature include
the transconductance and the output resistance of the
error amplifiers (A4-A7). For modest temperature variations, conservative over damping under worst-case room
temperature conditions is usually sufficient to guarantee
adequate stability at all temperatures.
One measure of stability margin is to vary the selected
values of both RC and CC by 2:1 in all four possible combinations. If the regulator response remains reasonably
well damped under all conditions, the regulator can be
considered fairly tolerant of parametric variations. Any
tendency towards an under damped (ringing) response
indicates that a more conservative compensation may
be needed.
4000f
26
LTC4000
Applications Information
Design Example
In this design example, the LTC4000 is paired with the
LT3845 buck converter to create a 10A, 3-cell LiFePO4
battery charger. The circuit is shown on the front page
and is repeated here in Figure 14.
• RCL is set at 24.9kΩ such that the voltage at the CL pin
is 1.25V. Similar to the IIMON pin, the regulation voltage
on the IBMON pin is clamped at 1V with an accurate
internal reference. Therefore, the charge current limit
is set at 10A according to the following formula:
• The input voltage monitor falling threshold is set at
14.3V according to the following formula:
0.050V 0.050V
=
= 10A
RCS
5mΩ
• The trickle charge current level is consequently set at
1.25A, according to the following formula:
⎛ 14.3V
⎞
RVM1 = ⎜
− 1⎟ • 100kΩ ≈ 1.10MΩ
⎝ 1.193V ⎠
• The IL pin is left open such that the voltage on this pin
is >1.05V. The regulation voltage on the IIMON pin is
clamped at 1.0V with an accurate internal reference.
Therefore, the input current limit is set at 10A according
to the following formula:
R IS =
ICLIM(MAX) =
0.050V
= 5mΩ
10A
ICLIM(TRKL) = 0.25µA •
24.9kΩ
= 1.25A
5mΩ
• The battery float voltage is set at 10.8V according to
the following formula:
⎞
⎛ 10.8
RBFB1 = ⎜
− 1⎟ • 133kΩ ≈ 1.13MΩ
⎝ 1.136 ⎠
• The bad battery detection time is set at 43 minutes
according to the following formula:
CTMR (nF) = tBADBAT (h) • 138.5 =
15V TO 60V
5mΩ
IN
LT3845
SHDN
ITH
CC
IID IGATE CSP
BAT
127k
LTC4000
FBG
133k
BFB
IIMON
10nF
1.13M
NTC
IBMON
10nF
Si7135DP
CSN
OFB
ENC
CHRG
FLT
3.0V
5mΩ
BGATE
VM
100k
1.15M
47nF
1µF
1.10M
12V, 15A
100µF
14.7k
RST
CLN
IN
Si7135DP
OUT
VC
43
• 138.5 = 100nF
60
TMR
IL
0.1µF
CL
22.1k
10k
GND BIAS
CX
22.1k
10k
10.8V FLOAT
10A MAX CHARGE
CURRENT
3-CELL Li-Ion
BATTERY PACK
1µF
NTHS0603
N02N1002J
4000 F14
Figure 14. 48V to 10.8V at 10A Buck Converter Charger for Three LiFePO4 Cells
4000f
27
LTC4000
Applications Information
• The charge termination time is set at 2.9 hours according to the following formula:
CTMR (nF) = t TERMINATE (h) • 34.6 = 2.9 • 34.6 = 100nF
• The C/X current termination level is programmed at 1A
according to the following formula:
RCX =
(1A • 5mΩ) + 0.5mV ≈ 22.1kΩ
0.25µA
Note that in this particular solution, the timer termination is selected since a capacitor connects to the TMR
pin. Therefore, this C/X current termination level only
applies to the CHRG indicator pin.
• The output voltage regulation level is set at 12V according to the following formula:
⎞
⎛ 12
ROFB1 = ⎜
− 1⎟ • 127kΩ ≈ 1.15MΩ
⎝ 1.193 ⎠
• The instant-on voltage level is consequently set at 9.79V
according to the following formula:
VINST _ ON =
• The range of valid temperature for charging is set at
–1.5°C to 41.5°C by picking a 10k Vishay Curve 2 NTC
thermistor that is thermally coupled to the battery, and
connecting this in series with a regular 10k resistor to
the BIAS pin.
• For compensation, the procedure described in the
empirical loop compensation section is followed. As
recommended, first a 1µF CC and 10k RC is used, which
sets all the loops to be stable. For an example of typical
transient responses, the charge current regulation loop
when VOFB is regulated to VOUT(INST_ON) is used here.
Figure 15 shows the recommended setup to inject a
DC-coupled charge current variation into this particular
loop. The input to the CL pin is a square wave at 70Hz
with the low level set at 120mV and the high level set
at 130mV, corresponding to a 1.2A and 1.3A charge
current (100mA charge current step). Therefore, in this
particular example the trickle charge current regulation
stability is examined. Note that the nominal trickle
charge current in this example is programmed at 1.25A
(RCL = 24.9kΩ).
1150kΩ + 127kΩ
• 0.974V = 9.79V
127kΩ
The worst-case power dissipation during instant-on
operation can be calculated as follows:
• During trickle charging:
B
A
10k
1500pF
1k
LTC4000
PTRKL = [0.86 • VFLOAT – VBAT ] • ICLIM _ TRKL
= [0.86 • 10.8] • 1A
= 9.3W
IBMON
0.015µF
CL
SQUARE WAVE
GENERATOR
f = 60Hz
4000 F15
• And beyond trickle charging:
PINST _ ON = [0.86 • VFLOAT – VBAT ] • ICLIM
= [0.86 • 10.8 – 7.33] • 10A
= 19.3W
Figure 15. Charge Current Regulation Loop Compensation Setup
Therefore, depending on the layout and heat sink available to the charging PMOS, the suggested PMOS over
temperature detection circuit included in Figure 7 may
need to be included. For the complete application circuit,
please refer to Figure 25.
4000f
28
LTC4000
Applications Information
With CC = 1µF, RC = 10k at VIN = 20V, VBAT = 7V, VCSP
regulated at 9.8V and a 0.2A output load condition at
CSP, the transient response for a 100mA charge current
step observed at IBMON is shown in Figure 16.
10
15
5
10
0
5
VIBMON (mV)
5mV/DIV
VIBMON (mV)
5mV/DIV
15
The transient response now indicates an overall under
damped system. As noted in the empirical loop compensation section, the value of RC is now increased iteratively
until RC = 20k. The transient response of the same loop
with CC = 22nF and RC = 20k is shown in Figure 18.
–5
–10
–15
–20 –15 –10
5
0
5ms/DIV
–5
10
15
20
–15
–20 –15 –10
Figure 16. Transient Response of Charge Current Regulation Loop
Observed at IBMON When VOFB is Regulated to VOUT(INST_ON) with
CC = 1µF, RC = 10k for a 100mA Charge Current Step
The transient response shows a small overshoot with slow
settling indicating a fast minor loop within a well damped
overall loop. Therefore, the value of CC is reduced iteratively until CC = 22nF. The transient response of the same
loop with CC = 22nF and RC = 10k is shown in Figure 17.
15
VIBMON (mV)
5mV/DIV
10
5
0
–5
–10
–5
5
0
5ms/DIV
10
15
20
–5
–10
25
4000 F16
–15
–20 –15 –10
0
25
4000 F17
Figure 17. Transient Response of Charge Current Regulation Loop
Observed at IBMON When VOFB is Regulated to VOUT(INST_ON) with
CC = 22nF, RC = 10k for a 100mA Charge Current Step
–5
5
0
5ms/DIV
10
15
20
25
4000 F18
Figure 18. Transient Response of Charge Current Regulation Loop
Observed at IBMON When VOFB is Regulated to VOUT(INST_ON) with
CC = 22nF, RC = 20k for a 100mA Charge Current Step
Note that the transient response is close to optimum
with some overshoot and fast settling. If after iteratively
increasing the value of RC, the transient response again
indicates an over damped system, the step of reducing
CC can be repeated. These steps of reducing CC followed
by increasing RC can be repeated continuously until one
arrives at a stable loop with the smallest value of CC and
the largest value of RC. In this particular example, these
values are found to be CC = 22nF and RC = 20kΩ.
After arriving at these final values of RC and CC, the stability
margin is checked by varying the values of both RC and
CC by 2:1 in all four possible combinations. After which
the setup condition is varied, including varying the input
voltage level and the output load level and the transient
response is checked at these different setup conditions.
Once the desired responses on all different conditions are
obtained, the values of RC and CC are noted.
4000f
29
LTC4000
Applications Information
This same procedure is then repeated for the other four
loops: the input current regulation, the output voltage
regulation, the battery float voltage regulation and finally
the charge current regulation when VOFB > VOUT(INST_ON).
Note that the resulting optimum values for each of the loops
may differ slightly. The final values of CC and RC are then
selected by combining the results and ensuring the most
conservative response for all the loops. This usually entails
picking the largest value of CC and the smallest value of
RC based on the results obtained for all the loops. In this
particular example, the value of CC is finally set to 47nF
and RC = 14.7kΩ.
Board Layout Considerations
In the majority of applications, the most important parameter of the system is the battery float voltage. Therefore,
the user needs to be extra careful when placing and routing
the feedback resistor RBFB1 and RBFB2. In particular, the
battery sense line connected to RBFB1 and the ground return
line for the LTC4000 must be Kelvined back to where the
battery output and the battery ground are located respectively. Figure 19 shows this Kelvin sense configuration.
For accurate current sensing, the sense lines from RIS
and RCS (Figure 19) must be Kelvined back all the way
to the sense resistors terminals. The two sense lines of
each resistor must also be routed close together and away
from noise sources to minimize error. Furthermore, current filtering capacitors should be placed strategically to
ensure that very little AC current is flowing through these
sense resistors as mentioned in the applications section.
The decoupling capacitors CIN and CBIAS must be placed
as close to the LTC4000 as possible. This allows as short
a route as possible from CIN to the IN and GND pins, as
well as from CBIAS to the BIAS and GND pins.
In a typical application, the LTC4000 is paired with an
external DC/DC converter. The operation of this converter
often involves high dV/dt switching voltage as well as high
currents. Isolate these switching voltages and currents
from the LTC4000 section of the board as much as possible by using good board layout practices. These include
separating noisy power and signal grounds, having a good
low impedance ground plane, shielding whenever necessary, and routing sensitive signals as short as possible
and away from noisy sections of the board.
SWITCHING
CONVERTER
GND
ITH
SYSTEM LOAD
RC
CC
ITH
CC
CLN
RIS
IID IGATE
CSP
RCS
LTC4000
IN
CSN
BGATE
BAT
RBFB1
BFB
VIN
RBFB2
GND
FBG
4000 F19
Figure 19. Kelvin Sense Lines Configuration for LTC4000
4000f
30
LTC4000
Applications Information
Appendix—The Loop Transfer Functions
The Input Current Regulation Loop
When a series resistor (RC) and capacitor (CC) is used
as the compensation network as shown in Figure 11, the
transfer function from the input of A4-A7 to the ITH pin
is simply as follows:
The feedback signal for the input current regulation loop
is the sense voltage across the input current sense resistor (RIS).
where gm4-7 is the transconductance of error amplifier A4A7, typically 0.5mA/V; gm10 is the output amplifier (A10)
transconductance, RO4-7 is the output impedance of the
error amplifier, typically 50mΩ; and RO10 is the effective
output impedance of the output amplifier, typically 10mΩ
with the ITH pin open circuit.
Note this simplification is valid when gm10 • RO10 • RO4-7
• CC = AV10 • RO4-7 • CC is much larger than any other
poles or zeroes in the system. Typically AV10 • RO4-7 = 5 •
1010 with the ITH pin open circuit. The exact value of gm10
and RO10 depends on the pull-up current and impedance
connected to the ITH pin respectively.
This voltage is amplified by a factor of 20 and compared
to the voltage on the IL pin by the transconductance error amplifier (A4). This amplifier then drives the output
transconductance amplifier (A10) to appropriately adjust
the voltage on the ITH pin driving the external DC/DC
converter to regulate the input current across the sense
resistor (RIS). This loop is shown in detail in Figure 20.
The simplified loop transmission is:
⎤
⎡⎛
1 ⎞
⎢ ⎜RC –
⎟ CCs + 1⎥
gm10 ⎠
⎝
⎥•
LIC (s) = gm4 ⎢
⎥
⎢
CCs
⎥
⎢
⎦
⎣
⎤
⎡ 20R (R2
IS
CIIMON s + 1)
⎥ • Gmip (s)
⎢
⎢⎣ (R1+ R2) CIIMONs + 1 ⎥⎦
where Gmip(s) is the transfer function from VITH to the
input current of the external DC/DC converter.
In most applications, compensation of the loops involves
picking the right values of RC and CC. Aside from picking
the values of RC and CC, the value of gm10 may also be
adjusted. The value of gm10 can be adjusted higher by
increasing the pull-up current into the ITH pin and its
value can be approximated as:
gm10
I + 5µA
= ITH
50mV
The higher the value of gm10, the smaller the lower limit
of the value of RC would be. This lower limit is to prevent
the presence of the right half plane zero.
Even though all the loops share this transfer function from
the error amplifier input to the ITH pin, each of the loops
has a slightly different dynamic due to differences in the
feedback signal path.
RIS
IN
IIN
CIN
IIMON
R2
20k
IN
CLN
+
–
⎤
⎡⎛
1 ⎞
s
+
1
C
⎥
⎢ ⎜RC –
⎟ C
gm10 ⎠
VITH
⎝
⎥
⎢
(s) = gm4-7
⎥
⎢
RO4-7 • CCs
VFB
⎥
⎢
⎦
⎣
A8
gm8 = 0.33m
BIAS
CIIMON
R1
60k
LTC4000
A4
gm4 = 0.5m
50µA
1V
IL
+
–
–
RO4
CC
A10
gm10 = 0.1m
–
+
CC
RC
ITH
RO10
RIL
4000 F20
Figure 20. Simplified Linear Model of the Input Current
Regulation Loop
4000f
31
LTC4000
Applications Information
The Output Voltage Regulation Loop
The Battery Float Voltage Regulation Loop
The feedback signal for the output voltage regulation loop
is the voltage on the OFB pin, which is connected to the
center node of the resistor divider between the output
voltage (connected to CSP) and the FBG pin. This voltage
is compared to an internal reference (1.193V typical) by
the transconductance error amplifier A7. This amplifier
then drives the output transconductance amplifier (A10)
to appropriately adjust the voltage on the ITH pin driving
the external DC/DC converter to regulate the output voltage observed by the OFB pin. This loop is shown in detail
in Figure 21.
The battery float voltage regulation loop is very similar to
the output float voltage regulation loop. Instead of observing the voltage at the OFB pin, the battery float voltage
regulation loop observes the voltage at the BFB pin.
LTC4000
INPUT
CC
A10
gm10 = 0.1m
–
+
ITH
RO10
A7
gm7 = 0.5m
RO7
CC
RC
+
–
+
–
Gmop(s)
LOAD
CSP
ROFB1
OFB
1.193V
CL
RL
One significant difference is that while the value of RL
in the output voltage loop can vary significantly, the
output resistance of the battery float voltage loop is a
small constant value approximately equal to the sum
of the on-resistance of the external PFET (RDS(ON)) and
the series internal resistance of the battery (RBAT). This
approximation is valid for any efficient system such that
most of the output power from the battery is delivered to
the system load and not dissipated on the battery internal resistance or the charging PFET on-resistance. For a
typical system, minimum RL is at least five times larger
than RDS(ON) + RBAT and RBFB is at least 106 times larger
than RBAT. Figure 22 shows the detail of the battery float
voltage regulation loop.
LTC4000
ROFB2
FBG
–
+
INTERNALLY
PULLED HIGH
4000 F21
⎡⎛
⎤
1 ⎞
⎢ ⎜R C –
⎟ CCs + 1⎥
gm10 ⎠
⎝
⎥•
L OV (s) = gm7 ⎢
⎢
⎥
CCs
⎢
⎥
⎣
⎦
RO6
CC
RC
ITH
RO10
A7
gm6 = 0.5m
Figure 21. Simplified Linear Model of the Output Voltage
Regulation Loop
The simplified loop transmission is as follows:
INPUT
CC
A10
gm10 = 0.1m
+
–
+
–
Gmop(s)
LOAD
BAT
RBFB1
BFB
RCS
CL
RL
RDS(ON)
1.136V
RBFB2
FBG
RBAT
INTERNALLY
PULLED HIGH
4000 F22
Figure 22. Simplified Linear Model of the Battery Float
Voltage Regulation Loop
⎡R
⎤ ⎡
⎤
RL
OFB2 •
⎢
⎥ ⎢
⎥ • Gmo p (s)
⎣ R OFB ⎦ ⎣ RL • CL s + 1⎦
where Gmop(s) is the transfer function from VITH to
the output current of the external DC/DC converter, and
ROFB = ROFB1 + ROFB2.
4000f
32
LTC4000
Applications Information
In Figure 22 the battery is approximated to be a signal
ground in series with the internal battery resistance RBAT.
Therefore, the simplified loop transmission is as follows:
⎤
⎡⎛
1 ⎞
⎢ ⎜RC –
⎟ CCs + 1⎥
gm10 ⎠
⎝
⎥•
LBV (s) = gm6 ⎢
⎥
⎢
CCs
⎥
⎢
⎦
⎣
⎡RBFB2 ⎤ ⎡
⎤
RLB
⎢
⎥•⎢
⎥ • Gmop (s)
⎣ RBFB ⎦ ⎣RLB • CL s + 1⎦
where Gmop(s) is the transfer function from VITH to
the output current of the external DC/DC converter,
RBFB = RBFB1 + RBFB2, and RLB = RL//(RDS(ON) + RCS +
RBAT) represents the effective output resistance from the
LOAD node to GND.
The Battery Charge Current Regulation Loop
The simplified loop transmission is:
⎤
⎡⎛
1 ⎞
⎢ ⎜RC –
⎟ CCs + 1⎥
gm10 ⎠
⎝
⎥•
LCC (s) = gm5 ⎢
⎥
⎢
CCs
⎥
⎢
⎦
⎣
⎡ (R2 • CIBMONs + 1) ⎤
RL
20RCS • ⎢
•
⎥•
⎢⎣ (R1+ R2) CIBMONs + 1⎥⎦ Rf + RL
⎤
⎡
1
⎥ • Gmop (s)
⎢
⎢⎣ (RL  Rf ) CL s + 1⎥⎦

where Gmop(s) is the transfer function from VITH to
the output current of the external DC/DC converter, Rf =
RCS + RDS(ON) + RBAT, and RL//Rf represents the effective
resistance value resulting from the parallel combination
of RL and Rf.
CC
RC
RO10
CC
ITH
LTC4000
INPUT
A10
gm10 = 0.1m
+
–
RO5
A5
gm5 = 0.5m
+
–
–
Due to the presence of the instant-on feature, description
of the charge current regulation loop has to be divided into
two separate operating regions. These regions of operation
depend on whether the voltage on the OFB pin is higher
or lower than the instant-on threshold (VOUT(INST_ON)).
In this operating region, the external charging PFET’s gate
is driven low and clamped at VBGATE(ON). The detail of this
loop is shown in Figure 23.
–
+
This final regulation loop combines certain dynamic characteristics that are found in all the other three loops. The
feedback signal for this charge current regulation loop is
the sense voltage across the charge current sense resistor (RCS). This voltage is amplified by a factor of 20 and
compared to the voltage on the CL pin by the transconductance error amplifier (A5). In a familiar fashion, this
amplifier drives the output transconductance amplifier
(A10) to appropriately adjust the voltage on the ITH pin
driving the external DC/DC converter to regulate the input
current across the sense resistor (RCS).
The Battery Charge Current Regulation Loop when
VOFB > VOUT(INST_ON)
BIAS
A8
gm8 = 0.33m
1V
50µA/
5µA
R1
60k
R2
20k
+
–
Gmop(s)
RL
CL
CSP
RCS
CSN
RDS(ON)
CL
RCL
IBMON
CIBMON
BAT
4000 F23
RBAT
Figure 23. Simplified Linear Model of the Charge Current
Regulation Loop with the External Charging PFET Driven On
4000f
33
LTC4000
Applications Information
The Battery Charge Current Regulation Loop when
VOFB is Regulated to VOUT(INST_ON)
The simplified loop transmission is:
When the battery voltage is below the instant-on level,
the external charging PFET is driven linearly to regulate
the voltage at the output (connected to the CSP pin). The
output voltage is regulated such that the voltage at the OFB
pin is equal to the instant-on threshold (VOUT(INST_ON)).
If this external PFET regulation is fast compared to the
unity crossover frequency of the battery charge current
regulation loop, then the voltage at the output can be considered a small signal ground. However, in the LTC4000
the external PFET regulation is purposely made slow to
allow for a broader selection of possible PFETs to be used.
Therefore, the linear model of the PFET has to be included
in the analysis of the charge current regulation loop. The
detail of this loop is shown in Figure 24.
CC
RC
RO10
CC
ITH
LTC4000
–
+
+
–
RO5
A5
gm5 = 0.5m
+
–
–
BIAS
A8
gm8 = 0.33m
1V
50µA/
5µA
R1
60k
R2
20k
+
–
CL
RCL
IBMON
CIBMON
⎡
⎤
1
•⎢
⎥•
R fIDC + RL ⎢⎣ (RL  R fIDC ) CL s + 1⎥⎦
RL
⎡
⎤
Cg
⎢
⎥
s+1
gmEXT
⎢
⎥
• Gmo p (s)
⎢⎛R + R
⎥
⎞ Cg
BAT
CS
⎢⎜
s + 1⎥
⎟
⎢⎣ ⎝ R fIDC ⎠ gmEXT
⎥⎦

RL
RCS
CSN
Cg
CL
Gmop(s)
CSP
⎡ (R2 • C
⎤
IBMONs + 1)
20R CS • ⎢
⎥•
⎢⎣ (R1+ R2 ) CIBMONs + 1⎥⎦
where Gmop(s) is the transfer function from VITH to the
output current of the external DC/DC converter, gmEXT is
the small signal transconductance of the output PFET,
RflDC = RCS + 1/gmEXT + RBAT and RL//RflDC represents
the effective resistance value resulting from the parallel
combination of RL and RflDC.
INPUT
A10
gm10 = 0.1m
⎡⎛
⎤
1 ⎞
⎢ ⎜R C –
⎟ CCs + 1⎥
gm10 ⎠
⎝
⎥•
L CC2 (s) = gm5 ⎢
⎢
⎥
CCs
⎢
⎥
⎣
⎦
1
gmEXT
BAT
4000 F24
RBAT
Figure 24. Simplified Linear Model of the Charge
Current Regulation Loop with the External Charging
PFET Linearly Regulated
4000f
34
LTC4000
Typical Applications
VIN
15V TO 60V
10A MAX
5mΩ
2.2µF
BSC123NO8NS3
47µF
VIN
TG
WÜRTH ELEKTRONIC
74435561100
10µH
SW
BG
CSS
B160
1.5nF
Si7135DP
3mΩ
BSC123NO8NS3
0.1µF
BURST_EN
SYNC
VOUT
12V, 15A
33µF
×3
BOOST
BAS521
SGND
1.15M
VCC
1µF
LT3845A
1N4148
182k
VFB
fSET
49.9k
16.2k
SENSE+
5mΩ
–
SHDN VC SENSE
14.7k
RST
ITH
NTHS0603
N02N1002J
47nF
CC
IID IGATE CSP
CSN
BGATE
CLN
IN
1.10M
10nF
1µF
BAT
OFB
VM
100k
BZX84C3VO
LTC4000
ENC
CHRG
FLT
127k
FBG
133k
IIMON
10nF
IBMON
10nF
Si7135DP
IL
CL
CX
TMR
24.9k
0.1µF
BFB
NTC
GND BIAS
22.1k
1µF
162k
10k
10k
IN+
1.13M
RNTC
2N7002L
V+ OUT
NTHS0603
N02N1002J
IN–
20k
LTC1540
HYST
3-CELL
LiFePO4
BATTERY
PACK
38.3k
REF
V–
GND
1M
4000 F25
Figure 25. 48V to 10.8V at 10A Buck Converter 3-Cell LiFePO4 Battery Charger with 2.9h Termination Timer,
1.25A Trickle Charge Current and Charging PFET Thermal Protection
4000f
35
LTC4000
Typical Applications
VIN
6V TO 18V
15A MAX
3.3mΩ
2.5mΩ
150µF
22µF
×4
PA1494.362NL
3.3µH
10Ω
1nF
10Ω
INTVCC
Si7135DP
BSC027N04
SENSE+
22µF
×5
BOOST
INTVCC
PLLINMODE
100k
INTVCC
LTC3786
RUN
SS
0.1µF
FREQ
0.1µF
150µF
SW
BG
PGOOD
4.7µF
1.87M
BSC027N04
TG
VBIAS
232k
VFB
GND
VOUT
22V, 5A
BAS140W
SENSE–
ITH
12.1k
10mΩ
28.7k
22nF
RST
CLN
IN
ITH
IID
CC
IGATE CSP
CSN
BGATE
Si7135DP
10nF
1µF
383k
VM
100k
BAT
OFB
LTC4000
ENC
CHRG
FLT
107k
FBG
107k
IIMON
10nF
IBMON
10nF
CX
IL CL
GND BIAS
TMR
BFB
NTC
4000 F26
22.1k
10µF
10k
22.1k
1µF
NTHS0603
N02N1002J
1.87M
RNTC
5-CELL
Li-Ion
BATTERY
PACK
TENERGY
SSIP PACK
30104
Figure 26. 6V to 21V at 5A Boost Converter 5-Cell Li-Ion Battery Charger with C/10 Termination and 0.55A Trickle Charge Current
4000f
36
VIN
18V TO 72V
2.2µF
×2
0.1µF
15.8k
221k
FS
VCC
681Ω
13.6k
0.04Ω VCC
150pF
ISO1
PS2801-1-K
BAS516
3.01k
68Ω
BAS516
•
•
•
BAS516
1.5k
VOUT
PDS1040
VOUT
10nF
24.9k
47nF
VM
150k
IN
10nF
IIMON IBMON IL CL
ENC
CHRG
FLT
RST
ITH
CC
100k
100µF
×3
22.1k
TMR
100nF
LTC4000
CLN
1µF
20mΩ
CX
NTC
FBG
BFB
OFB
BAT
BGATE
CSN
CSP
IGATE
GND BIAS
22.1k
IID
SiA923EDJ
Figure 27. 18V to 72VIN to 4.2V at 2.0A Isolated Flyback Single-Cell Li-Ion Battery Charger
with 2.9h Timer Termination and 0.22A Trickle Charge Current
SYNC GND FB OC
ITH
ISENSE
FDC2512
1µF
GATE
MMBTA42
VCC
LTC3805-5
75k
SSFLT
RUN
PDZ6.8B
6.8V
221k
TR1
PA1277NL
1µF
10k
115k
115k
309k
RNTC
309k
10nF
4000 F27
NTHS0603
N02N1002J
SINGLE-CELL
Li-Ion
BATTERY
PACK
SiA923EDJ
25mΩ
VSYS
4.4V, 2.5A
LTC4000
Typical Applications
4000f
37
LTC4000
Package Description
UFD Package
28-Lead Plastic QFN (4mm × 5mm)
(Reference LTC DWG # 05-08-1712 Rev B)
0.70 ±0.05
4.50 ± 0.05
3.10 ± 0.05
2.50 REF
2.65 ± 0.05
3.65 ± 0.05
PACKAGE OUTLINE
0.25 ±0.05
0.50 BSC
3.50 REF
4.10 ± 0.05
5.50 ± 0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
4.00 ± 0.10
(2 SIDES)
0.75 ± 0.05
R = 0.05
TYP
PIN 1 NOTCH
R = 0.20 OR 0.35
× 45° CHAMFER
2.50 REF
R = 0.115
TYP
27
28
0.40 ± 0.10
PIN 1
TOP MARK
(NOTE 6)
1
2
5.00 ± 0.10
(2 SIDES)
3.50 REF
3.65 ± 0.10
2.65 ± 0.10
(UFD28) QFN 0506 REV B
0.200 REF
0.00 – 0.05
0.25 ± 0.05
0.50 BSC
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE A JEDEC PACKAGE OUTLINE MO-220 VARIATION (WXXX-X).
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON THE TOP AND BOTTOM OF PACKAGE
4000f
38
LTC4000
Package Description
GN Package
28-Lead Plastic SSOP (Narrow .150 Inch)
(Reference LTC DWG # 05-08-1641)
.386 – .393*
(9.804 – 9.982)
.045 ± .005
28 27 26 25 24 23 22 21 20 19 18 17 1615
.254 MIN
.033
(0.838)
REF
.150 – .165
.229 – .244
(5.817 – 6.198)
.0165 ± .0015
.150 – .157**
(3.810 – 3.988)
.0250 BSC
1
RECOMMENDED SOLDER PAD LAYOUT
.015 ± .004
× 45°
(0.38 ± 0.10)
.0075 – .0098
(0.19 – 0.25)
2 3
4
5 6
7
8
9 10 11 12 13 14
.0532 – .0688
(1.35 – 1.75)
.004 – .0098
(0.102 – 0.249)
0° – 8° TYP
.016 – .050
(0.406 – 1.270)
NOTE:
1. CONTROLLING DIMENSION: INCHES
INCHES
2. DIMENSIONS ARE IN
(MILLIMETERS)
.008 – .012
(0.203 – 0.305)
TYP
.0250
(0.635)
BSC
GN28 (SSOP) 0204
3. DRAWING NOT TO SCALE
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
4000f
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
39
LTC4000
Typical Application
5.6Ω
IHLP6767GZ
ER4R7M01
4.7µH
390pF
3.6Ω
B240A
VIN
6V TO 36V
12.5A MAX
4mΩ
Q2
270µF
Q4
B240A
Q5
0.22µF
TG1
SW1
0.01Ω
Q3
Si7135DP
1.24k
BG1 SENSE+ SENSE– BG2
SW2
BOOST1
VOUT
15V, 5A
330µF
×2
0.22µF
0.01Ω
1.24k
3.3µF
×5
1800pF
22µF
×2
TG2
BOOST2
DFLS160
DFLS160
INTVCC
INTVCC
10µF
MODE/PLLIN
VIN
1µF
100k
LTC3789
VINSNS
PGOOD
309k
IOSENSE+
IOSENSE–
VOUTSNS
BZT52C5V6
121k
FREQ
EXTVCC
ILIM
154k
VFB
RUN
ITH
SS
SGND
10µF
PGND1
8.06k
0.01µF
10mΩ
14.7k
RST
CLN
IN
ITH
100nF
CC
IID IGATE CSP
CSN
BGATE
VM
100k
Q2: SiR422DP
Q3: SiR496DP
Q4: SiR4840BDY
Q5: SiR496DP
BAT
OFB
LTC4000
ENC
CHRG
FLT
BZX84C3VO
26.7k
FBG
118k
IIMON
10nF
Si7135DP
10nF
1µF
365k
IBMON
IL
10nF
CL
TMR
CX
BFB
NTC
GND BIAS
1.37M
4-CELL
LiFePO4
BATTERY
PACK
4000 F28
18.2k
0.1µF
22.1k
1µF
RNTC
10k
NTHS0603
N02N1002J
Figure 28. 6V to 36VIN to 14.4V at 4.5A Buck Boost Converter 4-Cell LiFePO4 Battery Charger
with 2.9h Timer Termination and 0.45A Trickle Charge Current
Related Parts
PART NUMBER
DESCRIPTION
COMMENTS
LTC3789
High Efficiency, Synchronous, 4 Switch Buck-Boost Controller
Improved LTC3780 with More Features
LT3845
High Voltage Synchronous Current Mode Step-Down Controller with
Adjustable Operating Frequency
For Medium/High Power, High Efficiency Supplies
LT3650
High Voltage 2A Monolithic Li-Ion Battery Charger
3mm × 3mm DFN-12 and MSOP-12 Packages
LT3651
High Voltage 4A Monolithic Li-Ion Battery Charger
4A Synchronous Version of LT3650 Family
LT3652/LT3652HV
Power Tracking 2A Battery Chargers
Multi-Chemistry, Onboard Termination
LTC4009
High Efficiency, Multi-Chemistry Battery Charger
Low Cost Version of LTC4008, 4mm × 4mm QFN-20
LTC4012
High Efficiency, Multi-Chemistry Battery Charger with PowerPath Control Similar to LTC4009 Adding PowerPath Control
LT3741
High Power, Constant Current, Constant Voltage, Step-Down Controller
Thermally Enhanced 4mm × 4mm QFN and 20-Pin TSSOP
4000f
40 Linear Technology Corporation
LT 0411 • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
 LINEAR TECHNOLOGY CORPORATION 2011
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