LT1370 500kHz High Efficiency 6A Switching Regulator U DESCRIPTION FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LT ®1370 is a monolithic high frequency current mode switching regulator. It can be operated in all standard switching configurations including boost, buck, flyback, forward, inverting and “Cuk.” A 6A high efficiency switch is included on the die, along with all oscillator, control and protection circuitry. Faster Switching with Increased Efficiency Uses Small Inductors: 4.7µH All Surface Mount Components Low Minimum Supply Voltage: 2.7V Quiescent Current: 4.5mA Typ Current Limited Power Switch: 6A Regulates Positive or Negative Outputs Shutdown Supply Current: 12µA Typ Easy External Synchronization Switch Resistance: 0.065Ω Typ The LT1370 typically consumes only 4.5mA quiescent current and has higher efficiency than previous parts. High frequency switching allows for very small inductors to be used. U APPLICATIONS ■ ■ ■ ■ New design techniques increase flexibility and maintain ease of use. Switching is easily synchronized to an external logic level source. A logic low on the Shutdown pin reduces supply current to 12µA. Unique error amplifier circuitry can regulate positive or negative output voltage while maintaining simple frequency compensation techniques. Nonlinear error amplifier transconductance reduces output overshoot on start-up or overload recovery. Oscillator frequency shifting protects external components during overload conditions. Boost Regulators Laptop Computer Supplies Multiple Output Flyback Supplies Inverting Supplies , LTC and LT are registered trademarks of Linear Technology Corporation. U TYPICAL APPLICATION 5V to 12V Boost Converter L1* VIN OFF ON S/S VOUT† 12V VSW R1 53.6k 1% FB + LT1370 + C1** 22µF 25V GND C2 0.047µF R3 2k VC R2 6.19k 1% C4** 22µF 25V ×2 *COILTRONICS UP2-4R7 (4.7µH) UP4-100 (10µH) **AVX TPSD226M025R0200 † MAX IOUT L1 IOUT 4.7µH 1.8A 10µH 2A C3 0.0047µF LT1370 • TA01 VIN = 5V L = 10µH 90 EFFICIENCY (%) 5V 12V Output Efficiency 92 D1 MBRD835L 88 86 84 82 80 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 LOAD CURRENT (A) LT1370 • TA02 1 LT1370 W W W AXI U U ABSOLUTE RATI GS Supply Voltage ....................................................... 30V Switch Voltage LT1370 ............................................................... 35V LT1370HV .......................................................... 42V S/S, SHDN, SYNC Pin Voltage ................................ 30V Feedback Pin Voltage (Transient, 10ms) .............. ±10V Feedback Pin Current ........................................... 10mA Negative Feedback Pin Voltage (Transient, 10ms) ............................................. ±10V Operating Ambient Temperature Range ...... 0°C to 70°C Operating Junction Temperature Range Commercial .......................................... 0°C to 125°C Industrial ......................................... – 40°C to 125°C Storage Temperature Range ................ – 65°C to 150°C Lead Temperature (Soldering, 10 sec)................. 300°C U W U PACKAGE/ORDER I FOR ATIO FRONT VIEW 7 6 5 4 3 2 1 TAB IS GND VIN NFB VSW GND S/S FB VC R PACKAGE 7-LEAD PLASTIC DD ORDER PART NUMBER LT1370CR LT1370HVCR LT1370IR LT1370HVIR FRONT VIEW 7 6 5 4 3 2 1 TAB IS GND TJMAX = 125°C, θJA = 30°C/W, θJC = 4°C/W VIN NFB VSW GND S/S FB VC T7 PACKAGE 7-LEAD TO-220 ORDER PART NUMBER LT1370CT7 LT1370HVCT7 LT1370IT7 LT1370HVIT7 TJMAX = 125°C, θJA = 50°C/W, θJC = 4°C/W WITH PACKAGE SOLDERED TO 0.5 INCH2 COPPER AREA OVER BACKSIDE GROUND PLANE OR INTERNAL POWER PLANE. θJA CAN VARY FROM 20°C/W TO > 40°C/W DEPENDING ON MOUNTING TECHNIQUE Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, TA = 25°C unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS VREF Reference Voltage Measured at Feedback Pin VC = 0.8V 1.230 1.225 1.245 1.245 1.260 1.265 V V 250 550 900 nA nA 0.01 0.03 %/V IFB Feedback Input Current ● VFB = VREF ● VNFR INFB gm 2 Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V ● Negative Feedback Reference Voltage Measured at Negative Feedback Pin Feedback Pin Open, VC = 0.8V ● – 2.525 – 2.560 – 2.48 – 2.48 – 2.435 – 2.400 V V Negative Feedback Input Current VNFB = VNFR ● – 45 – 30 – 15 µA Negative Feedback Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V ● 0.01 0.05 %/V Error Amplifier Transconductance ∆IC = ±25µA 1500 1900 2300 µmho µmho ● 1100 700 120 Error Amplifier Source Current VFB = VREF – 150mV, VC = 1.5V ● Error Amplifier Sink Current VFB = VREF + 150mV, VC = 1.5V ● 200 350 µA 1400 2400 µA LT1370 ELECTRICAL CHARACTERISTICS VIN = 5V, VC = 0.6V, VFB = VREF, VSW, S/S and NFB pins open, TA = 25°C unless otherwise noted. SYMBOL AV f PARAMETER CONDITIONS MIN TYP MAX UNITS Error Amplifier Clamp Voltage High Clamp, VFB = 1V 1.5 1.8 2.30 V Low Clamp, VFB = 1.5V 0.2 0.3 0.52 Error Amplifier Voltage Gain VC Pin Threshold Duty Cycle = 0% Switching Frequency 2.7V ≤ VIN ≤ 25V 0°C ≤ TJ ≤ 125°C – 40°C ≤ TJ ≤ 0°C (I-Grade) Maximum Switch Duty Cycle V/ V 0.9 1.1 1.35 V ● 460 440 400 500 500 550 580 580 kHz kHz kHz ● 85 95 35 42 40 Switch Current Limit Blanking Time 130 BV Output Switch Breakdown Voltage LT1370 ● ● LT1370HVC, 0°C ≤ TJ ≤ 125°C LT1370HVI, – 40°C ≤ TJ ≤ 0°C (I-Grade) VSAT Output Switch ON Resistance ISW = 6A ● ILIM Switch Current Limit Duty Cycle = 50% Duty Cycle = 80% (Note 1) ● ∆IIN ∆ISW % 300 ns 44 47 V V V 0.065 0.11 Ω 8 7 10 A A Supply Current Increase During Switch ON Time 22 33 mA/A Control Voltage to Switch Current Transconductance 10 Minimum Input Voltage IQ V 500 6 A/V ● 2.4 2.7 V 4.5 6 mA Supply Current 2.7V ≤ VIN ≤ 25V ● Shutdown Supply Current 2.7V ≤ VIN ≤ 25V, VS/S ≤ 0.6V ● 12 40 µA Shutdown Threshold 2.7V ≤ VIN ≤ 25V ● 0.6 1.3 2 V ● 4 12 25 µs Shutdown Delay S/S Input Current Synchronization Frequency Range The ● denotes specifications which apply over the full operating temperature range. 0V ≤ S/S ≤ 5V ● –7 10 µA ● 600 800 kHz Note 1: For duty cycles (DC) between 45% and 85%, minimum switch current limit is given by ILIM = 2.65(2.7 – DC). 3 LT1370 U W TYPICAL PERFORMANCE CHARACTERISTICS Switch Saturation Voltage vs Switch Current Switch Current Limit vs Duty Cycle 8.2 550 SWITCH CURRENT LIMIT (A) 75°C 400 25°C 350 300 0°C 250 200 150 100 2.8 7.8 7.6 7.4 7.2 6.6 1 0 4 2 3 SWITCH CURRENT (A) 5 2.0 18 1.8 SHUTDOWN DELAY (µs) 14 1.4 12 1.2 10 1.0 0.8 6 0.6 4 0.4 2 0.2 0 –50 –25 0 SHUTDOWN THRESHOLD (V) 1.6 MINIMUM SYNCHRONIZATION VOLTAGE (VP-P) 20 SHUTDOWN DELAY 0 25 50 75 100 125 150 TEMPERATURE (°C) 3.0 2.5 2.0 1.5 1.0 0.5 0 –50 –25 0 –1 –2 –3 –4 2 3 4 VOLTAGE (V) 5 6 7 LT1370 • G07 4 25°C –55°C 200 125°C 100 0 –100 –200 –300 25 50 75 100 125 150 TEMPERATURE (°C) –0.3 VREF –0.2 –0.1 FEEDBACK PIN VOLTAGE (V) Error Amplifier Transconductance vs Temperature 110 2000 100 1800 90 80 70 60 50 40 30 gm = ∆I (VC) ∆V (FB) 1600 1400 1200 1000 800 600 400 200 20 10 0.1 LT1370 • G06 TRANSCONDUCTANCE (µmho) SWITCHING FREQUENCY (% OF TYPICAL) INPUT CURRENT (µA) 0 1 300 Switching Frequency vs Feedback Pin Voltage 2 0 400 LT1370 • G05 S/S Pin Input Current vs Voltage –1 Error Amplifier Output Current vs Feedback Pin Voltage fSYNC = 700kHz LT1370 • G04 1 25 50 75 100 125 150 TEMPERATURE (°C) LT1370 • G03 Minimum Synchronization Voltage vs Temperature SHUTDOWN THRESHOLD 0 LT1370 • G02 Shutdown Delay and Threshold vs Temperature 8 2.2 1.8 –50 –25 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 0 6 LT1370 • G01 16 2.4 2.0 6.8 50 2.6 7.0 ERROR AMPLIFIER OUTPUT CURRENT (µA) SWITCH VOLTAGE (mV) 8.0 125°C 450 3.0 INPUT VOLTAGE (V) 500 0 Minimum Input Voltage vs Temperature 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 FEEDBACK PIN VOLTAGE (V) LT1370 • G08 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1370 • G09 LT1370 U W TYPICAL PERFORMANCE CHARACTERISTICS VC Pin Threshold and High Clamp Voltage vs Temperature Feedback Input Current vs Temperature 2.2 1.8 1.6 1.4 1.2 VC THRESHOLD 0 25 50 75 100 125 150 TEMPERATURE (°C) 700 NEGATIVE FEEDBACK INPUT CURRENT (µA) VC HIGH CLAMP FEEDBACK INPUT CURRENT (nA) VC VOLTAGE (V) 0 800 2.0 1.0 –50 –25 Negative Feedback Input Current vs Temperature VFB =VREF 600 500 400 300 200 100 0 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1370 • G10 LT1370 • G11 VNFB =VNFR –10 –20 –30 –40 –50 –50 –25 0 25 50 75 100 125 150 TEMPERATURE (°C) LT1370 • G12 U U U PIN FUNCTIONS VC: The Compensation pin is used for frequency compensation, current limiting and soft start. It is the output of the error amplifier and the input of the current comparator. Loop frequency compensation can be performed with an RC network connected from the VC pin to ground. See Applications Information. S/S: Shutdown and Synchronization Pin. The S/S pin is logic level compatible. Shutdown is active low and the shutdown threshold is typically 1.3V. For normal operation, pull the S/S pin high, tie it to VIN or leave it floating. To synchronize switching, drive the S/S pin between 600kHz and 800kHz. See Applications Information. FB: The Feedback pin is used for positive output voltage sensing and oscillator frequency shifting. It is the inverting input to the error amplifier. The noninverting input of this amplifier is internally tied to a 1.245V reference. VIN: Bypass Input Supply Pin with a Low ESR Capacitor, 10µF or More. The regulator goes into undervoltage lockout when VIN drops below 2.5V. Undervoltage lockout stops switching and pulls the VC pin low. NFB: The Negative Feedback pin is used for negative output voltage sensing. It is connected to the inverting input of the negative feedback amplifier through a 100k source resistor. VSW: The Switch pin is the collector of the power switch and has large currents flowing through it. Keep the traces to the switching components as short as possible to minimize radiation and voltage spikes. GND: Tie all ground pins to a good quality ground plane. See Applications Information. 5 LT1370 W BLOCK DIAGRAM VIN SHUTDOWN DELAY AND RESET SW LOW DROPOUT 2.3V REG ANTI-SAT S/S SYNC LOGIC OSC DRIVER SWITCH 5:1 FREQUENCY SHIFT + 100k NFB NFBA – COMP 50k – FB + 1.245V REF + EA IA VC GND SENSE AV ≈ 20 0.005Ω – GND LT1370 • BD U OPERATION The LT1370 is a current mode switcher. This means that switch duty cycle is directly controlled by switch current rather than by output voltage. Referring to the block diagram, the switch is turned ON at the start of each oscillator cycle. It is turned OFF when switch current reaches a predetermined level. Control of output voltage is obtained by using the output of a voltage sensing error amplifier to set current trip level. This technique has several advantages. First, it has immediate response to input voltage variations, unlike voltage mode switchers which have notoriously poor line transient response. Second, it reduces the 90° phase shift at midfrequencies in the energy storage inductor. This greatly simplifies closed-loop frequency compensation under widely varying input voltage or output load conditions. Finally, it allows simple pulse-by-pulse current limiting to provide maximum switch protection under output overload or short conditions. A low dropout internal regulator provides a 2.3V supply for all internal circuitry. This low dropout design allows input voltage to vary from 2.7V to 25V with virtually no change in device performance. A 6 500kHz oscillator is the basic clock for all internal timing. It turns on the output switch via the logic and driver circuitry. Special adaptive antisat circuitry detects onset of saturation in the power switch and adjusts driver current instantaneously to limit switch saturation. This minimizes driver dissipation and provides very rapid turn-off of the switch. A 1.245V bandgap reference biases the positive input of the error amplifier. The negative input of the amplifier is brought out for positive output voltage sensing. The error amplifier has nonlinear transconductance to reduce output overshoot on start-up or overload recovery. When the feedback voltage exceeds the reference by 40mV, error amplifier transconductance increases 10 times, which reduces output overshoot. The feedback input also invokes oscillator frequency shifting, which helps protect components during overload conditions. When the feedback voltage drops below 0.6V, the oscillator frequency is reduced 5:1. Lower switching frequency allows full control of switch current limit by reducing minimum switch duty cycle. LT1370 U OPERATION Unique error amplifier circuitry allows the LT1370 to directly regulate negative output voltages. The negative feedback amplifier’s 100k source resistor is brought out for negative output voltage sensing. The NFB pin regulates at – 2.48V while the amplifier output internally drives the FB pin to 1.245V. This architecture, which uses the same main error amplifier, prevents duplicating functions and maintains ease of use. Consult LTC Marketing for units that can regulate down to – 1.25V. The error signal developed at the amplifier output is brought out externally. This pin (VC) has three different functions. It is used for frequency compensation, current limit adjustment and soft starting. During normal regulator operation this pin sits at a voltage between 1V (low output current) and 1.9V (high output current). The error amplifier is a current output (gm) type, so this voltage can be externally clamped for lowering current limit. Likewise, a capacitor coupled external clamp will provide soft start. Switch duty cycle goes to zero if the VC pin is pulled below the control pin threshold, placing the LT1370 in an idle mode. U U W U APPLICATIO S I FOR ATIO VOUT Positive Output Voltage Setting The LT1370 develops a 1.245V reference (VREF) from the FB pin to ground. Output voltage is set by connecting the FB pin to an output resistor divider (Figure 1). The FB pin bias current represents a small error and can usually be ignored for values of R2 up to 7k. The suggested value for R2 is 6.19k. The NFB pin is normally left open for positive output applications. Positive fixed voltage versions are available (consult LTC Marketing). R1 FB PIN R2 Dual Polarity Output Voltage Sensing Certain applications benefit from sensing both positive and negative output voltages. One example is the “Dual Output Flyback Converter with Overvoltage Protection” circuit shown in the Typical Applications section. Each output voltage resistor divider is individually set as described above. When both the FB and NFB pins are used, R1 = R2 VOUT –1 1.245 VREF LT1370 • F01 Figure 1. Positive Output Resistor Divider Negative Output Voltage Setting The LT1370 develops a – 2.48V reference (VNFR) from the NFB pin to ground. Output voltage is set by connecting the NFB pin to an output resistor divider (Figure 2). The –30µA NFB pin bias current (INFB) can cause output voltage errors and should not be ignored. This has been accounted for in the formula in Figure 2. The suggested value for R2 is 2.49k. The FB pin is normally left open for negative output applications. ( ) ( ) VOUT = VREF 1 + R1 R2 –VOUT INFB ( ) R1 –VOUT = VNFB 1 + R1 + INFB (R1) R2 R2 R1 = NFB PIN VNFR VOUT– 2.48 ( )( 2.48 + 30 • 10– 6 R2 ) LT1370 • F02 Figure 2. Negative Output Resistor Divider the LT1370 acts to prevent either output from going beyond its set output voltage. For example, in this application if the positive output were more heavily loaded than the negative, the negative output would be greater and would regulate at the desired set-point voltage. The positive output would sag slightly below its set-point voltage. This technique prevents either output from going unregulated high at no load. 7 LT1370 U U W U APPLICATIO S I FOR ATIO Shutdown and Synchronization The device has a dual function S/S pin which is used for both shutdown and synchronization. This pin is logic level compatible and can be pulled high, tied to VIN or left floating for normal operation. A logic low on the S/S pin activates shutdown, reducing the part’s supply current to 12µA. Typical synchronization range is from 1.05 to 1.8 times the part’s natural switching frequency, but is only guaranteed between 600kHz and 800kHz. A 12µs resetable shutdown delay network guarantees the part will not go into shutdown while receiving a synchronization signal. Caution should be used when synchronizing above 700kHz because at higher sync frequencies the amplitude of the internal slope compensation used to prevent subharmonic switching is reduced. This type of subharmonic switching only occurs when the duty cycle of the switch is above 50%. Higher inductor values will tend to eliminate this problem. Thermal Considerations Care should be taken to ensure that the worst-case input voltage and load current conditions do not cause excessive die temperatures. Typical thermal resistance is 30°C/W for the R package and 50°C/W for the T7 package but these numbers will vary depending on the mounting techniques (copper area, airflow, etc.). Heat is transferred from the package via the tab. Average supply current (including driver current) is: IIN = 4.5mA + DC(ISW/45) ISW = Switch current DC = Switch duty cycle Switch power dissipation is given by: PSW = (ISW)2(RSW)(DC) RSW = Output switch ON resistance Total power dissipation of the die is the sum of supply current times supply voltage, plus switch power: PD(TOTAL) = (IIN)(VIN) + PSW 8 Surface mount heat sinks are available which can lower package thermal resistance by two or three times. One manufacturer, Wakefield Engineering, offers surface mount heat sinks for the R package and can be reached at (617) 245-5900 or at www.wakefield.com. Choosing the Inductor For most applications the inductor will fall in the range of 2.2µH to 22µH. Lower values are chosen to reduce physical size of the inductor. Higher values allow more output current because they reduce peak current seen by the power switch, which has a 6A limit. Higher values also reduce input ripple voltage and reduce core loss. When choosing an inductor you need to consider maximum load current, core and copper losses, allowable component height, output voltage ripple, EMI, fault current in the inductor, saturation and, of course, cost. The following procedure is suggested as a way of handling these somewhat complicated and conflicting requirements. 1. Assume that the average inductor current for a boost converter is equal to load current times VOUT / VIN and decide whether or not the inductor must withstand continuous overload conditions. If average inductor current at maximum load current is 3A, for instance, a 3A inductor may not survive a continuous 6A overload condition. Also be aware that boost converters are not short-circuit protected and that, under output short conditions, inductor current is limited only by the available current of the input supply. 2. Calculate peak inductor current at full load current to ensure that the inductor will not saturate. Peak current can be significantly higher than output current, especially with smaller inductors and lighter loads, so don’t omit this step. Powdered iron cores are forgiving because they saturate softly, whereas ferrite cores saturate abruptly and other core materials fall in between. The following formula assumes continuous mode operation but it errs only slightly on the high side for discontinuous mode, so it can be used for all conditions. LT1370 U U W U APPLICATIO S I FOR ATIO ) ) V (V –V ) V IPEAK = (IOUT) OUT + IN OUT IN 2(f)(L)(VOUT) VIN VIN = Minimum input voltage f = 500kHz switching frequency 3. Decide if the design can tolerate an “open” core geometry, like a rod or barrel, which has high magnetic field radiation, or whether it needs a closed core, like a toroid, to prevent EMI problems. One would not want an open core next to a magnetic storage media, for instance! This is a tough decision because the rods or barrels are temptingly cheap and small and there are no helpful guidelines to calculate when the magnetic field radiation will be a problem. 4. Start shopping for an inductor that meets the requirements of core shape, peak current (to avoid saturation), average current (to limit heating) and fault current. If the inductor gets too hot, wire insulation will melt and cause turn-to-turn shorts. Keep in mind that all good things like high efficiency, low profile and high temperature operation will increase cost, sometimes dramatically. 5. After making an initial choice, consider the secondary things like output voltage ripple, second sourcing, etc. Use the experts in the LTC Applications Department if you feel uncertain about the final choice. They have experience with a wide range of inductor types and can tell you about the latest developments in low profile, surface mounting, etc. Output Capacitor The output capacitor is normally chosen by its effective series resistance (ESR), because this is what determines output ripple voltage. At 500kHz any polarized capacitor is essentially resistive. To get low ESR takes volume, so physically smaller capacitors have high ESR. The ESR range needed for typical LT1370 applications is 0.025Ω to 0.2Ω. A typical output capacitor is an AVX type TPS, 22µF at 25V (two each), with a guaranteed ESR less than 0.2Ω. This is a “D” size surface mount solid tantalum capacitor. TPS capacitors are specially constructed and tested for low ESR, so they give the lowest ESR for a given volume. To further reduce ESR, multiple output capacitors can be used in parallel. The value in microfarads is not particularly critical, and values from 22µF to greater than 500µF work well, but you cannot cheat mother nature on ESR. If you find a tiny 22µF solid tantalum capacitor, it will have high ESR and output ripple voltage will be terrible. Table 1 shows some typical solid tantalum surface mount capacitors. Table 1. Surface Mount Solid Tantalum Capacitor ESR and Ripple Current E CASE SIZE ESR (MAX Ω) RIPPLE CURRENT (A) 0.1 to 0.3 0.7 to 0.9 0.7 to 1.1 0.4 0.1 to 0.3 0.9 to 2.0 0.7 to 1.1 0.36 to 0.24 0.2 (Typ) 1.8 to 3.0 0.5 (Typ) 0.22 to 0.17 2.5 to 10 0.16 to 0.08 AVX TPS, Sprague 593D AVX TAJ D CASE SIZE AVX TPS, Sprague 593D AVX TAJ C CASE SIZE AVX TPS AVX TAJ B CASE SIZE AVX TAJ Many engineers have heard that solid tantalum capacitors are prone to failure if they undergo high surge currents. This is historically true and AVX type TPS capacitors are specially tested for surge capability, but surge ruggedness is not a critical issue with the output capacitor. Solid tantalum capacitors fail during very high turn-on surges, which do not occur at the output of regulators. High discharge surges, such as when the regulator output is dead-shorted, do not harm the capacitors. Single inductor boost regulators have large RMS ripple current in the output capacitor, which must be rated to handle the current. The formula to calculate this is: Output Capacitor Ripple Current (RMS) DC IRIPPLE (RMS) = IOUT 1 – DC = IOUT VOUT – VIN VIN DC = Switch duty cycle 9 LT1370 U U W U APPLICATIO S I FOR ATIO Input Capacitors Output Diode The input capacitor of a boost converter is less critical due to the fact that the input current waveform is triangular and does not contain large squarewave currents as is found in the output capacitor. Capacitors in the range of 10µF to 100µF with an ESR of 0.1Ω or less work well up to full 6A switch current. Higher ESR capacitors may be acceptable at low switch currents. Input capacitor ripple current for a boost converter is : The suggested output diode (D1) is a Motorola MBRD835L. It is rated at 8A average forward current and 35V reverse voltage. Typical forward voltage is 0.4V at 3A. The diode conducts current only during switch OFF time. Peak reverse voltage for boost converters is equal to regulator output voltage. Average forward current in normal operation is equal to output current. Frequency Compensation IRIPPLE = 0.3(VIN)(VOUT – VIN) (f)(L)(VOUT) f = 500kHz switching frequency The input capacitor can see a very high surge current when a battery or high capacitance source is connected “live” and solid tantalum capacitors can fail under this condition. Several manufacturers have developed tantalum capacitors specially tested for surge capability (AVX TPS series, for instance) but even these units may fail if the input voltage approaches the maximum voltage rating of the capacitor during a high surge. AVX recommends derating capacitor voltage by 2:1 for high surge applications. Ceramic, OS-CON and aluminum electrolytic capacitors may also be used and have a high tolerance to turn-on surges. Ceramic Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low ESR. Unfortunately, the ESR is so low that it can cause loop stability problems. Solid tantalum capacitor ESR generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually resonate with their ESL before ESR becomes effective. They are appropriate for input bypassing because of their high ripple current ratings and tolerance of turn-on surges. 10 Loop frequency compensation is performed on the output of the error amplifier (VC pin) with a series RC network. The main pole is formed by the series capacitor and the output impedance (≈500kΩ) of the error amplifier. The pole falls in the range of 2Hz to 20Hz. The series resistor creates a “zero” at 1kHz to 5kHz, which improves loop stability and transient response. A second capacitor, typically one-tenth the size of the main compensation capacitor, is sometimes used to reduce the switching frequency ripple on the VC pin. VC pin ripple is caused by output voltage ripple attenuated by the output divider and multiplied by the error amplifier. Without the second capacitor, VC pin ripple is: VC Pin Ripple = 1.245(VRIPPLE)(gm)(RC) (VOUT) VRIPPLE = Output ripple (VP–P) gm = Error amplifier transconductance ( ≈1500µmho) RC = Series resistor on VC pin VOUT = DC output voltage To prevent irregular switching, VC pin ripple should be kept below 50mVP–P. Worst-case VC pin ripple occurs at maximum output load current and will also be increased if poor quality (high ESR) output capacitors are used. The addition of a 0.0047µF capacitor on the VC pin reduces switching frequency ripple to only a few millivolts. A low value for RC will also reduce VC pin ripple, but loop phase margin may be inadequate. LT1370 U U W U APPLICATIO S I FOR ATIO Layout Considerations For maximum efficiency, LT1370 switch rise and fall times are made as short as possible. To prevent radiation and high frequency resonance problems, proper layout of the components connected to the switch node is essential. B field (magnetic) radiation is minimized by keeping output diode, switch pin and output bypass capacitor leads as short as possible. Figure 3 shows recommended positions for these components. E field radiation is kept low by minimizing the length and area of all traces connected to the switch pin. A ground plane should always be used under the switcher circuitry to prevent interplane coupling. The high speed switching current path is shown schematically in Figure 4. Minimum lead length in this path is essential to ensure clean switching and low EMI. The path including the switch, output diode and output capacitor is the only one containing nanosecond rise and fall times. Keep this path as short as possible. More Help For more detailed information on switching regulator circuits, please see Application Note 19. Linear Technology also offers a computer software program, SwitcherCADTM, to assist in designing switching converters. In addition, our Applications Department is always ready to lend a helping hand. SwitcherCAD is a trademark of Linear Technology Corporation. FB VC GND NFB S/S VSW VIN C D C KEEP PATH FROM VSW, OUTPUT DIODE, OUTPUT CAPACITORS AND GROUND RETURN AS SHORT AS POSSIBLE LT1370 • F03 Figure 3. Layout Considerations— R Package L1 SWITCH NODE VOUT VIN HIGH FREQUENCY CIRCULATING PATH LOAD LT1370 • F04 Figure 4 11 LT1370 U TYPICAL APPLICATIONS N Positive-to-Negative Converter with Direct Feedback VIN 2.7V TO 13V + T1* 2 D2 P6KE-15A D3 1N4148 1 • C1 100µF VIN ON OFF VSW S/S 4 + • R2 2.49k 1% NFB VC C3 0.0047µF R3 2.49k 1% GND C2 0.047µF R1 2k –VOUT† –5V 3 D1 MBRD835L LT1370 C4 100µF ×2 *BH ELECTRONICS 501-0726 †MAX I OUT IOUT VIN 1.75A 3V 2.25A 5V 3A 9V LT1370 • TA03 Dual Output Flyback Converter with Overvoltage Protection R1 13k 1% R2 6.19k 1% VIN 2.7V TO 10V + C1 22µF FB OFF ON VIN VSW S/S MBRS360T3 T1* 2, 3 7 + P6KE-20A • 1N4148 8, 9 • LT1370 NFB VC C3 0.0047µF GND C2 0.047µF R3 2k *DALE LPE-5047-100MB 12 •4 10 + 1 MBRS360T3 VOUT 15V C4 47µF C5 47µF –VOUT –15V R4 12.1k 1% R5 2.49k 1% LT1370 • TA04 LT1370 U TYPICAL APPLICATIONS N Two Li-Ion Cells to 5V SEPIC Converter** VIN 4V TO 9V L1A* 6.8µH VIN ON OFF • VSW S/S C1 33µF 20V FB GND • VC C3 100µF 10V ×2 + L1B* 6.8µH R1 2k C4 0.047µF VOUT† 5V R2 18.7k 1% C2 4.7µF LT1370 + D1 MBRD835L R3 6.19k 1% C5 0.0047µF LT1370 • TA05 C1 = AVX TPSD 336M020R0200 C2 = TOKIN 1E475ZY5U-C304 C3 = AVX TPSD107M010R0100 * BH ELECTRONICS 501-0726 ** INPUT VOLTAGE MAY BE GREATER OR LESS THAN OUTPUT VOLTAGE †MAX I OUT IOUT VIN 2A 4V 2.2A 5V 2.6A 7V 2.8A 9V Single Li-Ion Cell to 5V L1* D1 MBRD835L VOUT† 5V VSW R1 18.7k 1% FB + VIN OFF ON S/S LT1370 + SINGLE Li-Ion CELL + C1** 100µF 10V GND VC R2 6.19k 1% C2 0.047µF R3 2k C4** 100µF 10V ×2 C3 0.0047µF LT1370 • TA06 *COILCRAFT DO3316P-472 **AVX TPSD107M010R0100 †MAX I OUT IOUT 2.5A 3A 3.3A VIN 2.7V 3.3V 3.6V 13 LT1370 U TYPICAL APPLICATIONS N Laser Power Supply 0.01µF 5kV 1800pF 10kV 47k 5W 1800pF 10kV 8 11 L1 1 4 5 HV DIODES 3 2 LASER + 2.2µF Q1 0.47µF 150Ω L2 82µH MUR405 VIN 12V TO 25V VSW 10k VIN + Q2 10k FB LT1370 2.2µF VC GND 0.1µF VIN 1N4002 (ALL) 190Ω 1% + 10µF LT1370 • TA07 14 L1 = COILTRONICS CTX02-11128 L2 = GOWANDA GA40-822K Q1, Q2 = ZETEX ZTX849 0.47µF = WIMA 3X 0.15µF TYPE MKP-20 HV DIODES = SEMTECH-FM-50 LASER = HUGHES 3121H-P COILTRONICS (407) 241-7876 LT1370 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. R Package 7-Lead Plastic DD Pak (LTC DWG # 05-08-1462) 0.256 (6.502) 0.060 (1.524) TYP 0.060 (1.524) 0.390 – 0.415 (9.906 – 10.541) 0.165 – 0.180 (4.191 – 4.572) 0.045 – 0.055 (1.143 – 1.397) 15° TYP 0.060 (1.524) 0.183 (4.648) 0.059 (1.499) TYP 0.330 – 0.370 (8.382 – 9.398) ( +0.203 0.102 –0.102 BOTTOM VIEW OF DD PAK HATCHED AREA IS SOLDER PLATED COPPER HEAT SINK ) 0.095 – 0.115 (2.413 – 2.921) 0.075 (1.905) 0.300 (7.620) +0.008 0.004 –0.004 ( +0.012 0.143 –0.020 +0.305 3.632 –0.508 0.040 – 0.060 (1.016 – 1.524) 0.026 – 0.036 (0.660 – 0.914) ) 0.013 – 0.023 (0.330 – 0.584) 0.050 ± 0.012 (1.270 ± 0.305) R (DD7) 0396 T7 Package 7-Lead Plastic TO-220 (Standard) (LTC DWG # 05-08-1422) 0.390 – 0.415 (9.906 – 10.541) 0.165 – 0.180 (4.191 – 4.572) 0.147 – 0.155 (3.734 – 3.937) DIA 0.045 – 0.055 (1.143 – 1.397) 0.230 – 0.270 (5.842 – 6.858) 0.460 – 0.500 (11.684 – 12.700) 0.570 – 0.620 (14.478 – 15.748) 0.330 – 0.370 (8.382 – 9.398) 0.620 (15.75) TYP 0.700 – 0.728 (17.780 – 18.491) 0.152 – 0.202 0.260 – 0.320 (3.860 – 5.130) (6.604 – 8.128) 0.040 – 0.060 (1.016 – 1.524) 0.095 – 0.115 (2.413 – 2.921) 0.013 – 0.023 (0.330 – 0.584) 0.026 – 0.036 (0.660 – 0.914) 0.135 – 0.165 (3.429 – 4.191) 0.155 – 0.195 (3.937 – 4.953) T7 (TO-220) (FORMED) 1197 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1370 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1171 100kHz 2.5A Boost Switching Regulator Good for Up to VIN = 40V LTC 1265 12V 1.2A Monolithic Buck Converter Converts 5V to 3.3V at 1A with 90% Efficiency LT1302 Micropower 2A Boost Converter Converts 2V to 5V at 600mA in SO-8 Packages LT1372 500kHz 1.5A Boost Switching Regulator Also Regulates Negative Flyback Outputs LT1373 Low Supply Current 250kHz 1.5A Boost Switching Regulator 90% Efficient Boost Converter with Constant Frequency LT1374 500kHz 4.5A Buck Switching Regulator Converts 12V to 3.3V at 2.5A in SO-8 Package LT1376 500kHz 1.5A Buck Switching Regulator Steps Down from Up to 25V Using 4.7µH Inductors LT1512 500kHz 1.5A SEPIC Battery Charger Input Voltage May Be Greater or Less Than Battery Voltage LT1513 500kHz 3A SEPIC Battery Charger Input Voltage May Be Greater or Less Than Battery Voltage ® 16 Linear Technology Corporation 1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● (408) 432-1900 FAX: (408) 434-0507● TELEX: 499-3977 ● www.linear-tech.com 1370f LT/TP 0198 4K • PRINTED IN THE USA LINEAR TECHNOLOGY CORPORATION 1998