LTC3802 Dual 550kHz Synchronous 2-Phase DC/DC Controller with Programmable Up/Down Tracking U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC®3802 is a dual switching regulator controller optimized for high efficiency step-down conversion from input voltages between 3V to 30V. The controller uses a leading edge modulation scheme to allow extremely low duty cycle operation. The constant frequency voltage mode controller allows a phase-lockable frequency between 330kHz and 750kHz. Power loss and noise due to the ESR of the input capacitors are minimized by operating the two controller output stages 180° out of phase. The synchronous buck architecture automatically shifts to Burst Mode operation as the output load decreases, ensuring maximum efficiency over a wide range of load currents. Wide VIN Range: 3V to 30V Operation with Line Feedforward Compensation Leading Edge Modulation Architecture for Extremely Low Duty Cycle Operation Phase-Lockable Fixed Frequency: 330kHz to 750kHz Two 180° Out-of-Phase Controllers Fast Programmable Power-Up/-Down Tracking Programmable Current Limit Without External Current Sense Resistor Optional Burst Mode® Operation at Light Load ±1% 0.6V Voltage Reference External N-Channel MOSFET Architecture Low Shutdown Current: <100µA Overvoltage Protection and PGOOD Flag Small 28-Lead SSOP and 32-Lead QFN Packages The LTC3802 features an onboard, trimmed 0.6V reference and provides better than 1% regulation at the converter outputs. A separate output sense provides real time overvoltage protection and PGOOD sensing. An FBT pin programs the power-up/-down tracking between the two channels to meet various sequencing requirements. A RUN/SS pin provides soft-start and externally programmable current limit protection functions. U APPLICATIO S ■ ■ ■ Notebook and Palmtop Computers Portable Instruments Battery-Operated Digital Devices DC Power Distribution Systems , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a registered trademark of Linear Technology Corporation. U.S. Patent Nos 5481178, 5846544, 6304066, 6580258, 5055767, 6307356 U ■ TYPICAL APPLICATIO VIN 5V TO 22V + 56µF 25V ×2 4.7µF 25V ×8 5V CMDSH-3 Si7860DP ×2 1µH VOUT1 3.3V 15A 0.1µF B340B 10k 10k + 330µF 4V ×3 Si7440DP ×2 47k 560pF 2.21k 2.21k 2200pF 1500pF 390Ω 15k 330pF + 10µF CMDSH-3 TG1 PVCC Si7860DP ×2 TG2 BOOST2 SW2 BOOST1 BG2 SW1 PLLIN BG1 PLLLPF PGND LTC3802 IMAX2 IMAX1 FBT CMPIN1 CMPIN2 COMP1 FB1 SGND FCB RUN/SS COMP2 FB2 PHASEMD VINFF PGOOD 1µH VOUT2 2.5V 15A 0.1µF Si7440DP ×2 47k B340B 330µF 4V ×3 + 10k 3.16k 10k 3.16k 5V VCC + 0.1µF 10Ω 10µF 2200pF 15k 560pF 1500pF 390Ω 330pF 3802 TA01 0.1µF 10k 5V VIN 0.47µF 3802f 1 LTC3802 U W W W ABSOLUTE AXI U RATI GS (Note 1) Supply Voltage VCC, PVCC ............................................................. 7V BOOSTn .............................................................. 37V BOOSTn – SWn .................................................... 7V SWn ........................................................ –1V to 30V Input Voltage VINFF .................................................................... 30V FBn, CMPINn, FBT, PLLIN, FCB, RUN/SS, PGOOD, PLLLPF, PHASEMD, EXTREF, IMAXn .......................... –0.3V to VCC + 0.3V Extended Commercial Operating Temperature Range (Note 2) .. –40°C to 85°C Storage Temperature Range LTC3802EGN ................................... – 65°C to 150°C LTC3802EUH ................................... – 65°C to 125°C Lead Temperature (Soldering, 10 sec) LTC3802EGN Only ........................................... 300°C U W U PACKAGE/ORDER I FOR ATIO 25 SW2 5 24 PLLIN 23 PLLLPF 6 IMAX1 7 22 IMAX2 FBT 8 21 CMPIN2 CMPIN1 9 20 VCC COMP1 10 FB1 11 SGND 12 FCB 13 RUN/SS 14 SW2 TG2 BOOST2 PGND 2 23 PGND PGND 3 22 PLLIN IMAX1 4 21 PLLLPF 33 FBT 5 20 IMAX2 CMPIN1 6 19 CMPIN2 COMP1 7 18 VCC 19 COMP2 18 FB2 UH PART MARKING 17 COMP2 FB1 8 9 10 11 12 13 14 15 16 17 PHASEMD SGND PGND 24 PGND 16 VINFF 15 PGOOD 3802 FB2 4 LTC3802EUH 32 31 30 29 28 27 26 25 SW1 1 PHASEMD TG1 SW1 LTC3802EGN BG2 26 TG2 VINFF 3 PVCC BOOST1 PGOOD 27 BOOST2 BG1 2 ORDER PART NUMBER TOP VIEW RUN/SS BG1 ORDER PART NUMBER BOOST1 28 BG2 EXTREF 1 TG1 PVCC FCB TOP VIEW UH PACKAGE 32-LEAD (5mm × 5mm) PLASTIC QFN GN PACKAGE 28-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 34°C/W EXPOSED PAD (PIN 33) IS GND MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 110°C/W Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range. (Note 3) VCC = PVCC = BOOST = 5V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS VCC VCC Supply Voltage PVCC PVCC Supply Voltage (Note 4) ● BVCC BOOST Pin Voltage VBOOST – VSW (Note 4) ● VUVLO Positive Undervoltage Lockout Measured at VCC Measured at VINFF IVCC VCC Supply Current VFB = VCOMP VRUN/SS = 0V, PLLIN Floating ● MIN TYP MAX 3 5 6 V 5 6 V 5 6 V 2.5 2.5 2.8 2.8 V V 6.5 100 9 150 mA µA 2.2 2.2 ● ● UNITS 3802f 2 LTC3802 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range. (Note 3) VCC = PVCC = BOOST = 5V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS IPVCC PVCC Supply Current IBOOST BOOST Pin Current MIN TYP MAX UNITS VFB = VCMPIN = 0V, No Load VRUN/SS = 0V (Notes 5, 6) 2 1 10 mA µA VFB = VCMPIN = 0V, No Load VRUN/SS = 0V (Notes 5, 6) 1 1 10 mA µA 0.600 0.600 0.606 0.609 Switcher Control Loop VFB Feedback Voltage VEXTREF = 5V, 0°C ≤ T ≤ 70°C VEXTREF = 5V ● 0.594 0.591 ∆VFB Feedback Voltage VCC Line Regulation VCC = 4.5V to 6V ∆VOUT Output Voltage Load Regulation (Note 7) AERR Error AMP DC Gain No Load, VEXTREF = VRUN/SS = VCC GBW Error AMP Gain Bandwidth Product ICOMP Error AMP Output Sink/Source Current IFB Voltage Feedback Input Current VFB = 0V to 1V ● ±1 µA ICMPIN Comparators Input Current VCMPIN = 0V to 1V ● ±1 µA IFBT FBT Input Current VFBT = 0V to 1V ● ±1 µA IEXTREF EXTREF Input Current VEXTREF = 0V to 5V ● ±1 µA VEXTREF External Reference Not to Affect VFB ALFF ∆ Drop in Duty Cycle/∆ VINFF RVINFF VINFF Input Resistance VPGOOD Positive Power Good Threshold Negative Power Good Threshold With Respect to 0.6V With Respect to 0.6V VOVP Overvoltage Threshold With Respect to VFB VBURRS (VCMPIN – VFB) to Reset Burst Mode Operation 15 –12 mV mV VSAW SAW Before Line Compensation 1.2 V IIMAX IMAX Source Current ILIM(TH) ISS VSHDN ±0.01 V V 0.1 % 80 dB f = 100kHz (Note 7) 10 MHz VRUN/SS = VCC ±12 mA ● ● 70 %/V ±0.2 1 VVINFF = 5V to 30V ● ● V 2.3 %/V 1 MΩ 5 –5 10 –10 15 –15 % % 3 5 9 % ● –9.0 –8.5 –10 –10 –11.0 –11.5 µA µA ● –15 0 5 1.5 15 mV V/V V/V –5 1.5 –7 2 100 –9 2.5 µA µA/µA µA mA 0.8 1.2 V VIMAX = 1V ILIM Comparator Offset VIMAX/ILIM Threshold Hard ILIM/ILIM Threshold VCMPIN = 0V RUN/SS Source Current RUN/SS Sink/Source Current Ratio RUN/SS Sink Current, ILIM RUN/SS Sink Current, Hard ILIM VCMPIN = VFBT = 0.6V, VPHASEMD = VCC VCMPIN = VFBT = 0.6V, VPHASEMD = 0V VCMPIN = VFBT = 0.6V VCMPIN = VFBT = 0V ● 1 RUN/SS Shutdown Threshold RUN/SS↑ ● 0.4 LOGIC and PGOOD IPHASEMD PHASEMD Pull-Up Current PHASEMD Pull-Down Current VPHASEMD = 0V VPHASEMD = 5V VIH PLLIN, FCB High Level Input Voltage ● VIL PLLIN, FCB Low Level Input Voltage ● IPGOOD VPGOOD Leakage Current VOLPG VPGOOD Output Low Voltage IPGOOD = 1mA ● tPGOOD VPGOOD Falling Edge Delay VPGOOD Rising Edge Delay (Note 8) (Note 8) ● µA µA –7 2 2.4 V Power Good 0.1 100 150 10 0.8 V ±1 µA 0.3 V µs µs 3802f 3 LTC3802 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range. (Note 3) VCC = PVCC = BOOST = 5V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS PLLIN Open or VPLLIN = 0V VPLLLPF = 1.2V VPLLLPF = 0V VPLLLPF = 2.4V 490 550 550 330 750 610 kHz kHz kHz kHz Switcher Switching Characteristics fOSC Oscillator Frequency RPLLIN PLLIN Pull-Down Current Source IPLLLPF Phase Detector Output Current Sourcing Capability Sinking Capability Phase TG1↓ vs PLLIN↓ TG1↓ vs PLLIN↓ TG2↓ vs PLLIN↓ TG2↓ vs PLLIN↓ VPHASEMD Shutdown Threshold Floating 90° Phase Threshold 250 650 400 850 5 µA fPLLIN > fOSC fPLLIN < fOSC –15 15 µA µA PHASEMD Floats VPHASEMD = 5V PHASEMD Floats VPHASEMD = 5V 0 90 180 270 Deg Deg Deg Deg 4.5 V V V 0 % 92 % ● 1.2 ● 3.5 1.7 2.0 4.0 86 89 DCMIN Minimum TG Duty Cycle VPLLIN = 0V ● DCMAX Maximum TG Duty Cycle VPLLIN = 0V, VCMPIN = 0.6V (Note 9) ● tON(MIN) TG Minimum Pulse Width BG Minimum Pulse Width (Notes 7, 10) VCMPIN = 0V (Note 9) tNOV Driver Nonoverlap No Load RDS(ON) TG High RDS(ON) TG Low RDS(ON) BG High RDS(ON) BG Low RDS(ON) IOUT = 100mA (Note 7) Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: The LTC3802 is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: All currents into device pins are positive; all currents out of device pins are negative. All voltages are referenced to ground unless otherwise specified. Note 4: To ensure proper operation, PVCC and BVCC (VBOOST – VSW) must be greater than VGS(ON) of the external MOSFETs. Note 5: Supply current in normal operation is dominated by the current needed to charge and discharge the external MOSFET gates. This current will vary with supply voltage and the external MOSFETs used. 50 400 10 ns ns 30 80 ns 1.6 1.3 1.8 0.7 2.20 1.80 2.50 1.00 Ω Ω Ω Ω Note 6: Supply current in shutdown is dominated by external MOSFET leakage and may be significantly higher than the quiescent current drawn by the LTC3802, especially at elevated temperature. Note 7: Guaranteed by design, not subject to test. Note 8: Rise and fall times are measured using 10% and 90% levels. Delay and nonoverlap times are measured using 50% levels. Note 9: If VCMPIN is less than 90% of its nominal value, BG minimum pulse width is limited to 400ns. Note 10: The LTC3802 leading edge modulation architecture does not have a minimum TG pulse width requirement. The TG minimum pulse width is limited by the rise and fall times. 3802f 4 LTC3802 U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs IOUT Efficiency vs VIN 100 TA = 25°C 98 CIRCUIT ON THE FIRST PAGE OF THIS DATA SHEET 96 90 EFFICIENCY (%) 80 CONTINUOUS MODE 75 70 65 60 50 45 40 0.1 1 10 92 2.5025 VOUT = 2.5V IOUT = 10A 90 88 0 –0.1 –0.2 84 2.4925 –0.3 82 –0.4 80 100 0 5 10 15 VIN (V) 20 30 25 2.4875 0 10 5 15 3802 G02 Line Regulation 3802 G03 VFB vs Temperature VCC = 5V IOUT = 5A TA = 25°C 0.4 604.8 0.3 603.6 0.6 0.2 602.4 0.4 0.1 601.2 0.2 600.0 0 –0.1 VCC = 5V 0.8 ∆VFB (%) VOUT (V) 606.0 0 2.4975 1.0 0.5 ∆VOUT (%) 2.5025 598.8 –0.2 –0.2 597.6 –0.4 2.4925 –0.3 596.4 –0.6 –0.4 595.2 –0.8 2.4875 –0.5 594.0 –50 –25 0 10 5 15 VIN (V) 20 25 30 50 25 0 75 TEMPERATURE (°C) 3802 G04 –1.0 125 Line Feedforward Transient 0.5 TA = 25°C 602.4 100 LTC1323 • TPC05 VFB vs VCC Supply Voltage 603.0 –0.5 20 IOUT (A) 3802 G01 2.5075 0.1 2.4975 IOUT (A) 2.5125 0.3 0.2 VOUT = 2.5V IOUT = 5A 86 VIN = 12V VOUT = 3.3V TA = 25°C CIRCUIT ON THE FIRST PAGE OF THIS DATA SHEET 55 94 0.4 ∆VOUT (%) VOUT 2.5V (NO LOAD) AC 50mV/DIV 0.4 0.3 601.2 0.2 600.6 0.1 600.0 0 599.4 –0.1 598.8 –0.2 598.2 –0.3 597.6 –0.4 ∆VFB (%) 601.8 VFB (mV) EFFICIENCY (%) 85 VIN = 12V TA = 25°C CIRCUIT ON THE FIRST PAGE 2.5075 OF THIS DATA SHEET VOUT (V) Burst Mode OPERATION 0.5 2.5125 VFB (mV) 95 Load Regulation 100 VCOMP AC 50mV/DIV VIN 5V TO 15V STEP 5V/DIV –0.5 597.0 3 3.5 5 4.5 4 5.5 VCC SUPPLY VOLTAGE (V) 6 CIN: 1µF/50V ×6 10µs/DIV SANYO 35CV220AX 3802 G07 3802 G06 3802f 5 LTC3802 U W TYPICAL PERFOR A CE CHARACTERISTICS Load Step in Continuous Mode VOUT 1.2V 50mV/DIV VOUT 1.2V 50mV/DIV VOUT 1.2V 50mV/DIV IOUT 1A TO 10A STEP 5A/DIV VIN = 12V 20µs/DIV CIRCUIT ON FIRST PAGE OF THIS DATA SHEET Burst Mode Waveform with 0.2A Load Load Step in Burst Mode Operation VSW 10V/DIV VSW 10V/DIV IOUT 1A TO 10A STEP 5A/DIV INDUCTOR CURRENT 5A/DIV 3802 G08 3802 G09 VIN = 12V 50µs/DIV CIRCUIT ON FIRST PAGE OF THIS DATA SHEET Burst Mode Waveform with 3A Load ILIM(TH) Offset vs VIMAX VOUT 1.2V 50mV/DIV ILIM(TH) Offset vs Temperature 10 10 TA = 25°C VCC = 5V ILIM(TH) OFFSET (mV) ILIM(TH) OFFSET (mV) INDUCTOR CURRENT 5A/DIV 2 –2 –10 3802 G11 2 –2 –6 –6 VIN = 12V 20µs/DIV CIRCUIT ON FIRST PAGE OF THIS DATA SHEET VCC = 5V VIMAX = 500mV 6 6 VSW 10V/DIV 200 0 600 400 VIMAX (mV) 800 –10 –50 1000 50 25 0 75 TEMPERATURE (°C) –25 Switching Frequency vs Temperature SWITCHING FREQUENCY (kHz) TG 10V/DIV RUN/SS 2V/DIV INDUCTOR CURRENT 20A/DIV 3802 G14 5µs/DIV CIRCUIT ON FIRST PAGE OF THIS DATA SHEET VIN = 12V, VOUT = 3.3V, CSS = 0.01µF, RI(MAX) = 47k, L = 1µH (TOKO-FDA1254-1ROM) 600 VCC = 5V 580 560 540 520 500 –50 –25 50 25 0 75 TEMPERATURE (°C) 125 Switching Frequency vs VCC Supply Voltage SWITCHING FREQUENCY (kHz) 600 100 3802 G13 3802 G12 Short-Circuit Test 3802 G10 VIN = 12V 50µs/DIV CIRCUIT ON FIRST PAGE OF THIS DATA SHEET 100 125 3802 G15 TA = 25°C 580 560 540 520 500 3 3.5 4 4.5 5 5.5 VCC SUPPLY VOLTAGE (V) 6 3802 G16 3802f 6 LTC3802 U W TYPICAL PERFOR A CE CHARACTERISTICS 750 100 VOUT1 5V AC 20mV/DIV TA = 25°C VCC = 5V 650 VCC = 5V 95 fSW = 550kHz MAXIMUM DUTY CYCLE (%) SWITCHING FREQUENCY (kHz) 850 Maximum Duty Cycle vs Temperature Continuous Mode Operation Switching Frequency vs VPLLLPF TG1 20V/DIV 550 VOUT2 1V AC 20mV/DIV 450 350 250 TG2 20V/DIV 0 0.4 0.8 1.2 1.6 2 2.4 90 VCMPIN > 0.54V 85 80 75 VCMPIN < 0.54V 70 65 VIN = 30V 60 – 50 – 25 3802 G18 0.5µs/DIV 75 50 25 TEMPERATURE (°C) 0 VPLLLPF (V) 100 3802 G17 3802 G19 Maximum Duty Cycle vs VCC Supply Voltage Maximum Duty Cycle vs Switching Frequency TA = 25°C 95 VCC = 5V 90 VCMPIN > 0.54V 80 VCMPIN < 0.54V 75 70 90 VCMPIN > 0.54V 85 80 75 VCMPIN < 0.54V 70 3 3.5 4 5 5.5 4.5 VCC SUPPLY VOLTAGE (V) 450 510 570 630 690 SWITCHING FREQUENCY (kHz) 9.5 –5 IRUN/SS SOURCE CURRENT TA = 25°C 10.3 9.7 9.1 –15 0 50 75 25 TEMPERATURE (°C) 100 –20 125 3802 G23 1 1.2 1.4 1.6 1.8 VCOMP (V) 2 2.2 8.5 3.5 4 4.5 5 5.5 VCC SUPPLY VOLTAGE (V) 6 3802 G24 TA = 25°C PVCC = VBOOST – VSW = 5V 60 IPVCC 40 IBOOST1 IBOOST2 20 0 3 2.4 Driver Supply Current vs Load 80 –10 8.5 TA = 25°C VCC = 5V VCMPIN = VFB 3802 G22 DRIVER SUPPLY CURRENT (mA) 0 IIMAX (µA) IIMAX (µA) 5 10.0 8.0 – 50 – 25 30 0.8 10.9 10 IIMAX 9.0 11.5 IRUN/SS (µA) 10.5 40 750 15 IRUN/SS SINK CURRENT 11.0 50 IIMAX vs VCC Supply Voltage 20 VCC = 5V VIN = 20V 60 3802 G21 IIMAX and IRUN/SS vs Temperature 11.5 VIN = 30V 0 390 3802 G20 12.0 70 10 60 330 6 VIN = 12V 20 65 65 VIN = 5V 80 90 DUTY CYCLE (%) MAXIMUM DUTY CYCLE (%) MAXIMUM DUTY CYCLE (%) TA = 25°C 95 fSW = 550kHz 0 Duty Cycle vs VCOMP 100 100 100 85 125 0 2000 6000 8000 4000 CTG, CBG LOAD (pF) 10000 3802 G25 3802f 7 LTC3802 U W TYPICAL PERFOR A CE CHARACTERISTICS Supply Current vs Temperature Supply Current vs Supply Voltage 10 10 1 IVCC IBOOST1 IBOOST2 (NO LOAD) 0.1 IVCC SHUTDOWN VCC = VPVCC = VBOOST – VSW = 5V 0.01 0 25 50 75 –50 –25 IPVCC (NO LOAD) 1 IBOOST1, IBOOST2 (NO LOAD) 0.1 IVCC SHUTDOWN 0.01 100 125 TEMPERATURE (°C) 3802 G26 U U U PI FU CTIO S TA = 25°C IPVCC (NO LOAD) SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) IVCC 3 3.5 4 4.5 5 SUPPLY VOLTAGE (V) 5.5 6 3802 G27 (28-Pin SSOP/32-Pin QFN Package) PVCC (Pin 1/Pin 29): Driver Power Supply Input. PVCC provides power to the two BG drivers and must be connected to an external voltage high enough to fully turn on the external MOSFETs, QB1 and QB2. PVCC requires at least a 10µF bypass capacitor directly to PGND. BG1 (Pin 2/Pin 30): Channel 1 Controller Bottom Gate Drive. The BG1 pin drives the gate of the bottom N-channel synchronous switch MOSFET, QB1. BG1 is designed to drive typically up to 10,000pF of gate capacitance. BOOST1 (Pin 3/Pin 31): Channel 1 Controller Top Gate Driver Supply. BOOST1 should be bootstrapped to SW1 with a 0.1µF capacitor. An external Schottky diode from PVCC to BOOST1 creates a complete floating chargepumped supply at BOOST1. No other external supplies are required. TG1 (Pin 4/Pin 32): Channel 1 Controller Top Gate Drive. The TG1 pin drives the top N-channel MOSFET with a voltage swing equal to PVCC superimposed on the switch node voltage SW1. TG1 is designed to drive typically up to 6000pF of gate capacitance. SW1 (Pin 5/Pin 1): Channel 1 Controller Switching Node. Connect SW1 to the switching node of the channel 1 converter. When the bottom MOSFET QB1 turns on, the current limit comparator and the burst comparator monitor the voltage at SW1. If the voltage drop across MOSFET QB1 is too large, the controller enters current limit; if it is too small, the switcher enters Burst Mode operation. See Current Limit and Burst Mode Applications Information. PGND (Pin 6/Pins 2, 3, 23, 24): Power Ground. The BG drivers return to this pin. Connect PGND to a high current ground node in close proximity to the sources of external MOSFETs QB1 and QB2 and the VIN, PVCC and VOUT bypass capacitors. IMAX1 (Pin 7/Pin 4): Channel 1 Controller Current Limit Set. The IMAX1 pin has an internal 10µA current source pull-up, allowing the current limit and burst comparator threshold to be programmed by a single external resistor to SGND. See Current Limit and Burst Mode Applications Information. 3802f 8 LTC3802 U U U PI FU CTIO S (28-Pin SSOP/32-Pin QFN Package) FBT (Pin 8/Pin 5): Feedback Tracking Input. FBT should be connected through a resistive divider network to VOUT1 to set the channel 1 output slew rate. Upon power-up/-down, the LTC3802 servos FBT and CMPIN2 to the same potential to control the output power-up/-down slew rate. To program both outputs to have the same slew rate, duplicate the CMPIN2 resistive divider at FBT. To have a ratiometric slew rate, short FBT to CMPIN1. To disable the tracking function, short FBT to CMPIN2. CMPIN1 (Pin 9/Pin 6): Channel 1 Controller Comparators Input. CMPIN1 should be connected through a resistive divider network to VOUT1 to monitor its real time output voltage. To improve transient response, a feedforward capacitor can be added to the resistive divider. The power good comparators, overvoltage comparator and Burst reset comparators monitor this node directly. CMPIN1 is a sensitive pin, avoid coupling noise into this pin. COMP1 (Pin 10/Pin 7): Channel 1 Controller Error Amplifier Output. The COMP1 pin is connected directly to the channel 1 error amplifier output and the input of the line feedforward circuit. Use an RC network between the COMP1 pin and the FB1 pin to compensate the feedback loop for optimum transient response. Under start-up conditions, the potential at RUN/SS controls the slew rate at COMP1. FB1 (Pin 11/Pin 8): Channel 1 Controller Error Amplifier Input. FB1 should be connected through a resistive divider network to VOUT1 to set the channel 1 switcher output voltage. Also, connect the channel 1 switcher loop compensation network to FB1. SGND (Pin 12/Pin 9): Signal Ground. All the internal low power circuitry returns to the SGND pin. Connect to a low impedance ground, separated from the PGND node. All feedback, compensation and soft-start connections should return to SGND. SGND and PGND should be connected only at a single point, near the PGND pin and the negative terminal of the VIN bypass capacitor. FCB (Pin 13/Pin 10): Force Continuous Bar. Internally pulled high. When FCB is shorted to GND, the controller forces both converters to maintain continuous synchronous operation regardless of load current. EXTREF (Pin 11, QFN Package Only): External Reference. The EXTREF pin and the internal bandgap voltage are used as the switcher control loop’s reference in a diode OR manner. If the potential at the EXTREF pin is less than 0.6V, it overrides the internal reference and lowers the switcher output voltages. If EXTREF potential is more than 1V, the internal bandgap voltage controls both channel output voltages. EXTREF has no effect on the PGOOD threshold. EXTREF is internally connected to the RUN/SS pin in the GN28 package. RUN/SS (Pin 14/Pin 12): Run Control and Soft-Start Input. An internal 7µA current source pull-up and an external capacitor to ground at this pin sets the start-up delaly (approximately 300ms/µF), the output ramp rate and the time delay for soft current limit. Forcing this pin below 0.8V with an open-drain/collector transistor shuts down the device. Pulling RUN/SS high with a current greater than 10µA can result in malfunctioning of tracking during start-up. Pulling RUN/SS high with currents higher than 50µA can interfere with current limit protection. PGOOD (Pin 15/Pin 13): Open-Drain Power Good Output. PGOOD is pulled to ground under shutdown condition or when any switcher output voltage is not within ±10% of its set point . VINFF (Pin 16/Pin 14): Line Feedforward Compensation Input. Connects to the VIN power supply to provide line feedforward compensation. A change in VIN immediately modulates the input to the PWM comparator and changes the pulse width in an inversely proportional manner, thus bypassing the feedback loop and providing excellent transient line regulation. VINFF is a sensitive pin, an external lowpass filter can be added to this pin to prevent noisy signals from affecting the loop gain. 3802f 9 LTC3802 U U U PI FU CTIO S (28-Pin SSOP/32-Pin QFN Package) PHASEMD (Pin 17/Pin 15): Phase Selector Input. This pin determines the phase relationships between controller␣ 1, controller 2 and the PLLIN signal. When PHASEMD is floating, its value is around 2V, and the internal phaselocked loop synchronizes the falling edge of TG1 to the falling edge of the PLLIN signal. When PHASEMD is forced high, PLLIN leads TG1 by 90°. TG1 and TG2 remain at 180° out of phase independent of the PHASEMD input. When PHASEMD is forced low, an internal current source discharges the RUN/SS slowly to provide power down tracking. Avoid coupling noise into this sensitive pin. FB2 (Pin 18/Pin 16): Channel 2 Controller Error Amplifier Input. See FB1. COMP2 (Pin 19/Pin 17): Channel 2 Controller Error Amplifier Output. See COMP1. VCC (Pin 20/Pin 18): Power Supply Input. All the internal circuits except the switcher output drivers are powered from this pin. VCC should be connected to a low noise 5V supply and should be bypassed to SGND with at least a 10µF capacitor in close proximity to the LTC3802. CMPIN2 (Pin 21/Pin 19): Channel 2 Controller Comparators Input. See CMPIN1. IMAX2 (Pin 22/Pin 20): Channel 2 Controller Current Limit Set. See IMAX1. PLLLPF (Pin 23/Pin 21): Phase-Locked Loop Lowpass Filter. The phase-locked loop’s lowpass filter is tied to this pin. Alternatively, this pin can be driven with an AC or DC voltage source to vary the frequency of the internal oscillator. PLLIN (Pin 24/Pin 22): Phase-Locked Loop Input/External Synchronization Input to the Phase Detector. The falling edge of this signal is used for frequency synchronization. When PLLIN floats or shorts to ground, the controllers free run at 550kHz. SW2 (Pin 25/Pin 25): Channel 2 Controller Switching Node. See SW1. TG2 (Pin 26/Pin 26): Channel 2 Controller Top Gate Drive. See TG1. BOOST2 (Pin 27/Pin 27): Channel 2 Controller Top Gate Driver Supply. See BOOST1. BG2 (Pin 28/Pin 28): Channel 2 Controller Bottom Gate Drive. See BG1. Exposed Pad (Pin 33, QFN Package Only): Exposed Pad is PGND, must be soldered to PCB. 3802f 10 LTC3802 W BLOCK DIAGRA TO CH2 PLLIN CHANNEL 1 SUBCIRCUIT DUPLICATE FOR SECOND CONTROLLER CHANNEL FCB VCC PLL AND OSC BOOST1, 2 PLLLPF PHASEMD TG1, 2 SW1, 2 DISABLE Burst Mode OPERATION – LOGIC PVCC STOP TOP GATE + PWM BG1, 2 PGND SGND –1 LINE FEEDFORWARD COMPENSATION BURST + COMP1, 2 ILIM – – VINFF ÷ 100 + 10µA ÷5 IMAX1,2 7µA SOFTSTART POWER DOWN RUN/SS ERR – FBT – + TRACK EXTREF (QFN PACKAGE ONLY) + CMPIN2 VREF 0.6V MAX – + + + – VREF + 15mV FB1, 2 NEG RESET – POS RESET VREF + 5% VREF – 12mV CMPIN1, 2 – NPG PGOOD + 100µs DELAY VREF – 10% VREF + 10% PPG + FROM CH2 PGOOD COMPARATORS – MPG PGOOD COMPARATORS 3802 BD 3802f 11 LTC3802 U W U U APPLICATIO S I FOR ATIO Switching Architecture The LTC3802 includes two step-down (buck) voltage mode feedback switching regulator controllers. These two controllers act independently of each other except at startup and current limit. For proper power-up sequencing, channel 1 is designated to be the higher output voltage channel (see Start-Up Tracking). Each channel uses two external sychronous N-channel MOSFETs. A floating topside driver and a simple external charge pump provide full gate drive to each upper MOSFET. The controller uses leading edge modulation architecture to allow extremely low duty cycle and fast load recovery operation. In a typical LTC3802 switching cycle, the PWM comparator turns on the top MOSFET and charges up the output capacitor. Some time later, an internal clock resets the top MOSFET, turns on the bottom MOSFET and reduces the output charging current. The top gate duty cycle is controlled by the feedback amplifier, which compares the divided output voltage with an internal reference. This switching cycle repeats itself at a fixed 550kHz frequency or in synchronization with an external oscillator. The internal master clock runs at 550kHz, turning off the top gate once every 1.8µs. Thus, feedback loop components and output inductors and capacitors can be scaled to a particular operating frequency. Noise generated by the circuit will always be in a known frequency band, with the 550kHz frequency designed to leave the 455kHz IF band free of interference. Subharmonic oscillation and slope compensation, common headaches with constant frequency current mode switchers, are absent in voltage mode designs like the LTC3802. Two LTC3802 channels run from a common clock, with the phasing chosen to be 180° from channel 1 to channel 2. This has the effect of doubling the frequency of the switching pulses seen by the input bypass capacitor, significantly lowering its RMS current and reducing the capacitance required. Feedback Control Each LTC3802 channel senses the output voltage at VOUT with an internal feedback op amp (see Block Diagram). This is a real op amp with a low impedance output, 80dB open-loop gain and 10MHz gain-bandwidth product. The positive input is connected to a level-shifted internal 600mV reference, while the negative input is connected to the level-shifted FB pin. The output is connected to COMP, which is in turn connected to the line feedforward circuit and from there to the PWM generator. To speed up the overshoot recovery time, the maximum potential at the COMP pin is internally clamped at a level corresponding to the maximum top gate duty cycle. Under start-up conditions, RUN/SS controls the COMP pin slew rate. At steady state, as shown in Figure 1, the output of the switching regulator is given the following equation R1 VOUT = VREF • 1 + RB Unlike many regulators that use a transconductance (gm) amplifier, the LTC3802 is designed to use an inverting summing amplifier topology with the FB pin configured as a virtual ground. This allows the feedback gain to be tightly controlled by external components, which is not possible with a simple gm amplifier. In addition, the voltage feedback amplifier allows flexibility in choosing pole and zero locations. In particular, it allows the use of “Type 3” compensation, which provides a phase boost at the LC pole frequency and significantly improves the control loop phase margin. In a typical LTC3802 circuit, the feedback loop consists of the line feedforward circuit, the modulator, the external inductor, the output capacitor and the feedback amplifier with its compensation network. All these components affect loop behavior and need to be accounted for in the loop compensation. The modulator consists of the PWM generator, the output MOSFET drivers and the external MOSFETs themselves. The modulator gain varies linearily with the input voltage. The line feedforward circuit compensates for this change in gain, and provides a constant gain from the error amplifier output to the inductor input regardless of input voltage. From a feedback loop point of view, the combination of the line feedforward circuit and the modulator looks like a linear voltage transfer function from COMP to the inductor input and has a gain roughly equal to 22V/V. It has fairly benign AC behavior at typical loop compensation frequencies with significant phase shift appearing at half the switching frequency. 3802f 12 LTC3802 U W U U APPLICATIO S I FOR ATIO Figure 1 shows a Type 3 amplifier. The transfer function of this amplifier is given by the following equation: –(1 + sC1R2)[1 + s(R1 + R3)C3] VCOMP = VOUT sR1(C1 + C2)[1 + s(C1//C2)R2](1 + sC3R3) The RC network across the error amplifier and the feedforward components R3 and C3 introduce two polezero pairs to obtain a phase boost at the system unity gain frequency, fC. In theory, the zeros and poles are placed symmetrically around fC, and the spread between the zeros and the poles is adjusted to give the desired phase boost at fC. However, in practice, if the crossover frequency is much higher than the LC double-pole frequency, this method of frequency compensation normally generates a phase dip within the unity bandwidth and creates some concern regarding conditional stability. If conditional stability is a concern, move the error amplifier’s zero to a lower frequency to avoid excessive phase dip. The following equations can be used to compute the feedback compensation components value: fC = Crossover frequency = fSW 10 1 2πR2C1 1 fC fZ2(RES) = = 5 2π(R1 + R3)C3 fZ1(ERR) = fLC = 1 2πR2(C1 // C2) 1 fP2(RES) = 5fC = 2πR3C3 fP1(ERR) = fESR = Required error amplifier gain at frequency fC: 2 2 f f ≈ 40 log 1 + C – 20 log 1 + C – 20 log( AMOD ) fLC fESR fLC fP2 (RES) fP2(RES) – fZ2(RES) + 1+ 1+ fC fC fZ2(RES) R2 ≈ 20 log • R1 fC fLC fP2(RES) + 1+ 1+ fESR fESR – fLC fC where AMOD is the modulator and line feedforward gain and is equal to: AMOD ≈ VIN(MAX) • DCMAX 30 • 0.89 = ≈ 22V/ V VSAW 1.2 Once the value of resistor R1, poles and zeros location have been decided, the value of R2, C1, C2, R3 and C3 can be obtained from the above equations. C2 VOUT C3 R1 R2 R3 RB VREF C1 FB + –1 GAIN – +1 0 COMP –1 PHASE (DEG) fSW = Switching frequency 1 fLC = 2π LCOUT 1 fESR = 2π RESR COUT choose: GAIN (dB) The external inductor/output capacitor combination makes a more significant contribution to loop behavior. These components cause a second order LC roll-off at the output with 180° phase shift. This roll-off is what filters the PWM waveform, resulting in the desired DC output voltage, but this phase shift causes stability issues in the feedback loop and must be frequency compensated. At higher frequencies, the reactance of the output capacitor will approach its ESR, and the roll-off due to the capacitor will stop, leaving – 20dB/decade and 90° of phase shift. FREQ –90 PHASE –180 BOOST –270 –380 3802 F01 Figure 1. Type 3 Amplifier Compensation 3802f 13 LTC3802 U W U U APPLICATIO S I FOR ATIO Compensating a switching power supply feedback loop is a complex task. The applications shown in this data sheet show typical values, optimized for the power components shown. Though similar power components should suffice, substantially changing even one major power component may degrade performance significantly. Stability also may depend on circuit board layout. To verify the calculated component values, all new circuit designs should be prototyped and tested for stability. Overvoltage Protection and Power Good Flag Notice that the FB pin is the feedback amplifier’s virtual ground node (offset by VREF). Because the typical compensation network does not include local DC feedback around the amplifier, the DC level at FB will be an accurate replica of the output voltage, divided down by the resistive divider. However, the compensation capacitors will tend to attenuate AC signals at FB, especially during quick transients. Because of this delay in the servo loop, the duty cycle is not able to adjust immediately to shifts in the output voltage. This problem is most apparent at high input and low output voltages. Under transient conditions, a slow reaction in the duty cycle could cause a large step in the output voltage. The LTC3802 avoids this voltage instability through the use of an additional comparator input pin, CMPIN, which provides real time measurement of the output voltage. A duplicate FB divider, R1 and RB should be connected to this pin. A small feedforward capacitor can be added across the top resistor to speed up the comparators. The MAX comparator monitors the output voltage through the CMPIN pin. If the output moves 5% above its nominal value, the comparator immediately turns the top MOSFET (QT) off and the bottom MOSFET (QB) on and maintains this state until the output falls back within 5% of its nominal value. This pulls the output down as fast as possible, preventing damage to the (often expensive) load. If CMPIN rises because the output is shorted to a higher supply, QB will stay on until the short goes away, the higher supply current limits or QB dies trying to save the load. This behavior provides maximum protection against overvoltage fault at the output, while allowing the circuit to resume normal operation when the fault is removed. CMPIN is also used as the input for the positive power good comparator PPG and the negative power good comparator NPG. The PPG comparator goes high if the potential at CMPIN is 10% above the nominal value. The NPG comparator fires if CMPIN potential is 10% lower than the nominal value. The output of PPG and NPG is connected to the PGOOD pin through the transistor MPG (see Block Diagram). PGOOD is an open-drain output and requires an external pull-up resistor. If channel 1 and 2 regulator output voltages are within ±10% of their nominal values, the transistor MPG shuts off and PGOOD is pulled high by the external pull-up resistor. If any of the two outputs is outside the 10% window for more than 100µs, PGOOD pulls low indicating that at least one output is out of regulation. For PGOOD to go high, both switcher outputs must be in regulation. PGOOD remains active during soft-start and current limit. Upon power-up, PGOOD is forced low. As soon as the RUN/SS pin rises above the shutdown threshold, the power good comparators take over and control the transistor MPG directly. The 100µs delay ensures that short output transient glitches that are successfully “caught” by the power good comparators don’t cause momentary glitches at the PGOOD pin. Current Limit Protection The LTC3802 includes an onboard current limit circuit that limits the maximum output current to a user-programmed level. It works by sensing the voltage drop across QB when QB is on and comparing that voltage to a user-programmed voltage at IMAX. The IMAX pin includes a trimmed 10µA pull-up, enabling the user to set the voltage at IMAX with a single resistor, RIMAX, to ground. The current comparator reference input is equal to VIMAX divided by 5 (see Block Diagram). Any time QB is on and the current flowing to the output is reasonably large, the SW node at the drain of QB will be somewhat negative with respect to PGND. Since QB looks like a low value resistor during its on-time, the voltage drop across it is proportional to the current flowing in it. The LTC3802 senses this voltage, inverts it and compares it to the current comparator reference. The current comparator begins limiting the output current when the magnitude of the negative voltage is larger than its reference. 3802f 14 LTC3802 U W U U APPLICATIO S I FOR ATIO The current limit detector is connected to an internal 100µA current source. Once current limit occurs, this current begins to discharge the soft-start capacitor at RUN/SS, reducing the duty cycle and controlling the output voltage until the current drops below the limit. The soft-start capacitor needs to move a fair amount before it has any effect on the duty cycle, adding a delay until the current limit takes effect. This allows the LTC3802 to experience brief overload conditions without affecting the output voltage regulation. The delay also acts as a pole in the current limit loop to enhance loop stability. Under severe short-circuit conditions, if the load current is 1.5 times larger than the programmed current limit threshold, the LTC3802 shuts off the top MOSFET immediately. This stops the increase in the inductor current. At this moment, if CMPIN is 10% lower than its nominal value, the LTC3802 hard current limit latches and discharges the RUN/SS capacitor with a current source of more than 1mA until RUN/SS hits its shutdown threshold. Once RUN/SS is completely discharged, the LTC3802 cycles its softstart again. Programming the current limit on the LTC3802 is straightforward. To set the current limit, calculate the expected voltage drop across QB at the maximum desired current: VPROG = (ILIMIT)(RDS(ON)) ILIMIT should be set much higher than the expected operating current, to allow for MOSFET RDS(ON) changes with temperature. Power MOSFET RDS(ON) varies from MOSFET to MOSFET, limiting the accuracy obtainable from the LTC3802 current limit loop. Setting ILIMIT to 150% of the maximum normal operating current is usually safe and will adequately protect the power components if they are chosen properly. Note that ringing on the switch node can cause an error for the current limit threshold. This factor will change depending on the layout. The SW node should have minimum routing from the MOSFETs to the LTC3802 to reduce parasitic inductor and hence ringing. VPROG is then programmed at the IMAX pin using the internal 10µA pull-up current and an external resistor: RIMAX = 5 • VPROG 10µA The resulting value of RIMAX should be checked in an actual circuit to ensure that the current circuit kicks in as expected. Circuits that use very low values for RIMAX (<25k) should be checked carefully, since small changes in RIMAX can cause large ILIMIT changes when the switch node ringing makes up a large percentage of the total VPROG value. If VPROG is set too low, the LTC3802 may fail to start up. The LTC3802 current limit is designed primarily as a disaster preventing, “no blow up” circuit, and is not useful as a precision current regulator. The LTC3802 bottom MOSFET VDS current sensing architecture not only eliminates the external current sense resistors and the corresponding power losses in the high current paths, it allows a wide range of output voltage setting, including extremely low duty cycle operation. On the other hand, for high input voltage with small output inductance applications, care must be taken to avoid inductor saturation during dead-short conditions. As soon as the output short circuits, the controller instantaneously enters maximum duty cycle operation. During the top MOSFET on interval, the current comparator is not monitoring the current and there is no current limit action until the bottom MOSFET turns on and the inductor current exceeds its hard current limit threshold. Typically, the top MOSFET and the inductor need to withstand one clock period of transient high current operation until the hard current limit operation engages. Peak currents can exceed 6 times the maximum DC output current during this period. Most MOSFETs allow 10µs of high current and this short duration of current should not damage the MOSFET. Nevertheless, it is a good idea to reduce the peak inductor current. This can be achieved by having a larger inductance to limit the short-circuit current slew rate, or an inductor with a saturation current that is higher than the hard current limit threshold. Alternatively, an inductor core material with a softer saturation characteristic such as iron powder can be used. Shutdown/Soft-Start The RUN/SS pin performs two functions: when pulled to ground it shuts down the LTC3802, and it acts as a conventional soft-start pin, enforcing a maximum duty cycle limit proportional to the voltage at RUN/SS. An 3802f 15 LTC3802 U W U U APPLICATIO S I FOR ATIO internal 7µA current source pull-up is connected to the RUN/SS pin, allowing a soft-start ramp to be generated with a single external capacitor to ground. The 7µA current source is active even when the LTC3802 is shut down, ensuring the device will start when the external pull-down at RUN/SS is released. Under shutdown conditions, the LTC3802 goes into a micropower sleep mode, and the quiescent current drops to 100µA. The RUN/SS pin shuts down the LTC3802 when it falls below 0.8V (Figure 2). Between about 0.8V and 2V, the LTC3802 wakes up and the duty cycle is kept to a miminum. As the potential at RUN/SS goes higher, the duty cycle increases linearly between 2V and 3.2V, reaching its final value of 89% when RUN/SS exceeds 3.2V. Prior to this point, the feedback amplifier will assume control of the loop and the output will come into regulation. Note that the RUN/SS linear range varies with the potential at VINFF; for 5V input voltage, the RUN/SS active range reduces to 2V-2.25V. The value of the soft-start capacitor, CSS, may depend on the input and output voltages, inductor value, output capacitance and load current. The inductor’s start-up current (from VOUT = 0V), can be much higher than its steady-state current. The difference depends on the input power supply slew rate, the input and output voltages, the LTC3802 soft-start slew rate, and the inductor and output capacitor values. For a given application, the known input and output requirements determine the output inductor and capacitor VOUT 0V NORMAL OPERATION CURRENT LIMIT START-UP 5V RUN/SS COMP CONTROLS DUTY CYCLE 3.2V 2V RUN/SS CONTROLS DUTY CYCLE MINIMUM DUTY CYCLE 0.8V 0V 3802 F02 POWER DOWN MODE LTC3802 ENABLE Figure 2. Soft-Start Operation in Start-Up and Current Limit values. These values establish the transient load recovery time. In general, a low value inductor combined with a high value capacitor yields a short transient load recovery time at the expense of higher inductor ripple and start-up current. These components, together with a small softstart capacitor, can also cause high inrash current. This triggers the LTC3802 current limit comparator and forces the LTC3802 to repeat the soft-start cycle, never allowing the supply to start. Start-up problems can also occur when a small soft-start capacitor is used with a small output inductor and capacitor. High input voltages generate high inrash currents, charging the output capacitor quickly and causing the output to overshoot. The LTC3802 OVP comparator turns off the top MOSFET once the output is 5% higher than its nominal value. However, the residual energy in the inductor will continue to charge the output capacitor, forcing the output voltage to increase further until the inductor energy is depleted. This overshoot at the output causes the feedback loop to operate nonlinearly; the output tends to ring for several cycles until the loop mechanism is restored. Therefore, select CSS with start-up in mind. Choosing CSS to ensure that there is no output overshoot and the inrush current is not able to trigger the current comparator. A minimum recommended soft-start capacitor of CSS = 0.1µF will be sufficient for most applications. Undervoltage Lockout The LTC3802 is designed for wide VIN operation. The internal UVLO circuit monitors the VCC and VINFF potential and starts operation as long as they are above their 2.5V UVLO thresholds. For high VIN supply operation, the low UVLO threshold should not cause any problem under typical application conditions. Upon power-up, once the VIN potential is higher than the UVLO threshold, the LTC3802 releases the RUN/SS node and allows the startup current to charge the soft-start capacitor. The time interval for the RUN/SS potential to ramp from 0.8V to 2V allows the VIN supply to slew to its steady-state potential. A 0.1µF soft-start capacitor creates a 17ms time delay before the driver starts switching. Most power supplies have a start-up time well within this time interval. For some 3802f 16 LTC3802 U W U U APPLICATIO S I FOR ATIO special power supplies with a slow start-up slew rate, the LTC3802 drivers might start switching before the input supply reaches its steady-state value. The high inrush current through the input power cable might cause the VIN supply to dip below the UVLO threshold and cause startup problems. Figure␣ 3 shows a simple circuit to fix this problem. The selection of the zener voltage allows the VIN UVLO trip point to be programmed externally. LTC3802 VIN > VZ 1N4699 VZ = 12V 100k 100k Q1 2N3904 The LTC3802 can be configured to give two different power-up/power-down slew rates to meet different application requirements: ratiometric and coincident tracking configurations (Figure 4). With a ratiometric configuration, the LTC3802 produces two different output slew rates (with VOUT1 > VOUT2). Because each channel’s slew rate is proportional to its corresponding output voltage, the two output voltages reach their steady-state value at about the same time. The coincident configuration produces the same slew rate at both outputs, so that the lower output voltage channel reaches its steady state first. RUN/SS Q2 2N3904 CSS 10k 3802 F03 Figure 3. External UVLO Setting Figure 4 shows the simplified schematic to realize this power-up function. During power-up, the tracking amplifier TRACK servos the tracking feedback loop and forces FBT to be at the same potential as CMPIN2. For ratiometric start-up, set: RT5 = R51 Start-Up Tracking Many DSP chips, microprocessors, FPGAs and ASICs require multiple power supplies for the core and I/O sections. Internally, the core and I/O blocks are isolated by structures which may become forward biased if the supply voltages are not at specified levels. During power-up and power-down operations, differences in the starting point and ramp rates of the two supplies may cause current to flow between the isolation structures which, when prolonged and excessive, can reduce the useable life of the semiconductor device. These currents can also trigger latch-up in devices, leading to device failure. Of greater concern than internal isolation of core and I/O structures are system-level concerns, such as bus contention between the I/O pins of the DSP and external peripheral devices. Power supply sequencing between the core and I/O may be required to prevent bidirectional I/O pins of the DSP and a peripheral device from opposing each other. Since the bus control logic originates in the core section, powering the I/O prior to the core may cause the DSP and peripheral pins to be configured simulatneously as outputs. If the data values on each side are opposing, then the output drivers contend for control, causing excessive current flow and eventually device failure. or remove resistors RT4 and RT5 and short FBT to CMPIN1. At power-up, if the channel 2 output voltage slew rate is too fast, or CMPIN2 is higher than FBT, the tracking amplifier will force a smaller channel 2 duty cycle. Channel␣ 1’s duty cycle is controlled by the RUN/SS pin and is not affected by the tracking amplifier. For coincident start-up, set: RT5 = R52 During power-up, if the channel 1 output voltage is higher than that of channel 2, or if FBT is higher than CMPIN2, the tracking amplifier TRACK starts to discharge the CSS capacitor and forces both channels to have the same duty cycle and output voltage. The tracking amplifier stops discharging once channel 2 reaches its negative power good threshold. To have the proper power-down sequence, ground the PHASEMD pin. This turns on an internal current source which slowly discharges the soft-start capacitor. Once the RUN/SS potential is low enough to control the duty cycle, the tracking amplifier takes control and servos the feedback loop to produce the selected output ramp. The LTC3802 tracking function can be easily disabled by disconnecting the FBT resistive divider and shorting FBT to CMPIN2. 3802f 17 LTC3802 U W U U APPLICATIO S I FOR ATIO L1 VOUT1 LFF AND PWM + RT4 R41 R11 COUT1 EXTREF + REF SAW1 SAW2 7µA RUN/SS CH1 DUTY CYCLE CONTROL 1.7V PHASEMD – RT5 R51 CSS + – CH2 DUTY CYCLE CONTROL VOUT2 L2 LFF AND PWM + COUT2 R12 R42 EXTREF + REF 14µA – POWER-UP/-DOWN OUTPUTS RB2 RB1 R52 CMPIN1 FBT VOUT1 MUST BE HIGHER THAN VOUT2 R11 = R41 = RT4 = R12 = R42 RB1 = R51, RB2 = R52 TRACK – + CMPIN2 3802 F04 RATIOMETRIC TRACKING RT5 = R51 CSS = 1µF VOUT1 WITH 10Ω LOAD 0.5V/DIV VOUT2 WITH 10Ω LOAD 10ms/DIV COINCIDENT TRACKING RT5 = R52 CSS = 1µF VOUT1 WITH 10Ω LOAD 0.5V/DIV VOUT2 WITH 10Ω LOAD 10ms/DIV Figure 4. Simplified Power-Up/Power-Down Output Tracking Schematic The QFN version of the LTC3802 provides an additional reference pin for external ratiometric start-up. If the potential at the EXTREF pin is less than 0.6V, it overrides the internal reference. This pin can be connected to an external ramp to control the output slew rate. If external tracking is not required, connect EXTREF to a potential somewhat larger than 0.6V or short EXTREF to the RUN/SS pin. The EXTREF pin should never be allowed to float. In the GN28 package, EXTREF is internally shorted to the RUN/SS pin. Burst Mode Operation The LTC3802 switcher supply has two modes of operation. Under heavy loads, it operates as a fully synchronous, continuous conduction switching regulator. In this mode of operation (continuous mode), the current in the inductor flows in the positive direction (towards the output) during the entire switching cycle, constantly supplying current to the load. In this mode, the synchronous switch (QB) is on whenever QT is off, so the current always flows through a low impedance switch, minimizing voltage drop and power loss. This is the most efficient mode of operation at heavy loads, where the resistive losses in the power devices are the dominant loss term. Continuous mode works efficiently when the load current is greater than half of the ripple current in the inductor. In a buck converter like the LTC3802, the average current in the inductor (averaged over one switching cycle) is equal to the load current. The ripple current is the difference between the maximum and the minimum current during a switching cycle (see Figure 5a). The ripple current depends on inductor value, clock frequency and output voltage, but is constant regardless of load as long as the LTC3802 remains in continuous mode. See the Inductor Selection section for a detailed description of ripple current. 3802f 18 LTC3802 U W U U APPLICATIO S I FOR ATIO INDUCTOR CURRENT As the output load current decreases in continuous mode, the average current in the inductor will reach a point where it drops below half the ripple current. At this point, the current in the inductor will reverse during a portion of the switching cycle, or begin to flow from the output back to the input. This does not adversely affect regulation, but does cause additional losses as a portion of the inductor current flows back and forth through the resistive power switches, giving away a little more power each time and lowering the efficiency. There are some benefits to allowing this reverse current flow: the circuit will maintain regulation even if the load current drops to zero and the output ripple voltage and frequency remain constant at all loads, easing filtering requirements. However, continuous mode at low output current does cause losses in efficiency. A portion of the inductor current flows back and forth through the resistive power switches, causing I2R losses. The drivers continue to switch QT and QB on and off once a cycle. Each time an external MOSFET is turned on, the internal driver must charge its gate to a potential above the MOSFET’s source voltage; when the MOSFET is turned off, that charge is lost to ground or SW. At the high switching frequencies, the lost gate charges can add up to tens of millicoulombs. As the load current continues to drop, these charges quickly become the dominant power loss term, reducing efficiency once again. IAVERAGE IRIPPLE TIME 3802 F05a INDUCTOR CURRENT Figure 5a. Continuous Mode IRIPPLE IAVERAGE TIME Figure 5b. Burst Mode Operation 3802 F05b To minimize the switching loss and reverse current flow at light loads, the LTC3802 switches to a second mode of operation: Burst Mode operation (Figure 5b). In Burst Mode operation, at the end of the QB cycle, if the inductor current approaches zero or goes negative, the LTC3802 turns off both drivers. The actual cutoff threshold is proportional to the IMAX setting and is equal to: – VIMAX – 3mV 100 The –3mV built-in offset overcomes the random mismatch in the burst compararator trip point and allows Burst Mode operation at no load. Once both MOSFETs shut off, the voltage at the SW pin will float around VOUT, and the inductor current and the voltage across the inductor will be close to zero. This prevents current from flowing backwards in QB, eliminating that power loss term. The moment the LTC3802 enters Burst Mode operation, both drivers skip a number of switching cycles until the internal 36µs timeout forces the switcher to return to continuous operation. This timeout eliminates the audible noise from certain types of inductors when they are lightly loaded. After the 36µs timeout, the LTC3802 forces one continuous mode cycle and checks the inductor current at the end of the period. If it is still too small, it enters Burst Mode operation again. This pattern repeats until the output is loaded. The LTC3802 returns to continuous mode operation if it detects that CMPIN potential is 12mV below or 15mV above its nominal bandgap voltage. Immediately after returning to continuous mode operation, the regulator output might continue to droop slightly until the feedback loop responds and requests an increase in duty cycle. During sudden transient steps, the regulator output ripple is limited by the feedback loop transient response and is independent of the mode of operation. The small 15mV and –12mV offset at the POS and NEG RESET comparators ensure that after a transient load step, the LTC3802 returns to continuous mode quickly. This minimizes the output ripple under Burst Mode operation. For proper Burst Mode operation, the LTC3802 requires very precise CMPIN and FB sensing. To realize this, CMPIN and FB must use the same resistive divider values 3802f 19 LTC3802 U W U U APPLICATIO S I FOR ATIO and all resistors should have better than 1% tolerance. If this is not possible and Burst Mode operation is required, the potential at CMPIN can be set slightly higher than FB by using a slightly bigger resistor from CMPIN to ground. This removes the requirement of having expensive resistors at the FB and CMPIN pins, at the expense of having a higher Burst Mode ripple and slightly different overvoltage and power good thresholds. To ensure clean Burst Mode operation, the CMPIN and FB resistive divider requires good layout technique. Both resistive dividers must be connected to the same nodes and away from high current paths. Low load current efficiency depends strongly on proper Burst Mode operation. In an ideal system, the gate drive is the dominant loss term at low load currents. Burst Mode operation turns off all output switching for several clock cycles in a row, significantly cutting gate drive losses. As the load current in Burst Mode operation falls toward zero, the current drawn by the LTC3802 falls to a quiescent level—about 6.5mA. To maximize low load efficiency, make sure the LTC3802 is allowed to enter Burst Mode operation as cleanly as possible. Operating Frequency/Frequency Synchronization The LTC3802 controller uses a constant frequency, phaselockable internal oscillator with its frequency determined by an internal capacitor. This capacitor is charged by a fixed current plus an additional current that is proportional to the voltage applied to the PLLLPF pin. When the PLLIN pin is not used, an internal pull-down current source forces PLLIN to ground and the controller runs at a fixed 550kHz switching frequency. The phase-locked loop allows the internal oscillator to be synchronized to an external source via the PLLIN pin. The phase-locked loop consists of an internal voltage controlled oscillator, a divide by 12 frequency divider and a phase detector. The voltage controlled oscillator monitors the output of the phase detector at the PLLLPF pin. It provides a linear relationship between the PLLLPF potential and the master oscillator frequency. A DC voltage input from 0.5V to 1.9V corresponds to a 330kHz to 750kHz master switching frequency. The phase detector used is an edge sensitive digital circuit which provides zero degree phase shift between the external and internal oscillators. This type of phase detector will not lock up on an input frequency close to the harmonics of the VCO center frequency. The output of the phase detector is a complementary pair of current sources charging or discharging the external filter network on the PLLLPF pin. A simplified block diagram is shown in Figure␣ 6. If the external frequency, fPLLIN, is greater than the oscillator frequency, fOSC, current is sourced continuously, pulling up the PLLLPF pin. When fPLLIN is less than fOSC, current is sunk continuously, pulling down the PLLLPF pin. If fPLLIN and fOSC are the same but exhibit a phase difference, the current sources turn on for a period corresponding to the phase difference. Thus the voltage on the PLLLPF pin is adjusted until the phase and frequency of the external and internal oscillators are identical. At this stable operating point the phase comparator output is open and the filter capacitor, CLP, holds the voltage. When locked, the PLL aligns the turn off of the top MOSFET to the falling edge of the synchronizing signal. The loop filter components, CLP and RLP, smooth out the current pulses from the phase detector and provide a stable input to the voltage controlled oscillator. The filter components, CLP and RLP, determine how fast the loop acquires lock. Typically RLP = 10k and CLP is between 0.01µF and 0.1µF. The PHASMD pin determines the relative phases between the TG1, TG2 and the PLLIN signals. When PHASEMD is VCC LTC3802 PHASEMD PLLIN PHASE DETECTOR PLLLPF RLP CLP ÷12 VCO INTERNAL MASTER CLOCK 3806 F06 Figure 6. Phase-Locked Loop Block Diagram 3802f 20 LTC3802 U W U U APPLICATIO S I FOR ATIO floating, it sits at around 2V and the internal phase-locked loop synchronizes TG1’s falling edge to the falling edge of the PLLIN signal. When PHASEMD is high, these two signals are 90° out of phase. TG1 and TG2 remains 180° out of phase independent of PHASEMD input. The PHASEMD signal together with the PLL circuit can be used to synchronize an additional LTC3802 power supply circuit to provide a 4-phase, 4-output solution. Compared to an in-phase multiple controller solution, the LTC3802’s 4-phase design reduces the input capacitor ripple current requirements and efficiency losses because the peak current drawn from the input capacitor is spaced out within the switching cycle. EXTERNAL COMPONENTS SELECTION VCC and PVCC Power Supplies Power for the top and bottom MOSFET drivers is derived from the PVCC pin; the internal controller circuitry is derived from the VCC pin. Under typical operating conditions, the total current consumption at these two pins should be well below 100mA. Hence, PVCC and VCC can be connected to an external auxiliary 5V power supply. If an auxiliary supply is not available, a simple zener diode and a darlington NPN buffer can be used to power up these two pins as shown in Figure 7. To prevent switching noise from coupling to the sensitive analog control circuitry, VCC should VIN RZ 2k + CIN 100Ω Q1 DCP VINFF BOOST L VOUT QT CCP TG SW + COUT D1 QB BG LTC3802 + VZ 5.6V 10µF PVCC 0.1µF PGND 10Ω + 10µF VCC 0.1µF SGND 3802 F07 Q1: ZETEX FZT603 VZ: MM5Z6V2ST1 Figure 7. LTC3802 Power Supply Inputs have a 10µF bypassed capacitor close to the device. The BiCMOS process that allows the LTC3802 to include large on-chip MOSFET drivers also limits the maximum PVCC and VCC voltage to 7V. This limits the practical maximum auxiliary supply to a loosely regulated 7V rail. If VCC drops below 2.5V or PVCC drops below VCC by more than 1V, the LTC3802 goes into undervoltage lockout and prevents the power switches from turning on. Top MOSFET Driver Supply An external bootstrap capacitor, CCP, connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DCP from PVCC when the switch node is low. When the top MOSFET turns on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + PVCC. The boost capacitor needs to store about 100 times the gate charge required by the top MOSFET. In most applications a 0.1µF to 1µF, X5R or X7R dielectric capacitor is adequate. Power MOSFET Selection The LTC3802 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the threshold voltage V(GS)TH, breakdown voltage V(BR)DSS, maximum current IDS(MAX), on-resistance RDS(ON) and input capacitance. The gate drive voltage is set by the 5V PVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC3802 applications. If the PVCC voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered. Pay close attention to the V(BR)DSS specification, because most logic-level MOSFETs are limited to 30V or less. The MOSFETs selected should have a V(BR)DSS rating greater than the maximum input voltage and some margin should be added for transients and spikes. The MOSFETs selected should also have an IDS(MAX) rating of at least two times the maximum power stage output current. Still, this may not be a sufficient margin so it is advisable to calculate the MOSFET’s junction temperature to ensure that it is not exceeded. The LTC3802 uses the bottom MOSFET as the current sense element, particular attention must be paid to its 3802f 21 LTC3802 U W U U APPLICATIO S I FOR ATIO on-resistance. MOSFET on-resistance is typically specified with a maximum value RDS(ON)(MAX) at 25°C. In this case, additional margin is required to accommodate the rise in MOSFET on-resistance due to self heating and higher ambient temperature: RDS(ON)(MAX) (T) = ρT • RDS(ON)(MAX) (25°C) The ρT term is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with temperature, typically about 0.4%/°C as shown in Figure 8a. For a maximum junction temperature of 100°C, using a value ρT = 1.3 is reasonable. MOSFET input capacitance is a combination of several components but can be taken from the typical “gate charge” curve included on most data sheets (Figure 8b). The curve is generated by forcing a constant input current into the gate of a common source, current source loaded stage and then plotting the gate voltage versus time. The initial slope is the effect of the gate-to-source and the gateto-drain capacitance. The flat portion of the curve is the result of the Miller multiplication effect of the drain-to-gate When the controller is operating in continuous mode the duty cycles for the top and bottom MOSFETs are given by: Top Gate Duty Cycle = VOUT VIN V –V Bottom Gate Duty Cycle = IN OUT VIN 2.0 ρT NORMALIZED ON-RESISTANCE capacitance as the drain drops the voltage across the current source load. The upper sloping line is due to the drain-to-gate accumulation capacitance and the gate-tosource capacitance. The Miller charge (the increase in coulombs on the horizontal axis from a to b while the curve is flat) is specified for a given VDS drain voltage, but can be adjusted for different VDS voltages by multiplying by the ratio of the application VDS to the curve specified VDS values. A way to estimate the CMILLER term is to take the change in gate charge from points a and b on a manufacturers data sheet and divide by the stated VDS voltage specified. CMILLER is the most important selection criteria for determining the transition loss term in the top MOSFET but is not directly specified on MOSFET data sheets. CRSS and COS are specified sometimes but definitions of these parameters are not included. 1.5 The power dissipation for the top and bottom MOSFETs at maximum output current are given by: 1.0 PTOP = 0.5 0 –50 50 100 0 JUNCTION TEMPERATURE (°C) 150 Figure 8a. Typical MOSFET RDS(ON) vs Temperature MILLER EFFECT b QIN CMILLER = (QB – QA)/VDS )( ) PBOT = ( )( )( VIN – VOUT IOUT (MAX)2 ρT (TOP) RDS(ON)(MAX) VIN ) V VGS a )( 1 1 + PV – V • fsw CC TH(IL) VTH(IL) 3802 F08a VIN ( VOUT IOUT (MAX)2 ρT (TOP) RDS(ON)(MAX) VIN IOUT (MAX) +VIN2 (RDR )(C MILLER ) • 2 + VGS +V DS – – 3802 F08b Figure 8b. Gate Charge Characteristics where: RDR = Effective top driver resistance VTH(IL) = MOSFET data sheet specified typical gate threshold voltage at the specified drain current 3802f 22 LTC3802 U W U U APPLICATIO S I FOR ATIO CMILLER = Calulated Miller capacitance using the gate charge curve from the MOSFET data sheet fSW = Switching frequency Both MOSFETs have conduction losses (I2R) while the topside N-channel equation includes an additional term for transition losses, which peak at the highest input voltage. For VIN < 12V, the high current efficiency generally improves with larger MOSFETs, while for VIN > 12V, the transition losses rapidly increase to the point that the use of a higher RDS(ON) device with lower CMILLER actually provides higher efficiency. The bottom MOSFET losses are greatest at high input voltage when the top switch duty factor is low or during a short circuit when the bottom switch is on close to 100% of the period. Schottky Diode D1/D2 Selection The Schottky diode D1 shown in Figure 7 conducts during the dead time between the conduction of the power MOSFET switches. It is intended to prevent the body diode of the bottom MOSFET from turning on and storing a charge during the dead time, which can cause a modest (about 1%) efficiency loss. The diode can be rated for about one half to one fifth of the full load current since it is on for only a fraction of the duty cycle. In order for the diode to be effective, the inductance between it and the bottom MOSFET must be as small as possible, mandating that these components be placed adjacently. CIN Selection The input bypass capacitor in an LTC3802 circuit is common to both channels. The input bypass capacitor gets exercised in three ways: its ESR must be low enough to keep the supply drop low as the top MOSFETs turn on, its RMS current capability must be adequate to withstand the ripple current at the input, and the capacitance must be large enough to maintain the input voltage until the input supply can make up the difference. Generally, a capacitor (particularly a non-ceramic type) that meets the first two parameters will have far more capacitance than is required to keep capacitance-based droop under control. The input capacitor’s voltage rating should be at least 1.4 times the maximum input voltage. Power loss due to ESR occurs not only as I2R dissipation in the capacitor itself, but also in overall battery efficiency. For mobile applications, the input capacitors should store adequate charge to keep the peak battery current within the manufacturer’s specifications. The input capacitor RMS current requirement is simplified by the multiphase architecture and its impact on the worst-case RMS current drawn through the input network (battery/fuse/capacitor). It can be shown that the worstcase RMS current occurs when only one controller is operating. The controller with the highest (VOUT)(IOUT) product needs to be used to determine the maximum RMS current requirement. Increasing the output current drawn from the other out-of-phase controller will actually decrease the input RMS ripple current from this maximum value. The out-of-phase technique typically reduces the input capacitor’s RMS ripple current by a factor of 30% to 70% when compared to a single phase power supply solution. In continuous mode, the source current of the top N-channel MOSFET is approximately a square wave of duty cycle VOUT/VIN. The maximum RMS capacitor current is given by: IRMS ≈ IOUT(MAX) VOUT ( VIN – VOUT ) VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. The total RMS current is lower when both controllers are operating due to the interleaving of current pulses through the input capacitors. This is why the input capacitance requirement calculated above for the worst-case controller is adequate for the dual controller design. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. Always consult the manufacturer if there is any question. 3802f 23 LTC3802 U W U U APPLICATIO S I FOR ATIO Medium voltage (20V to 35V) ceramic, tantalum, OS-CON and switcher-rated electrolytic capacitors can be used as input capacitors, but each has drawbacks: ceramics have high voltage coefficients of capacitance and may have audible piezoelectric effects; tantalums need to be surgerated; OS-CONs suffer from higher inductance, larger case size and limited surface mount applicability; and electrolytics’ higher ESR and dryout possibility require several to be used. Sanyo OS-CON SVP, SVPD series; Sanyo POSCAP TQC series or aluminum electrolytic capacitors from Panasonic WA series or Cornel Dublilier SPV series, in parallel with a couple of high performance ceramic capacitors, can be used as an effective means of achieving low ESR and its big bulk capacitance goal for the input bypass. COUT Selection The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The output ripple ∆VOUT is approximately bounded by: 1 ∆VOUT ≤ ∆IL ESR + 8 • fSW • COUT where ∆IL is the inductor ripple current. ∆IL may be calculated using the equation: ∆IL = VOUT VOUT 1– L • fSW VIN Since ∆IL increases with input voltage, the output ripple voltage is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Manufacturers such as Sanyo, Panasonic and Cornell Dublilier should be considered for high performance through-hole capacitors. The OS-CON semiconductor electrolyte capacitor available from Sanyo has a good (ESR)(size) product. An additional ceramic capacitor in parallel with OS-CON capacitors is recommended to offset the effect of lead inductance. In surface mount applications, multiple capacitors may have to be paralleled to meet the ESR or transient current handling requirements of the application. Aluminum electrolytic and dry tantalum capacitors are both available in surface mount configurations. New special polymer surface mount capacitors offer very low ESR also but have much lower capacitive density per unit volume. In the case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. Several excellent output capacitor choices are the Sanyo POSCAP TPD, POSCAP TPB, AVX TPS, AVX TPSV, the Kemet T510 series of surface mount tantalums,Kemet AO-CAPs or the Panasonic SP series of surface mount special polymer capacitors available in case heights ranging from 2mm to 4mm. Other capacitor types include Nichicon PL series and Sprague 595D series. Consult the manufacturer for other specific recommendations. Inductor Selection The inductor in a typical LTC3802 circuit is chosen primarily for inductance value and saturation current. The inductor should not saturate below the hard current limit threshold. The inductor value sets the ripple current, which is commonly chosen at around 40% of the anticipated full load current. Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Highest efficiency is obtained at low frequency with small ripple current. However, achieving high efficiency requires a large inductor and generates higher output voltage excursion during load transients. There is a tradeoff between component size, efficiency and operating frequency. Given a specified limit for ripple current, the inductor value can be obtained using the following equation: L= VOUT V • 1 – OUT fSW • ∆IL(MAX) VIN(MAX) Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy 3802f 24 LTC3802 U W U U APPLICATIO S I FOR ATIO or Kool Mµ® cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko. PC Board Layout Checklist When laying out the printed circuit board, start with the power device. Be sure to orient the power circuitry so that a clean power flow path is achieved. Conductor widths should be maximized and lengths minimized. After you are satisfied with the power path, the control circuitry should be laid out. It is much easier to find routes for the relatively small traces in the control circuits than it is to find circuitous routes for high current paths. After the layout, the following checklist should be used to ensure proper operation of the LTC3802. 1. Place the top N-channel MOSFETs QT1 and QT2 within 1cm of each other with a common drain connection at CIN. Do not attempt to split the input decoupling for the two channels because doing so can create a resonant loop. 2. Place CIN, COUT, the MOSFETs, Schottky diode and the inductor together in one compact area. 3. Split the signal and power grounds. The path formed by the top and bottom N-channel MOSFETs, Schottky diode, and the CIN capacitor should have short leads and PC trace lengths. The output capacitor (–) terminals should be connected as close as possible to the (– ) terminals of the input capacitor by placing the capacitors next to each other. The combined LTC3802 signal ground pin and the ground return of CVCC must return to the combined COUT (–) terminals. Use a modified “star ground” technique: a low impedance, large copper area central grounding point on the same side of the PC board as the input and output capacitors, with tie-ins for the bottom of the VCC decoupling capacitor, the bottom of the voltage feedback resistive divider and the SGND pin of the IC. 4. Keep the switching nodes (SW1, SW2), top gate nodes (TG1, TG2), and boost nodes (BOOST1, BOOST2) away from sensitive small-signal nodes, especially from the opposite channel’s voltage and current sensing feedback pins. All of these nodes have very large and fast moving signals and therefore should be kept on the “output side” of the LTC3802 and occupy minimum PC trace area. 5. Reduce the parasitic inductance at the SW and PGND connections to allow proper Burst Mode operation. Use multiple vias if possible. 6. Use the same resistor values for the FB and CMPIN resistive divider. Connect these dividers to the same node: the (+) terminals of COUT and signal ground. The dividers should be connected to a node away from any high current path. 7. Place the VCC and PVCC decoupling capacitor close to the IC, between the VCC and the signal ground, and between PVCC and PGND. The VCC capacitor provides a quiet supply for the sensitive analog circuits and the PVCC capacitor carries the MOSFET drivers current peaks. An additional 1µF ceramic capacitor placed immediately next to the VCC and SGND pins can substantially improve noise performance. Checking Transient Response For all new LTC3802 PCB circuits, transient tests need to be performed to verify the proper feedback loop operation. The regulator loop response can be checked by looking at the load current transient response. Switching regulators take several cycles to respond to a step in DC (resistive) load current. When a load step occurs, VOUT shifts by an amount equal to ∆ILOAD • (ESR), where ESR is the effective series resistance of COUT. ∆ILOAD also begins to charge or discharge COUT generating the feedback error signal that forces the regulator to adapt to the current change and return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for excessive overshoot or ringing which would indicate a stability problem. Measuring transient response presents a challenge in two respects: obtaining an accurate measurement and generating a suitable transient for testing the circuit. Output measurements should be taken with a scope probe directly across the output capacitor. Proper high frequency probing techniques should be used. Do not use the 6" ground lead that comes with the probe! Use an adapter that fits on the tip of the probe and has a short ground clip Kool Mµ is a registered trademark of Magnetics, Inc. 3802f 25 LTC3802 U W U U APPLICATIO S I FOR ATIO to ensure that inductance in the ground path doesn’t cause a bigger spike than the transient signal being measured. The typical probe tip ground clip is spaced just right to span the leads of a typical output capacitor. In general, it is best to take this measurement with the 20MHz bandwidth limit on the oscilloscope turned on to limit high frequency noise. Note that microprocessor manufacturers typically specify ripple ≤ 20MHz, as energy above 20MHz is generally radiated and not conducted and will not affect the load even if it appears at the output capacitor. Now that we know how to measure the signal, we need to have something to measure. The ideal situation is to use the actual load for the test, switching it on and off while watching the output. If this isn’t convenient, a current step generator is needed. This generator needs to be able to turn on and off in nanoseconds to simulate a typical switching logic load, so stray inductance and long clip leads between the LTC3802 and the transient generator must be minimized. element—many power resistors use an inductive spiral pattern and are not suitable for use here. A simple solution is to take ten 1/4W film resistors and wire them in parallel to get the desired value. This gives a noninductive resistive load which can dissipate 2.5W continuously or 250W if pulsed with a 1% duty cycle, enough for most LTC3802 circuits. Solder the MOSFET and the resistor(s) as close to the output of the LTC3802 circuit as possible and set up the signal generator to pulse at a 100Hz rate with a 1% duty cycle. This pulses the LTC3802 with 100µs transients 10ms apart, adequate for viewing the entire transient recovery time for both positive and negative transitions while keeping the load resistor cool. VOUT LTC3802 RLOAD 50Ω PULSE GENERATOR IRFZ44 OR EQUIVALENT 10k 0V TO 10V 100Hz, 1% DUTY CYCLE 3802 F09 LOCATE CLOSE TO THE OUTPUT Figure 9 shows an example of a simple transient generator. Be sure to use a noninductive resistor as the load Figure 9. Transient Load Generator U PACKAGE DESCRIPTIO GN Package 28-Lead Plastic SSOP (Narrow .150 Inch) (Reference LTC DWG # 05-08-1641) .386 – .393* (9.804 – 9.982) .045 ±.005 28 27 26 25 24 23 22 21 20 19 18 17 1615 .254 MIN .033 (0.838) REF .150 – .165 .229 – .244 (5.817 – 6.198) .0165 ± .0015 .150 – .157** (3.810 – 3.988) .0250 TYP 1 RECOMMENDED SOLDER PAD LAYOUT .015 ± .004 × 45° (0.38 ± 0.10) .0075 – .0098 (0.191 – 0.249) 2 3 4 5 6 7 8 .053 – .069 (1.351 – 1.748) 9 10 11 12 13 14 .004 – .009 (0.102 – 0.249) 0° – 8° TYP .016 – .050 (0.406 – 1.270) .008 – .012 (0.203 – 0.305) .0250 (0.635) BSC NOTE: *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH 1. CONTROLLING DIMENSION: INCHES SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE INCHES 2. DIMENSIONS ARE IN (MILLIMETERS) **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 3. DRAWING NOT TO SCALE GN28 (SSOP) 0502 3802f 26 LTC3802 U PACKAGE DESCRIPTIO UH Package 32-Lead Plastic QFN (5mm × 5mm) (Reference LTC DWG # 05-08-1693) 0.70 ±0.05 5.50 ±0.05 4.10 ±0.05 3.45 ±0.05 (4 SIDES) PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT 5.00 ± 0.10 (4 SIDES) BOTTOM VIEW—EXPOSED PAD 0.23 TYP (4 SIDES) R = 0.115 TYP 0.75 ± 0.05 0.00 – 0.05 31 32 0.40 ± 0.10 PIN 1 TOP MARK (NOTE 6) 1 2 3.45 ± 0.10 (4-SIDES) (UH) QFN 0603 0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE M0-220 VARIATION WHHD-(X) (TO BE APPROVED) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.25 ± 0.05 0.50 BSC 3802f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 27 LTC3802 U TYPICAL APPLICATIO 2.5V/15A and 1.8V/15A Outputs with Start-Up Tracking and External Synchronization 1.5µF 35V ×8 2k 100Ω 5.6V MM5Z6V2ST1 Q1 ZETEX FZT603 VCC + 10µF QT1 Si7884DP L1 1µH VOUT1 2.5V 15A 560pF R11 10k RT4 10k R41 10k + 0.1µF D1 B340B COUT1 330µF 4V ×3 QB1 Si7884DP ×2 RIMAX1 62k RATIOMETRIC RB1 3.16k RT5 4.99k TRACKING GND C31 1500pF R51 3.16k R31 390Ω R21 15k C11 2200pF C21 330pF BURST CONTINUOUS 1 PVCC 4 TG1 3 BOOST1 5 SW1 2 BG1 6 PGND D3 BAS40-06LT1 26 TG2 27 BOOST2 25 SW2 28 BG2 24 PLLIN 23 PLLLPF QT2 Si7884DP QB2 Si7884DP ×2 10k 0.1µF 10k 2k CSS 0.1µF COUT1, COUT2: SANYO 4TPD330M L1, L2: TOKO FDA1254-1R0M 22 IMAX2 21 CMPIN2 20 VCC 19 COMP2 18 FB2 17 PHASEMD 16 VINFF 15 PGOOD 10k VCC VIN CINFF 0.47µF EXT SYNC RIMAX2 62k LTC3802 7 IMAX1 8 FBT 9 CMPIN1 10 COMP1 11 FB1 12 SGND 13 FCB 14 RUN/SS L2 1µH VOUT2 1.8V C32 15A 1500pF 0.1µF COUT2 330µF 4V ×3 + R42 10k R12 10k R32 390Ω D2 B340B GND 10Ω VCC + 0.1µF 10µF 0 DEG 90 DEG CIN 22µF 35V ×4 + POWER DOWN VIN 8V TO 28V R52 4.99k 560pF RB2 4.99k C22 R22 330pF C12 15k 2200pF 3802 TA02 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1530 High Power Synchronous Step-Down Controller SO-8 with Current Limit. No RSENSETM Required LTC1628/LTC1628-PG 2-Phase, Dual Output Synchronous Step-Down DC/DC LTC1628-SYNC Controllers Constant Frequency, Standby 5V and 3.3V LDOs, Power Good LTC1702A Dual PolyPhase® Synchronous Step-Down Switching Regulator 550kHz Operation, No RSENSE, 3V ≤ VIN ≤ 7V, Voltage Mode LTC1704 550kHz Synchronous Switching Regulator Controller Plus Linear Regulator Controller 550kHz, 25MHz GBW, Voltage Mode Switching Regulator Plus 2A Linear Regulator Controller LTC1735 Synchronous Step-Down DC/DC Controller 3.5V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ 6V, Current Mode LTC1778 No RSENSE Current Mode Synchronous Step-Down Controller Up to 97% Efficiency, 4V ≤ VIN ≤ 36V, 0.8V ≤ VOUT ≤ (0.9)(VIN), IOUT Up to 20A LTC3703 High Input Synchronous Step-Down Controller VIN ≤ 100V, 100kHz to 600kHz Operation LTC3708 Dual, 2-Phase, No RSENSE Synchronous Step-Down DC/DC Controller 2-Phase, No RSENSE Controller with Output Tracking LTC3728 550kHz, 2-Phase Dual Output Synchronous Step-Down Controller Synchronizable, Current Mode, 3.5V ≤ VIN ≤ 36V, SSOP and QFN Packages LTC3832 High Power Step-Down Synchronous DC/DC Controller for Low Voltage Operation Constant Frequency, Voltage Mode with Current Limit 3V ≤ VIN ≤ 8V, 0.6V ≤ VOUT ≤ (0.9)(VIN) PolyPhase is a registered trademark of Linear Technology Corporation. No RSENSE is a trademark of Linear Technology Corporation. 3802f 28 Linear Technology Corporation LT/TP 0404 1K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2004