LTC4089-1 USB Power Manager with High Voltage Switching Charger DESCRIPTIO FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ U ■ Seamless Transition Between Power Sources: Li-Ion Battery, USB, and 6V to 36V External Supply High Efficiency 1.2A Charger from 6V to 36V Input Load Dependent Charging from USB Input Guarantees Current Compliance 215m Internal Ideal Diode plus Optional External Ideal Diode Controller Provides Low Loss Power Path When External Supply/USB Not Present Constant-Current/Constant-Voltage Operation with Thermal Feedback to Maximize Charging Rate without Risk of Overheating Selectable 100% or 20% Current Limit (e.g., 500mA/ 100mA) from USB Input Preset 4.1V Charge Voltage with 0.8% Accuracy C/10 Charge Current Detection Output NTC Thermistor Input for Temperature Qualified Charging Tiny (6mm 3mm 0.75mm) 22-Pin DFN Package U APPLICATIO S ■ The LTC®4089-1 is a USB power manager plus high voltage Li-Ion battery charger. This device controls the total current used by the USB peripheral for operation and battery charging. Battery charge current is automatically reduced such that the sum of the load current and the charge current does not exceed the programmed input current limit. The LTC4089-1 also accommodates high voltage power supplies, such as 12V AC-DC wall adapters, FireWire, or automotive power. The LTC4089-1 provides a fixed 5V output from the high voltage input to charge single cell Li-Ion batteries. The charge current is programmable and an end-of-charge status output (CHRG) indicates full charge. Also featured is programmable total charge time, an NTC thermistor input used to monitor battery temperature while charging and automatic recharging of the battery. , LT, LTC and LTM are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Burst Mode is a registered trademark of Linear Technology Corporation. Protected by U.S. Patents including 6522118 and 6700364. Portable USB Devices—GPS Receivers, Cameras, MP3 Players, PDAs TYPICAL APPLICATIO U 0.1 F HIGH (6V-36V) VOLTAGE INPUT 1 F SW BOOST HVIN LTC4089-1 High Voltage Battery Charger Efficiency 10 H 90 CC CURRENT = 970mA 85 NO OUTPUT LOAD FIGURE 10 SCHEMATIC 80 WITH R PROG = 52k 10 F HVEN HVOUT IN 4.7 F HVPR LTC4089-1 1k 4.7 F OUT EFFICIENCY (%) 5V (NOM) FROM USB CABLE VBUS TO LDOs REGS, ETC. BAT 2k 65 60 55 HVIN = 8V HVIN = 12V HVIN = 24V HVIN = 36V 50 TIMER CLPROG GND PROG 0.1 F LTC4089-1 75 70 45 100k + Li-Ion BATTERY VOUT (TYP) 5V 5V VBAT AVAILABLE INPUT HV INPUT (LTC4089-1) USB ONLY BAT ONLY 40 2.5 3.5 3 4 BATTERY VOLTAGE (V) 4.5 4089-1 TA01b 4089-1 TAO1 40891f 1 LTC4089-1 AXI U RATI GS U W W W ABSOLUTE U W U PACKAGE/ORDER I FOR ATIO (Notes 1, 2, 3, 4, 5) TOP VIEW Terminal Voltage BOOST ...................................................... –0.3V to 50V BOOST above SW .....................................................25V HVIN, HVEN .............................................. –0.3V to 40V IN, OUT, HVOUT t < 1ms and Duty Cycle < 1% .................. –0.3V to 7V DC............................................................ –0.3V to 6V BAT .............................................................. –0.3V to 6V NTC, TIMER, PROG, CLPROG .......–0.3V to (VCC + 0.3V) CHRG, HPWR, SUSP, HVPR......................... –0.3V to 6V Pin Current, DC IN, OUT, BAT (Note 6) ..............................................2.5A Operating Temperature Range ................. –40°C to 85°C Maximum Operating Junction Temperature .......... 110°C Storage Temperature Range................... –65°C to 125°C GND 1 22 HVEN GND 2 21 HVIN HVOUT 3 20 BOOST VC 4 19 SW NTC 5 18 HVOUT 23 VNTC 6 HVPR 7 16 SUSP CHRG 8 15 HPWR PROG 9 14 CLPROG GATE 10 17 TIMER 13 OUT BAT 11 12 IN DJC PACKAGE 22-LEAD (6mm × 3mm) PLASTIC DFN EXPOSED PAD (PIN 23) IS GND (MUST BE SOLDERED TO PCB) TJMAX = 110°C, JA = 40°C/W ORDER PART NUMBER DJC PART MARKING LTC4089EDJC-1 40891 Order Options Tape and Reel: Add #TR Lead Free: Add #PBF Lead Free Tape and Reel: Add #TRPBF Lead Free Part Marking: http://www.linear.com/leadfree/ Consult LTC Marketing for parts specified with wider operating temperature ranges. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. HVIN = 12V, BOOST = 17V, VIN = 5V, VBAT = 3.7V, HVEN = 12V, HPWR = 5V, RPROG = 100k, RCLPROG = 2k, SUSP = 0V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS USB Input Current Limit VIN USB Input Supply Voltage IIN Input Bias Current IBAT = 0 (Note 7) Suspend Mode; SUSP = 5V ● ● ILIM Current Limit RCLPROG = 2k, HPWR = 5V RCLPROG = 2k, HPWR = 0V ● ● IIN(MAX) Maximum Input Current Limit (Note 8) RON ON Resistance VIN to VOUT IOUT = 80mA Load VCLPROG CLPROG Pin Voltage RCLPROG = 2k RCLPROG = 1k ISS Soft-Start Inrush Current VCLEN Input Current Limit Enable Threshold Voltage (VIN – VOUT) ● 4.35 475 90 5.5 V 0.5 50 1 100 mA µA 500 100 525 110 mA mA 2.4 A 0.215 ● ● 0.98 0.98 1.00 1.00 20 –80 50 –50 1.02 1.02 5 (VIN – VOUT) Rising (VIN – VOUT) Falling V V mA/µs 80 –20 mV mV 40891f 2 LTC4089-1 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. HVIN = 12V, BOOST = 17V, VIN = 5V, VBAT = 3.7V, HVEN = 12V, HPWR = 5V, RPROG = 100k, RCLPROG = 2k, SUSP = 0V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS VUVLO Input Undervoltage Lockout VIN Powers Part, Rising Threshold dVUVLO Input Undervoltage Lockout Hysteresis VIN Rising – VIN Falling ● MIN TYP MAX 3.6 3.8 4 130 UNITS V mV High Voltage Regulator VHVIN HVIN Supply Voltage 6 IHVIN HVIN Bias Current Not Switching Shutdown; HVEN = 0V VOUT Output Voltage with HVIN Present Assumes HVOUT to OUT Connection VHVUVLO High Voltage Input Undervoltage Lockout VHVIN Rising fSW Switching Frequency VHVOUT > 3.95V VHVOUT = 0V DCMAX Maximum Duty Cycle ISW(MAX) Switch Current Limit (Note 9) VSAT Switch VCESAT ISW = 1A ILK Switch Leakage Current VSWD Minimum Boost Voltage Above SW ISW = 1A IBST BOOST Pin Current ISW = 1A ● ● 36 V 1.9 0.01 2.5 2 mA µA 5 5.15 V 4.7 5 V 685 750 35 815 88 95 1.5 1.95 4.85 kHz kHz % 2.3 330 A mV 2 µA 1.85 2.2 V 30 50 mA Battery Management VBAT Input Voltage BAT IBAT Battery Drain Current VBAT = 4.3V, Charging Stopped Suspend Mode; SUSP = 5V VHVIN = VIN = 0V, BAT Powers OUT, No Load VFLOAT Regulated Output Voltage IBAT = 2mA IBAT = 2mA; (0°C – 85°C) ICHG Current Mode Charge Current RPROG = 100k, No Load RPROG = 50k, No Load; (0°C – 85°C) ICHG(MAX) Maximum Charge Current (Note 8) VPROG PROG Pin Voltage RPROG = 100k RPROG = 50k ● ● 0.98 0.98 1.00 1.00 1.02 1.02 V V kEOC Ratio of End-of-Charge Current to Charge Current VBAT = VFLOAT (4.1V) ● 0.085 0.1 0.11 mA/mA ITRIKL Trickle Charge Current VBAT = 2V, RPROG = 100k 35 50 60 mA VTRIKL Trickle Charge Threshold Voltage 2.75 2.9 3 V VCEN Charger Enable Threshold Voltage VRECHRG Recharge Battery Threshold Voltage VFLOAT - VRECHRG tTIMER TIMER Accuracy VBAT = 4.3V Recharge Time Percent of Total Charge Time 50 % Low Battery Trickle Charge Time Percent of Total Charge Time, VBAT < 2.8V 25 % 105 °C TLIM Junction Temperature in Constant Temperature Mode 4.3 V 15 22 60 27 35 100 µA µA µA 4.066 4.059 4.100 4.100 4.134 4.141 V V 465 900 500 1000 535 1080 mA mA ● ● ● ● 1.2 ● (VOUT – VBAT) Falling; VBAT = 4V (VOUT – VBAT) Rising; VBAT = 4V A 55 80 ● 65 100 –10 mV mV 135 mV 10 % 40891f 3 LTC4089-1 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. HVIN = 12V, BOOST = 17V, VIN = 5V, VBAT = 3.7V, HVEN = 12V, HPWR = 5V, RPROG = 100k, RCLPROG = 2k, SUSP = 0V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Internal Ideal Diode RFWD Incremental Resistance, VON Regulation IBAT = 100mA 125 m RDIO,ON ON Resistance VBAT to VOUT IBAT = 600mA 215 m VFWD Voltage Forward Drop (VBAT - VOUT) IBAT = 5mA IBAT = 100mA IBAT = 600mA VOFF Diode Disable Battery Voltage 2.8 V IFWD Load Current Limit, for VON Regulation 550 mA ID(MAX) Diode Current Limit 2.2 A 20 mV ● 10 30 55 160 50 mV mV mV External Ideal Diode VFWD, EXT External Diode Forward Voltage Logic VOL Output Low Voltage (CHRG, HVPR) ISINK = 5mA 0.1 VIH Input High Voltage HVEN, SUSP, HPWR Pin Low to High VIL Input Low Voltage HVEN, SUSP, HPWR Pin High to Low IPULLDN Logic Input Pull Down Current SUSP, HPWR 2 IHVEN HVEN Pin Bias Current VHVEN = 2.3V VHVEN = 0V 6 0.01 VCHG,SD Charger Shutdown Threshold Voltage on TIMER ICHG,SD Charger Shutdown Pull-Up Current on TIMER IVNTC ● 0.4 2.3 V V 0.3 V µA 20 0.1 µA µA 0.4 V ● 0.14 VTIMER = 0V ● 5 14 VNTC Pin Current VVNTC = 2.5V ● 1.4 2.5 VVNTC VNTC Bias Voltage IVNTC = 500µA ● 4.4 4.85 INTC NTC Input Leakage Current VNTC = 1V VCOLD Cold Temperature Fault Threshold Voltage Rising Threshold Hysteresis 0.738 • VVNTC 0.018 • VVNTC V V VHOT Hot Temperature Fault Threshold Voltage Falling Threshold Hysteresis 0.326 • VVNTC 0.015 • VVNTC V V VDIS NTC Disable Voltage NTC Input Voltage to GND (Falling) Hysteresis µA NTC Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: VCC is the greater of VIN, VOUT or VBAT Note 3: All voltage values are with respect to GND. Note 4: This IC includes over-temperature protection that is intended to protect the device during momentary overload conditions. Junction temperatures will exceed 110°C when over-temperature protection is active. Continuous operation above the specified maximum operating junction temperature may result in device degradation or failure. 4 0 ● 75 100 35 3.5 mA V ±1 125 µA mV mV Note 5: The LTC4089-1 is guaranteed to meet specified performance from 0°C to 85°C and are designed, characterized and expected to meet these extended temperature limits, but is not tested at –40°C and 85°C. Note 6: Guaranteed by long term current density limitations. Note 7: Total input current is equal to this specification plus 1.002 • IBAT where IBAT is the charge current. Note 8: Accuracy of programmed current may degrade for currents greater than 1.5A. Note 9: Current limit guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at high duty cycle. 40891f LTC4089-1 TYPICAL PERFORMANCE CHARACTERISTICS Battery Regulation (Float) Voltage vs Temperature VFLOAT Load Regulation 4.20 TA = 25°C, unless otherwise specified. 4.120 RPROG = 34k 5 VIN = 5V IBAT = 2mA 4.115 4.15 Battery Current and Voltage vs Time 4 VFLOAT (V) VFLOAT (V) 4.00 4.105 4.100 4.095 4.090 3.95 3.90 200 400 600 IBAT (mA) 800 4.080 –50 1000 –25 0 50 25 TEMPERATURE (°C) 75 0 100 1250mAh CELL HVIN = 12V RPROG = 50k VIN = 5V VOUT = NO LOAD 500 RPROG = 100k RCLPROG = 2k VBAT = 3.7V 900 VIN = 0V 800 700 1.5 IOUT (mA) IBAT (mA) 1 300 2 2.5 VBAT (V) 3 3.5 4 4.5 100 VIN = 5V VBAT = 3.5V θJA = 50°C/W 0 50 –50 –25 25 75 0 TEMPERATURE (°C) 100 –50 0 50 100 0 125 0 50 100 VFWD (mV) 150 4089-1 G05 C C C C 200 4089-1 G06 High Voltage Regulator Efficiency vs Output Load 100 VBAT = 3.7V 4500 VIN = 0V Si2333 PFET 4000 HVIN = 8V 95 HVIN = 12V 90 EFFICIENCY (%) 3500 3000 2500 2000 1500 –50°C 0°C 50°C 100°C 1000 500 20 400 100 5000 0 500 300 4089-1 G04 0 600 200 Ideal Diode Current vs Forward Voltage and Temperature with External Device IOUT (mA) 0.5 0 200 150 1000 200 HPWR = 5V HPWR = 0V 0 100 4089-1 G03 400 100 50 Ideal Diode Current vs Forward Voltage and Temperature (No External Device) 500 400 200 0 300 TERMINATION TIME (MIN) 600 600 600 C/10 Charge Current vs Temperature (Thermal Regulation) 300 900 4089-1 G02 Charging from USB, IBAT vs VBAT IBAT (mA) 2 1 4089-1 G01 0 3 4.085 0 1200 VBAT VOUT VCHRG IBAT IBAT (mA) 4.05 VBAT, VOUT, VCHRG (V) 4.110 4.10 1500 60 40 VFWD (mV) 80 100 4089-1 G17 85 80 75 HVIN = 24V 70 HVIN = 36V 65 60 FIGURE 10 SCHEMATIC VBAT = 4.11V (IBAT = 0) 55 50 0 0.2 0.6 0.4 IOUT (A) 0.8 1.0 4089-1 G09 40891f 5 LTC4089-1 TYPICAL PERFORMANCE CHARACTERISTICS High Voltage Regulator Maximum Load Current, L = 10µH High Voltage Regulator Maximum Load Current, L = 33µH 1.6 1.8 1.5 1.6 TYPICAL 500 TYPICAL 450 1.3 1.2 MINIMUM 1.1 1.0 0.9 0.9 10 15 MINIMUM 20 VIN (V) 25 30 35 50 5 10 15 20 VIN (V) 25 30 720 700 680 660 640 25 50 75 100 125 150 TEMPERATURE (°C) 2.0 1.8 700 600 500 400 300 200 0 0 1.9 6.8 1.8 6.6 1.7 6.4 INPUT VOLTAGE (V) CURRENT LIMIT (A) 1.0 0.8 0.6 0.4 0 0 0.25 0.50 0.75 1 1.25 1.50 1.75 SHDN PIN VOLTAGE (V) 1.6 1.5 1.4 1.3 TA = –40°C TA = –5°C TA = 25°C TA = 90°C 2 4089-1 G15 High Voltage Regulator Typical Minimum Input Voltage 7.0 0 1.2 4089-1 G14 2.0 1.0 1.4 100 200 300 400 500 600 700 800 FEEDBACK VOLTAGE (mV) High Voltage Switch Current Limit 1.1 1.6 0.2 4089-1 G13 1.2 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 SWITCH CURRENT (A) High Voltage Regulator Soft-Start 100 620 0 4089-1 G12 SWITCH CURRENT LIMIT (A) SWITCHING FREQUENCY (kHz) FREQUENCY (kHz) 0 35 800 780 740 TA = –40°C 200 High Voltage Regulator Frequency Foldback 800 0 300 250 4089-1 G11 High Voltage Regulator Switch Frequency 600 –50 –25 350 100 4089-1 G10 760 TA = 25°C 150 1.1 1.0 5 VCE(SW) (mV) IOUT (A) 1.2 TA = 85°C 400 1.4 1.3 High Voltage Regulator Switch Voltage Drop 550 1.5 1.4 IOUT (A) TA = 25°C, unless otherwise specified. TO START 6.2 6.0 5.8 TO RUN 5.6 5.4 5.2 10 20 30 40 50 60 70 80 90 100 DUTY CYCLE (%) 4089-1 G16 5.0 1 10 100 LOAD CURRENT (mA) 1000 4089-1 G17 40891f 6 LTC4089-1 TYPICAL PERFORMANCE CHARACTERISTICS Input Disconnect Waveforms Input Connect Waveforms VIN 5V/DIV VOUT 5V/DIV VIN 5V/DIV VOUT 5V/DIV IIN 0.5A/DIV IBAT 0.5A/DIV IIN 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.85V IOUT = 100mA 1ms/DIV TA = 25°C, unless otherwise specified. 4089-1 G18 Response to HPWR HPWR 5V/DIV IIN 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.85V IOUT = 100mA 1ms/DIV 4089-1 G19 VBAT = 3.85V IOUT = 50mA Wall Disconnect Waveforms Wall Connect Waveforms SUSP 5V/DIV VOUT 5V/DIV IWALL 0.5A/DIV IBAT 0.5A/DIV IWALL 0.5A/DIV IBAT 0.5A/DIV 4089-1 G20 Response to Suspend WALL 5V/DIV WALL 5V/DIV VOUT 5V/DIV 100 s/DIV VOUT 5V/DIV IIN 0.5A/DIV IBAT 0.5A/DIV VBAT = 3.85V IOUT = 100mA RPROG = 100k 1ms/DIV 4089-1 G21 VBAT = 3.85V IOUT = 100mA RPROG = 100k 1ms/DIV 4089-1 G22 High Voltage Regulator Load Transient 100 s/DIV 4089-1 G23 High Voltage Regulator Load Transient HVOUT 50mV/DIV HVOUT 50mV/DIV IOUT 0.5A/DIV IL 0.5A/DIV 20 S/DIV VBAT = 3.85V IOUT = 50mA 4089-1 G24 20 S/DIV 4089-1 G25 40891f 7 LTC4089-1 PI FU CTIO S U U U GND (Pins 1, 2): Ground. Tie the GND pin to a local ground plane below the LTC4089-1 and the circuit components. HVOUT (Pins 3, 18): Voltage Output of the High Voltage Regulator. When sufficient voltage is present at HVOUT, the low voltage power path from IN to OUT will be disconnected and the HVPR pin will be pulled low to indicate that a high voltage wall adapter has been detected. The LTC4089-1 high voltage regulator will provide a fixed 5V output to the battery charger MOSFET. HVOUT should be bypassed with at least 10µF to GND. Connect pins 3 and 18 with a resistance no greater than 1 . VC (Pin 4): Leave the VC pin floating or bypass to ground with a 10pF capacitor. This optional 10pF capacitor reduces HVOUT ripple in discontinuous mode. NTC (Pin 5): Input to the NTC Thermistor Monitoring Circuits. The NTC pin connects to a negative temperature coeffcient thermistor which is typically co-packaged with the battery pack to determine if the battery is too hot or too cold to charge. If the battery’s temperature is out of range, charging is paused until the battery temperature reenters the valid range. A low drift bias resistor is required from VNTC to NTC and a thermistor is required from NTC to ground. If the NTC function is not desired, the NTC pin should be grounded. VNTC (Pin 6): Output Bias Voltage for NTC. A resistor from this pin to the NTC pin will bias the NTC thermistor. HVPR (Pin 7): High Voltage Present Output. Active low open drain output pin. A low on this pin indicates that the high voltage regulator has sufficient voltage to charge the battery. This feature is disabled if no power is present on HVIN, IN or BAT (i.e., below UVLO thresholds). H R G (Pin 8): Open-Drain Charge Status Output. When the C battery is being charged, the CHRG pin is pulled low by an internal N-channel MOSFET. When the timer runs out or the charge current drops below 10% of the programmed charge current or the input supply is removed, the CHRG pin is forced to a high impedance state. PROG (Pin 9): Charge Current Program. Connecting a resistor, RPROG, to ground programs the battery charge current. The battery charge current is programmed as follows: 50, 000 V ICHG( A) = RPROG GATE (Pin 10): External ideal diode gate pin. This pin can be used to drive the gate of an optional external PFET connected between BAT (drain) and OUT (source). By doing so, the impedance of the ideal diode between BAT and OUT can be reduced. When not in use, this pin should be left floating. It is important to maintain a high impedance on this pin and minimize all leakage paths. BAT (Pin 11): Connect to a single cell Li-Ion battery. This pin is used as an output when charging the battery and as an input when supplying power to OUT. When the OUT pin potential drops below the BAT pin potential, an ideal diode function connects BAT to OUT and prevents VOUT from dropping more than 100mV below VBAT. A precision internal resistor divider sets the final float (charging) potential on this pin. The internal resistor divider is disconnected when IN and HVIN are in undervoltage lockout. IN (Pin 12): Input Supply. Connect to USB supply, VBUS. Input current to this pin is limited to either 20% or 100% of the current programmed by the CLPROG pin as determined by the state of the HPWR pin. Charge current (to the BAT pin) supplied through the input is set to the current programmed by the PROG pin but will be limited by the input current limit if charge current is set greater than the input current limit. OUT (Pin 13): Voltage Output. This pin is used to provide controlled power to a USB device from either USB VBUS (IN), an external high voltage supply (HVIN), or the battery (BAT) when no other supply is present. The high voltage supply is prioritized over the USB VBUS input. OUT should be bypassed with at least 4.7µF to GND. 40891f 8 LTC4089-1 PI FU CTIO S U U U CLPROG (Pin 14): Current Limit Program and Input Current Monitor. Connecting a resistor, RCLPROG, to ground programs the input to output current limit. The current limit is programmed as follows: 1000 V ICL ( A) = R CLPROG In USB applications, the resistor RCLPROG should be set to no less than 2.1k. The voltage on the CLPROG pin is always proportional to the current flowing through the IN to OUT power path. This current can be calculated as follows: V IIN( A) = CLPROG • 1000 RCLPROG HPWR (Pin 15): High Power Select. This logic input is used to control the input current limit. A voltage greater than 2.3V on the pin will set the input current limit to 100% of the current programmed by the CLPROG pin. A voltage less than 0.3V on the pin will set the input current limit to 20% of the current programmed by the CLPROG pin. A 2µA pull-down current is internally connected to this pin to ensure it is low at power up when the pin is not being driven externally. SUSP (Pin 16): Suspend Mode Input. Pulling this pin above 2.3V will disable the power path from IN to OUT. The supply current from IN will be reduced to comply with the USB specification for suspend mode. Both the ability to charge the battery from HVIN and the ideal diode function (from BAT to OUT) will remain active. Suspend mode will reset the charge timer if VOUT is less than VBAT while in suspend mode. If VOUT is kept greater than VBAT, such as when the high voltage input is present, the charge timer will not be reset when the part is put in suspend. A 2µA pull-down current is internally applied to this pin to ensure it is low at power-up when the pin is not being driven externally. TIMER (Pin 17): Timer Capacitor. Placing a capacitor, CTIMER, to GND sets the timer period. The timer period is: t TIMER (hours) = C TIMER • R PROG • 3hours 0 . 1µF • 100k Charge time is increased if charge current is reduced due to undervoltage current limit, load current, thermal regulation and current limit selection (HPWR). Shorting the TIMER pin to GND disables the battery charging functions. SW (Pin 19): The SW pin is the output of the internal high voltage power switch. Connect this pin to the inductor, catch diode and boost capacitor. BOOST (Pin 20): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. HVIN (Pin 21): The HVIN pin supplies current to the internal high voltage regulator and to the internal high voltage power switch. The presence of a high voltage input takes priority over the USB VBUS input (i.e., when a high voltage input supply is detected, the USB IN to OUT path is disconnected). This pin must be locally bypassed. HVEN (Pin 22): The HVEN pin is used to disable the high voltage input path. Tie to ground to disable the high voltage input or tie to at least 2.3V to enable the high voltage path. If this feature is not used, tie to the HVIN pin. This pin can also be used to soft-start the high voltage regulator; see the Applications Information section. EXPOSED PAD (Pin 23): Ground. The exposed package pad is ground and must be soldered to the PC board for proper functionality and for maximum heat transfer (use several vias directly under the LTC4089-1). 40891f 9 LTC4089-1 BLOCK DIAGRA W D2 BOOST 10 C3 HVIN L1 SW Q1 + + D1 – R S – Q Q DRIVER DRIVER OSCILLATOR 10 VSET 5V – VC GM 10pF + PART NUMBER LTC4089-1 HVOUT VSET 5V C1 – 1.8V ENABLE R3 10 + – HVEN + 75mV (RISING) 25mV (FALLING) C4 + CURRENT LIMIT 22 1V + CLPROG – DIE TEMP 2k 13 HPWR ILIM CNTL ILIM CURRENT CONTROL CL IN ENABLE 21 CC/CV REGULATOR CHARGER ENABLE 105°C 500mA/100mA + – 2µA 25mV + EDA IN OUT BAT 21 21 BAT + ICHG SOFT-START2 1V GATE – BAT CHARGE CONTROL OUT 25mV IDEAL DIODE TA + + – IIN 1000 SOFT-START HVPR 19 – 4.25V (RISING) 3.15V (FALLING) + – 10 IN – 0.25V + 2.8V BATTERY UVLO CHG – 23 PROG – 100k VOLTAGE DETECT 15 VNTC + UVLO – 10k 14 RECHRG NTCERR + NTC TIMER OSCILLATOR CONTROL LOGIC HOLD NTC – 100k – BAT UV TOO COLD RESET TOO HOT 4V RECHARGE CHRG CLK COUNTER STOP 21 18 + + 0.1V C/10 EOC NTC ENABLE 2µA – 16 GND 11 SUSP 4089-1 TA01 40891f 10 U OPERATIO LTC4089-1 (Refer to Block Diagram) The LTC4089-1 is a complete PowerPath™ controller for battery powered USB applications. The LTC4089-1 is designed to receive power from a low voltage source (e.g., USB or 5V wall adapter), a high voltage source (e.g., FireWire/IEEE1394, automotive battery, 12V wall adapter, etc.), and a single-cell Li-Ion battery. It can then deliver power to an application connected to the OUT pin and a battery connected to the BAT pin (assuming that an external supply other than the battery is present). Power supplies that have limited current resources (such as USB VBUS supplies) should be connected to the IN pin which has a programmable current limit. Battery charge current will be adjusted to ensure that the sum of the charge current and load current does not exceed the programmed input current limit (see Figure 1). forward biased. The forward biased ideal diode will then provide the output power to the load from the battery. An ideal diode function provides power from the battery when output / load current exceeds the input current limit or when input power is removed. Powering the load through the ideal diode instead of connecting the load directly to the battery allows a fully charged battery to remain fully charged until external power is removed. Once external power is removed the output drops until the ideal diode is Input Current Limit HVIN The LTC4089-1 also includes a high voltage switching regulator which has the ability to receive power from a high voltage input. This input takes priority over the USB VBUS input (i.e., if both HVIN and IN are present, load current and charge current will be delivered via the high voltage path). When enabled, the high voltage regulator regulates the HVOUT voltage using a 750kHz constant frequency, current mode regulator. An external PFET between HVOUT (drain) and OUT (source) is turned on via the HVPR pin allowing OUT to charge the battery and/or supply power to the application. The LTC4089-1 provides a fixed 5V output. Whenever the input power path is enabled (i.e., SUSP = 0V and HVIN = 0V) and power is available at IN, power is delivered to OUT. The current limit and charger control circuits of the LTC4089-1 are designed to limit input current as well as control battery charge current as a function PowerPath is a registered trademark of Linear Technology Corporation. SW Q1 L1 D1 HIGH VOLTAGE BUCK REGULATOR HVOUT C1 + 4.25V (RISING) 3.15V (FALLING) – HVPR 19 + – + – ENABLE LOAD 75mV (RISING) 25mV (FALLING) OUT 21 USB CURRENT LIMIT CC/CV REGULATOR CHARGER IDEAL DIODE + – 25mV + – IN OUT 25mV + EDA 21 GATE – BAT 21 4089-1 F01 Figure 1. Simplified PowerPath Block Diagram BAT + LI-ION 40891f 11 LTC4089-1 U OPERATIO of IOUT. The input current limit, ICL, can be programmed using the following formula: 1000 1000 V ICL = • VCLPROG = RCLPROG RCLPROG where VCLPROG is the CLPROG pin voltage (typically 1V) and RCLPROG is the total resistance from the CLPROG pin to ground. For best stability over temperature and time, 1% metal film resistors are recommended. The programmed battery charge current, I CHG , is defined as: 50, 000 50, 000 V ICHG = • VPROG = RPROG RPROG Input current, IIN, is equal to the sum of the BAT pin output current and the OUT pin output current. VCLPROG will typically servo to 1V, however, if IOUT + IBAT < ICL then VCLPROG will track the input current according to the following equation: V IIN = IOUT + IBAT = CLPROG • 1000 RCLPROG The current limiting circuitry in the LTC4089-1 can and should be configured to limit current to 500mA for USB applications (selectable using the HPWR pin and programmed using the CLPROG pin). IIN 500 The LTC4089-1 reduces battery charge current such that the sum of the battery charge current and the load current does not exceed the programmed input current limit (onefifth of the programmed input current limit when HPWR is low, see Figure 2). The battery charge current goes to zero when load current exceeds the programmed input current limit (one-fifth of the limit when HPWR is low). Even if the battery charge current is set to exceed the allowable USB current, the USB specification will not be violated. The battery charger will reduce its current as needed to ensure that the USB specification is not exceeded. If the load current is greater than the current limit, the output voltage will drop to just under the battery voltage where the ideal diode circuit will take over and the excess load current will be drawn from the battery. In USB applications, the minimum value for RCLPROG should be 2.1k. This will prevent the input current from exceeding 500mA due to LTC4089-1 tolerances and quiescent currents. A 2.1k CLPROG resistor will give a typical current limit of 476mA in high power mode (HPWR = 1) or 95mA in low power mode (HPWR = 0). When SUSP is driven to a logic high, the input power path is disabled and the ideal diode from BAT to OUT will supply power to the application. IIN 100 500 IIN 80 200 100 400 ILOAD 60 40 20 IBAT IBAT (CHARGING) 0 0 4089-1 F02a 100 200 300 CURRENT (mA) ILOAD 300 CURRENT (mA) CURRENT (mA) 400 ILOAD 300 IBAT = ICHG 200 IBAT = ICL = IOUT 100 IBAT (CHARGING) (CHARGING) 400 500 IBAT ILOAD(mA) (IDEAL DIODE) (a) High Power Mode/Full Charge RPROG = 100k and RCLPROG = 2k 0 0 4089-1 F02b 20 40 60 80 100 IBAT ILOAD(mA) (IDEAL DIODE) (b) Low Power Mode/Full Charge RPROG = 100k and RCLPROG = 2k 0 0 4089-1 F02c 100 200 300 400 500 IBAT ILOAD (mA) (IDEAL DIODE) (c) High Power Mode with ICL = 500mA and ICHG = 250mA RPROG = 200k and RCLPROG = 2k Figure 2. Input and Battery Currents as a Function of Load Current 40891f 12 LTC4089-1 U OPERATIO High Voltage Step Down Regulator Ideal Diode from BAT to OUT The power delivered from HVIN to HVOUT is controlled by a 750kHz constant frequency, current mode step down regulator. An external P-channel MOSFET directs this power to OUT and prevents reverse conduction from OUT to HVOUT (and ultimately HVIN). The LTC4089-1 has an internal ideal diode as well as a controller for an optional external ideal diode. If a battery is the only power supply available, or if the load current exceeds the programmed input current limit, then the battery will automatically deliver power to the load via an ideal diode circuit between the BAT and OUT pins. The ideal diode circuit (along with the recommended 4.7µF capacitor on the OUT pin) allows the LTC4089-1 to handle large transient loads and wall adapter or USB VBUS connect/disconnect scenarios without the need for large bulk capacitors. The ideal diode responds within a few microseconds and prevents the OUT pin voltage from dropping significantly below the BAT pin voltage. A comparison of the I-V curve of the ideal diode and a Schottky diode can be seen in Figure 3. A 750kHz oscillator enables an RS flip-flop, turning on the internal 1.95A power switch Q1. An amplifier and comparator monitor the current flowing between the HVIN and SW pins, turning the switch off when this current reaches a level determined by the voltage at VC. An error amplifier servos the VC node to maintain 5V at HVOUT. An active clamp on the VC node provides current limit. The VC node is also clamped to the voltage on the HVEN pin; soft-start is implemented by a voltage ramp at the HVEN pin using an external resistor and capacitor. An internal regulator provides power to the control circuitry. This regulator includes an undervoltage lockout to prevent switching when HVIN is less than about 4.7V. The HVEN pin is used to disable the high voltage regulator. HVIN input current is reduced to less than 2µA and the external P-channel MOSFET disconnects HVOUT from OUT when the high voltage regulator is disabled. The switch driver operates from either the high voltage input or from the BOOST pin. An external capacitor and diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to fully saturate the internal bipolar NPN power switch for efficient operation. If the input current increases beyond the programmed input current limit additional current will be drawn from the battery via the internal ideal diode. Furthermore, if power to IN (USB VBUS) or HVIN (high voltage input) is removed, then all of the application power will be provided by the battery via the ideal diode. A 4.7µF capacitor at OUT is sufficient to keep a transition from input power to battery power from causing significant output voltage droop. The ideal diode consists of a precision amplifier that enables a large P-channel MOSFET transistor whenever the voltage at OUT is approximately 20mV (VFWD) below the voltage at BAT. The resistance of the internal ideal diode is approximately 200m . If this is sufficient for the When HVOUT is below 3.95V the operating frequency is reduced. This frequency foldback helps to control the regulator output current during start-up and overload. LTC4089 CONSTANT I0N SLOPE: 1/RDIO(ON) CONSTANT R0N CURRENT (A) IMAX IFWD SLOPE: 1/RFWD 0 VFWD SCHOTTKY DIODE CONSTANT V0N FORWARD VOLTAGE (V) 4089-1 F03 Figure 3. LTC4089-1 Versus Schottky Diode Forward Voltage Drop 40891f 13 LTC4089-1 U OPERATIO application then no external components are necessary. However, if more conductance is needed, an external P-channel MOSFET can be added from BAT to OUT. The GATE pin of the LTC4089-1 drives the gate of the PFET for automatic ideal diode control. The source of the external MOSFET should be connected to OUT and the drain should be connected to BAT. In order to help protect the external MOSFET in over-current situations, it should be placed in close thermal contact to the LTC4089-1. Battery Charger The battery charger circuits of the LTC4089-1 are designed for charging single cell lithium-ion batteries. Featuring an internal P-channel power MOSFET, the charger uses a constant-current/constant-voltage charge algorithm with programmable current and a programmable timer for charge termination. Charge current can be programmed up to 1.2A. The final float voltage accuracy is ±0.8% typical. No blocking diode or sense resistor is required when powering either the IN or the HVIN pins. The CHRG open-drain status output provides information regarding the charging status of the LTC4089-1 at all times. An NTC input provides the option of charge qualification using battery temperature. An internal thermal limit reduces the programmed charge current if the die temperature attempts to rise above a preset value of approximately 115°C. This feature protects the LTC4089-1 from excessive temperature, and allows the user to push the limits of the power handling capability of a given circuit board without risk of damaging the LTC4089-1. Another benefit of the LTC4089-1 thermal limit is that charge current can be set according to typical, not worst-case, ambient temperatures for a given application with the assurance that the charger will automatically reduce the current in worst-case conditions. The charge cycle begins when the voltage at the OUT pin rises above the battery voltage and the battery voltage is below the recharge threshold. No charge current actually flows until the OUT voltage is 100mV above the BAT voltage. At the beginning of the charge cycle, if the battery voltage is below 2.8V, the charger goes into trickle charge mode to bring the cell voltage up to a safe level for charging. The charger goes into the fast charge constant-current mode once the voltage on the BAT pin rises above 2.8V. In constant current mode, the charge current is set by RPROG. When the battery approaches the final float voltage, the charge current begins to decrease as the LTC4089-1 switches to constant-voltage mode. When the charge current drops below 10% of the programmed charge current while in constant-voltage mode the CHRG pin assumes a high impedance state. An external capacitor on the TIMER pin sets the total minimum charge time. When this time elapses the charge cycle terminates and the CHRG pin assumes a high impedance state, if it has not already done so. While charging in constant current mode, if the charge current is decreased by thermal regulation or in order to maintain the programmed input current limit the charge time is automatically increased. In other words, the charge time is extended inversely proportional to the actual charge current delivered to the battery. For Li-Ion and similar batteries that require accurate final float potential, the internal bandgap reference, voltage amplifier and the resistor divider provide regulation with ±0.8% accuracy. Trickle Charge and Defective Battery Detection At the beginning of a charge cycle, if the battery voltage is low (below 2.8V) the charger goes into trickle charge reducing the charge current to 10% of the fullscale current. If the low battery voltage persists for one quarter of the total charge time, the battery is assumed to be defective, the charge cycle is terminated and the CHRG pin output assumes a high impedance state. If for any reason the battery voltage rises above ~2.8V the charge cycle will be restarted. To restart the charge cycle (i.e., when the dead battery is replaced with a discharged battery), simply remove the input voltage and reapply it or cycle the TIMER pin to 0V. Programming Charge Current The formula for the battery charge current is: V ICHG = IPROG • 50, 000 = PROG • 50, 000 RPROG where VPROG is the PROG pin voltage and RPROG is the total resistance from the PROG pin to ground. Keep in mind that when the LTC4089-1 is powered from the IN pin, 40891f 14 LTC4089-1 U OPERATIO the programmed input current limit takes precedent over the charge current. In such a scenario, the charge current cannot exceed the programmed input current limit. For example, if typical 500mA charge current is required, calculate: 1V RPROG = • 50, 000 = 100k 500mA For best stability over temperature and time, 1% metal film resistors are recommended. Under trickle charge conditions, this current is reduced to 10% of the full-scale value. The Charge Timer The programmable charge timer is used to terminate the charge cycle. The timer duration is programmed by an external capacitor at the TIMER pin. The charge time is typically: C •R • 3hours t TIMER (hours) = TIMER PROG 0 . 1µF • 100k The timer starts when an input voltage greater than the undervoltage lockout threshold level is applied or when leaving shutdown and the voltage on the battery is less than the recharge threshold. At power-up or exiting shutdown with the battery voltage less than the recharge threshold the charge time is a full cycle. If the battery is greater than the recharge threshold the timer will not start and charging is prevented. If after power-up the battery voltage drops below the recharge threshold, or if after a charge cycle the battery voltage is still below the recharge threshold, the charge time is set to one-half of a full cycle. The LTC4089-1 has a feature that extends charge time automatically. Charge time is extended if the charge current in constant current mode is reduced due to load current or thermal regulation. This change in charge time is inversely proportional to the change in charge current. As the LTC4089-1 approaches constant voltage mode the charge current begins to drop. This change in charge current is due to normal charging operation and does not affect the timer duration. Consider, for example, a USB charge condition where RCLPROG = 2k, RPROG = 100k and CTIMER = 0.1µF. This corresponds to a three hour charge cycle. However, if the HPWR input is set to a logic low, then the input current limit will be reduced from 500mA to 100mA. With no additional system load, this means the charge current will be reduced to 100mA. Therefore, the termination timer will automatically slow down by a factor of five until the charger reaches constant voltage mode (i.e. VBAT = 4.1V) or HPWR is returned to a logic high. The charge cycle is automatically lengthened to account for the reduced charge current. The exact time of the charge cycle will depend on how long the charger remains in constant-current mode and/or how long the HPWR pin remains a logic low. Once a time-out occurs and the voltage on the battery is greater than the recharge threshold, the charge current stops, and the CHRG output assumes a high impedance state if it has not already done so. Connecting the TIMER pin to ground disables the battery charger. CHRG Status Output Pin When the charge cycle starts, the CHRG pin is pulled to ground by an internal N-channel MOSFET capable of driving an LED. When the charge current drops below 10% of the programmed full charge current while in constant voltage mode, the pin assumes a high impedance state, but charge current continues to flow until the charge time elapses. If this state is not reached before the end of the programmable charge time, the pin will assume a high impedance state when a time-out occurs. The CHRG current detection threshold can be calculated by the following equation: IDETECT = 0 . 1V 5000 V • 50, 000 = RPROG RPROG For example, if the full charge current is programmed to 500mA with a 100k PROG resistor the CHRG pin will change state at a battery charge current of 50mA. Note: The end-of-charge (EOC) comparator that monitors the charge current latches its decision. Therefore, the first time the charge current drops below 10% of the programmed full charge current while in constant voltage mode, it will toggle CHRG to a high impedance state. If, for some reason the charge current rises back above 40891f 15 LTC4089-1 U OPERATIO the threshold, the CHRG pin will not resume the strong pull-down state. The EOC latch can be reset by a recharge cycle (i.e., VBAT drops below the recharge threshold) or toggling the input power to the part. NTC Thermistor—Battery Temperature Charge Qualification The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the battery pack. The NTC circuitry is shown in Figure 4. To use this feature, connect the NTC thermistor (RNTC) between the NTC pin and ground and a resistor (RNOM) from the NTC pin to VNTC. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC thermistor at 25°C (this value is 10k for a Vishay NTHS0603N02N1002J thermistor). The LTC4089-1 goes into hold mode when the resistance (RHOT) of the NTC thermistor drops to 0.48 times the value of RNOM, or approximately 4.8k, which should be at 45°C. The hold mode freezes the timer and stops the charge cycle until the thermistor indicates a return to a valid temperature. As the temperature drops, the resistance of the NTC thermistor rises. The LTC4089-1 is designed to go into hold mode when the value of the NTC thermistor increases to 2.82 times the value of RNOM. This resistance is RCOLD. For a Vishay NTHS0603N02N1002J thermistor, this value is 28.2k which corresponds to approximately 0°C. The hot and cold comparators each have approximately 2°C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin will disable the NTC function. Current Limit Undervoltage Lockout An internal undervoltage lockout circuit monitors the input voltage and disables the input current limit circuits until VIN rises above the undervoltage lockout threshold. The current limit UVLO circuit has a built-in hysteresis of 125mV. Furthermore, to protect against reverse current in the power MOSFET, the current limit UVLO circuit disables the current limit (i.e., forces the input power path to a high impedance state) if VOUT exceeds VIN. If the current limit UVLO comparator is tripped, the current limit circuits will not come out of shutdown until VOUT falls 50mV below the VIN voltage. Charger Undervoltage Lockout An internal undervoltage lockout circuit monitors the VOUT voltage and disables the battery charger circuits until VOUT rises above the undervoltage lockout threshold. The battery charger UVLO circuit has a built-in hysteresis of 125mV. Furthermore, to protect against reverse current in the power MOSFET, the charger UVLO circuit keeps the charger shut down if VBAT exceeds VOUT. If the charger UVLO comparator is tripped, the charger circuits will not come out of shutdown until VOUT exceeds VBAT by 50mV. Suspend The LTC4089-1 can be put in suspend mode by forcing the SUSP pin greater than 2.3V. In suspend mode, the ideal diode function from BAT to OUT is kept alive. If power is applied to the HVIN pin, then charging will be unaffected. Current drawn from the IN pin is reduced to 50µA. Suspend mode is intended to comply with the USB power specification mode of the same name. VNTC LTC4089-1 6 RNOM 10k NTC 0.738 • VNTC – TOO_COLD 5 + RNTC 10k – 0.326 • VNTC TOO_HOT + + NTC_ENABLE 0.1V – 4089-1 F04 Figure 4. NTC Circuit 40891f 16 LTC4089-1 APPLICATIO S I FOR ATIO U U W U USB and 5V Wall Adapter Power Although the LTC4089-1 is designed to draw power from a USB port, a higher power 5V wall adapter can also be used to power the application and charge the battery (higher voltage wall adapters can be connected directly to HVIN). Figure 5 shows an example of combining a 5V wall adapter and a USB power input. With its gate grounded by 1k, P-channel MOSFET MP1 provides USB power to the LTC4089-1 when 5V wall power is not available. When 5V wall power is available, D1 both supplies power to the LTC4089, pulls the gate of MN1 high to increase the charge current (by increasing the input current limit), and pulls the gate of MP1 high to disable it and prevent conduction back to the USB port. 5V WALL ADAPTER 850mA ICHG USB POWER 500mA ICHG ICHG BAT D1 LTC4089-1 MP1 1k IN + PROG CLPROG MN1 2.87k 2k Li-Ion BATTERY 59k will be the programmed charge current plus the largest expected application load current. For robust operation in fault conditions, the saturation current should be ~2.3A. To keep efficiency high, the series resistance (DCR) should be less than 0.1 . Table 1 lists several vendors and types that are suitable. Table 1: Inductor Vendors VENDOR Sumida URL PART SERIES INDUCTANCE (µH) SIZE (mm) 8.2, 10 6 6 3 CDRH6D38 10 7 7 4 www.sumida.com CDRH5D28 TDK www.tdk.com SLF6028T 10 6 6 2.8 Toko www.toko.com D63LCB 10 6.3 6.3 3 Catch Diode Depending on load current, a 1A to 2A Schottky diode is recommended for the D1 catch diode. The diode must have a reverse voltage rating equal to, or greater than, the maximum input voltage. The ON Semiconductor MBRM140 and the Diodes Inc. DFLS140/160/240 are good choices. High Voltage Regulator Capacitor Selection 4089-1 F05 Figure 5. USB or 5V Wall Adapter Power Inductor Selection and Maximum Output Current A good choice for the inductor value is L = 10µH. With this value the maximum load current will be 1A. The RMS current rating of the inductor must be greater than the maximum load current and its saturation current should be about 30% higher. Note that the maximum load current Bypass the HVIN pin of the LTC4089-1 circuit with a 1µF, or higher value ceramic capacitor of X7R or X5R type. Y5V types have poor performance over temperature and applied voltage and should not be used. A 1µF ceramic is adequate to bypass the high voltage input and will easily handle the ripple current. However, if the input power source has high impedance, or there is significant inductance due to long wires or cables, additional bulk capacitance may be necessary. This can be provided with a low performance electrolytic capacitor. 40891f 17 LTC4089-1 APPLICATIO S I FOR ATIO U U W U The high voltage regulator output capacitor controls output ripple, supplies transient load currents, and stabilizes the regulator control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple performance. A good value is 10µF. Use X5R or X7R types, and note that a ceramic capacitor biased with VHVOUT will have less than its nominal capacitance. Table 2 lists several capacitor vendors. Table 2: Capacitor Vendors VENDOR PHONE Panasonic (714) 373-7366 URL www.panasonic.com PART SERIES COMMENTS Ceramic, EEF Series Polymer, Tantalum Kemet (864) 963-6300 www.kemet.com Ceramic, Tantalum T494, T495 Sanyo (408) 749-9714 www.sanyovideo.com Ceramic, Polymer, Tantalum POSCAP Murata (404) 436-1300 www.murata.com Ceramic www.avxcorp.com Ceramic, Tantalum www.taiyo-yuden.com Ceramic AVX Taiyo Yuden (864) 963-6300 TPS Series BOOST Pin Considerations Capacitor C3 and diode D2 (see Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases, a 0.1µF capacitor and fast-switching diode (such as the 1N4148 or 1N914) will work well. The BOOST pin must be at least 2.2V above the SW pin for proper operation. High Voltage Regulator Soft-Start The HVEN pin can be used to soft-start the high voltage regulator and reduce the maximum input current during start-up. A voltage ramp at the HVEN pin can be created by driving the pin through an external RC filter (see Figure 6). By choosing a large RC time constant, the peak start-up current will not overshoot the current that is required to regulate the output. Choose the value of the resistor so that it can supply 20µA when the HVEN pin reaches 2.3V. RUN 15k LTC4089-1 HVEN 0.1µF GND 4089-1 F06 Figure 6. Using the HVEN Pin to Soft-Start the High Voltage Regulator. Alternate NTC Thermistors The LTC4089-1 NTC trip points were designed to work with thermistors whose resistance-temperature characteristics follow Vishay Dale’s “R-T Curve 2.” The Vishay NTHS0603N02N1002J is an example of such a thermistor. However, Vishay Dale has many thermistor products that follow the “R-T Curve 2” characteristic in a variety of sizes. Furthermore, any thermistor whose ratio of RCOLD to RHOT is about 6.0 will also work (Vishay Dale R-T Curve 2 shows a ratio of 2.816/0.4839 = 5.82). Power conscious designs may want to use thermistors whose room temperature value is greater than 10k. Vishay Dale has a number of values of thermistor from 10k to 100k that follow the “R-T Curve 1.” Using these as indicated in the NTC Thermistor section will give temperature trip points of approximately 3°C and 42°C, a delta of 39°C. This delta in temperature can be moved in either direction by changing the value of RNOM with respect to RNTC. 40891f 18 LTC4089-1 APPLICATIO S I FOR ATIO U W U U Increasing RNOM will move both trip points to lower temperatures. Likewise, a decrease in RNOM with respect to RNTC will move the trip points to higher temperatures. To calculate RNOM for a shift to lower temperature, for example, use the following equation: RNOM R = COLD • RNTC at 25 °C 2 . 816 where RCOLD is the resistance ratio of RNTC at the desired cold temperature trip point. To shift the trip points to higher temperatures use the following equation: RNOM = RHOT • RNTC at 25 °C 0 . 484 where RHOT is the resistance ratio of RNTC at the desired hot temperature trip point. The following example uses a 100K R-T Curve 1 Thermistor from Vishay Dale. The difference between the trip points is 39°C, from before—and the desired cold trip point of 0°C, would put the hot trip point at about 39°C. The RNOM needed is calculated as follows: R RNOM = COLD • RNTC at 25°C = 2.816 3.266 • 100kΩ = 116kΩ 2.816 The nearest 1% value for RNOM is 115k. This is the value used to bias the NTC thermistor to get cold and hot trip points of approximately 0°C and 39°C, respectively. To extend the delta between the cold and hot trip points, a resistor (R1) can be added in series with RNTC (see Figure 7). The values of the resistors are calculated as follows: RNOM − RHOT R = COLD 2 . 816 − 0 . 484 where RNOM is the value of the bias resistor, RHOT and RCOLD are the values of RNTC at the desired temperature trip points. Continuing the forementioned example with a desired hot trip point of 50°C: RNOM = = R COLD − R HOT 2 . 816 − 0 . 484 100k • (3 . 266 − 0 . 36 0 2) 2 . 816 − 0 . 484 = 124 . 6k,124k nearest 1 % 0 . 484 2 . 816 − 0 . 484 • R1 = 100k • ( 3 . 266 − 0 . 3602) − 0 . 3 6 02 = 24 . 3k The final solution is shown in Figure 7, where RNOM = 124k, R1 = 24.3k and RNTC = 100k at 25°C VNTC LTC4089-1 15 RNOM 124k NTC 0.738 • VNTC – TOO_COLD 14 + R1 24.3k – 0.326 • VNTC RNTC 100k TOO_HOT + + NTC_ENABLE 0 . 484 R1 = • [RCOLD − RHOT ] − RHOT 2 . 816 − 0 . 4 8 4 0.1V – 4089-1 F07 Figure 7. Modified NTC Circuit 40891f 19 LTC4089-1 APPLICATIO S I FOR ATIO U W U U Power Dissipation and High Temperature Considerations The die temperature of the LTC4089-1 must be lower than the maximum rating of 110°C. This is generally not a concern unless the ambient temperature is above 85°C. The total power dissipated inside the LTC4089-1 depends on many factors, including input voltage (IN or HVIN), battery voltage, programmed charge current, programmed input current limit, and load current. In general, if the LTC4089-1 is being powered from IN the power dissipation can be calculated as follows: PD = ( VIN − VBAT ) • IBAT + ( VIN − VOUT ) • IOUT where PD is the power dissipated, IBAT is the battery charge current, and IOUT is the application load current. For a typical application, an example of this calculation would be: PD = (5V − 3 . 7 V) • 0 . 4A + (5V − 4 . 75V) • 0 . 1A = 545mW This example assumes VIN = 5V, VOUT = 4.75V, VBAT = 3.7V, IBAT = 400mA, and IOUT = 100mA resulting in slightly more than 0.5W total dissipation. If the LTC4089-1 is being powered from HVIN, the power dissipation can be estimated by calculating the regulator power loss from an efficiency measurement, and subtracting the catch diode loss. PD = (1 − η) • ( VHVOUT • (IBAT + IOUT )) − VD • VHVOUT 1 − V • (IBAT + IOUT ) + 0 . 3V • IBAT HV IN where is the efficiency of the high voltage regulator and VD is the forward voltage of the catch diode at I = IBAT + IOUT. The first term corresponds to the power lost in converting VHVIN to VHVOUT, the second term subtracts the catch diode loss, and the third term is the power dissipated in the battery charger. For a typical application, an example of this calculation would be: PD = (1− 0 . 87) • [ 4V • (0 . 7A + 0 . 3A)] − 0 . 4V • 4V 1− 12V • (0 . 7A + 0 . 3A) + 0 . 3V • 0 . 7A = 463mW This example assumes 87% efficiency, VHVIN = 12V, VBAT = 3.7V (VHVOUT is about 4V), IBAT = 700mA, IOUT = 300mA resulting in less than 0.5W total dissipation. If the LTC4089-5 is being powered from HVIN, the power dissipation can be estimated by calculating the regulator power loss from an efficiency measurement and subtracting the catch diode loss. PD = (1− η) • (5V • (IBAT + IOUT )) 5V − VD • 1− • (IBAT + IOUT ) VHVIN +(5V − VBAT ) • IBAT The difference between this equation and the LTC4089-1 is the last term which represents the power dissipation in the battery charger. For a typical application, an example of this calculation would be: PD = (1− 0.87) • (5V • (0.7 A + 0.3A)) 5V −0.4V • (1− ) • (0.7 A + 0.3A) 12V +(5V − 3.7 V) • 0.7 A = 1327 , mW Like the LTC4089-1 example, this example assumes 87% efficiency, VHVIN = 12V, VBAT = 3.7V, IBAT = 700mA, IOUT = 300mA resulting in 1.3W total dissipation. To prevent power dissipation of this magnitude from causing high die temperature, it is important to solder the exposed backside of the package to a ground plane. This ground should be tied to other copper layers below with thermal vias; these layers will spread the heat dissipated by the LTC4089-1. Additional vias should be placed near the catch diodes. Adding more copper to the top and bottom layers, and tying this copper to the internal planes with vias, can reduce thermal resistance further. With these steps, the thermal resistance from die (i.e., junction) to ambient can be reduced to JA = 40°C/W. 40891f 20 LTC4089-1 APPLICATIO S I FOR ATIO U W U U The power dissipation in the other power components—catch diodes, MOSFETs, boost diodes and inductors—causes additional copper heating and can further increase the “ambient” temperature of the IC. Board Layout Considerations As discussed in the previous section, it is critical that the exposed metal pad on the backside of the LTC4089-1 package be soldered to the PC board ground. Furthermore, proper operation and minimum EMI requires a careful printed circuit board (PCB) layout. Note that large, switched currents flow in the power switch (between the HVIN and SW pins), the catch diode and the HVIN input capacitor. These components, along with the inductor and output capacitor, should be placed on the same side of the circuit board, and their connections should be made on that layer. Place a local, unbroken ground plane below these components. The loop formed by these components should be as small as possible. Additionally, the SW and BOOST nodes should be kept as small as possible. Figure 8 shows the recommended component placement with trace and via locations. High frequency currents, such as the high voltage input current of the LTC4089-1, tend to find their way along the ground plane on a mirror path directly beneath the incident path on the top of the board. If there are slits or cuts in the ground plane due to other traces on that layer, the current will be forced to go around the slits. If high frequency currents are not allowed to flow back through their natural least-area path, excessive voltage will build up and radiated emissions will occur. See Figure 9. 4089 F09 Figure 9. Ground Currents Follow Their Incident Path at High Speed. Slices in the Ground Plane Cause High Voltage and Increased Emissions. VIN and VHVIN Bypass Capacitor C1 AND D1 GND PADS SIDE-BY-SIDE AND SEPERATED WITH C3 GND PAD MINIMIZE D1, L1, C3, U1, SW PIN LOOP Many types of capacitors can be used for input bypassing, however, caution must be exercised when using multilayer ceramic capacitors. Because of the self-resonant and high Q characteristics of some types of ceramic capacitors, high voltage transients can be generated under some start-up conditions, such as from connecting the charger input to a hot power source. For more information, refer to Application Note 88. Battery Charger Stability Considerations U1 THERMAL PAD SOLDERED TO PCB. VIAS CONNECTED TO ALL GND PLANES WITHOUT THERMAL RELIEF MINIMIZE TRACE LENGTH 4089-1 F08 The constant-voltage mode feedback loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1µF from BAT to GND. Furthermore, a 4.7µF capacitor with a 0.2 to 1 series resistor to GND is recommended at the BAT pin to keep ripple voltage low when the battery is disconnected. Figure 8. Suggested Board Layout 40891f 21 LTC4089-1 APPLICATIO S I FOR ATIO U U W U D2 SD101AWS VIN 6V TO 36V VIN E1 + E2 GND 21 C9 22 F 50V R1 1M 1% C1 1 F 50V 20 HVIN BOOST SW 19 L1 C2 10 H 0.1 F 6.3V SLF6028T-100M1R3 E16 HVOUT C3 22 F 6.3V D1 DLFS160 JP1 VIN 1 ON 22 2 C7 1000pF 50V 3 OFF USB E3 4.35V TO 5.5V JP2 CURRENT USB 500mA 100mA 1 2 3 E8 HPWR JP3 USB ON/OFF 1 OFF 2 ON 3 15 16 C4 0.1 F 10% R3 2.1k 1% R4 71.5k 1% HVOUT HVEN HVOUT 12 C5 4.7 F 6.3V R2 1 LTC4089-1 17 HVPR IN OUT HPWR SUSP GATE BAT TIMER CHRG 14 9 CLPROG VNTC NTC PROG VC 4 10pF GND GND 2 1 3 R7 680 18 7 13 Q1 Si2333DS R6 1k 1% 10 D3 HVPR RED E4 OUT C6 4.7 F 6.3V Q2 Si2333DS GND 11 8 R8 680 6 5 D4 CHGR GRN E6 LI-ION+ C8 4.7 F 6.3V R9 1 E7 GND R5 10k 1% E9 CHGR E11 NTC JP4 NTC 4089-1 F10 1 E13 SUSP 2 E10 CLPROG 3 R10 10k 1% E12 PROG EXT INT 4089-1 TA02 Figure 10. Typical Application Diagram 40891f 22 LTC4089-1 PACKAGE DESCRIPTIO U DJC Package 22-Lead Plastic DFN (6mm 3mm) (Reference LTC DWG # 05-08-1714) 0.889 0.70 ±0.05 R = 0.10 0.889 3.60 ±0.05 1.65 ±0.05 2.20 ±0.05 (2 SIDES) PACKAGE OUTLINE 0.25 ± 0.05 0.50 BSC 5.35 ± 0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS NOTE: 1. DIMENSIONS ARE IN MILLIMETERS 2. APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3. DRAWING IS NOT TO SCALE 6.00 ±0.10 (2 SIDES) 0.889 R = 0.10 TYP PIN 1 TOP MARK (NOTE 6) 0.200 REF 3.00 ±0.10 (2 SIDES) R = 0.115 TYP 22 0.889 1.65 ± 0.10 (2 SIDES) 0.75 ±0.05 0.00 – 0.05 0.40 ± 0.05 12 11 5.35 ± 0.10 (2 SIDES) 1 0.25 ± 0.05 0.50 BSC PIN #1 NOTCH R0.30 TYP OR 0.25mm × 45° CHAMFER (DJC) DFN 0605 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WXXX) IN JEDEC PACKAGE OUTLINE M0-229 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE 40891f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 23 LTC4089-1 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1733 Monolithic Lithium-Ion Linear Battery Charger Standalone Charger with Programmable Timer, Up to 1.5A Charge Current LTC1734 Lithium-Ion Linear Battery Charger in ThinSOTTM Simple ThinSOT Charger, No Blocking Diode, No Sense Resistor Needed LTC4002 Switch Mode Lithium-Ion Battery Charger Standalone, 4.7V ≤ VIN ≤ 24V, 500kHz Frequency, 3 Hour Charge Termination LTC4053 USB Compatible Monolithic Li-Ion Battery Charger Standalone Charger with Programmable Timer, Up to 1.25A Charge Current LTC4054 Standalone Linear Li-Ion Battery Charger with Integrated Pass Transistor in ThinSOT Thermal Regulation Prevents Overheating, C/10 Termination, C/10 Indicator, Up to 800mA Charge Current LTC4057 Lithium-Ion Linear Battery Charger Up to 800mA Charge Current, Thermal Regulation, ThinSOT Package LTC4058 Standalone 950mA Lithium-Ion Charger C/10 Charge Termination, Battery Kelvin Sensing, ±7% Charge Accuracy in DFN LTC4059 900mA Linear Lithium-Ion Battery Charger Battery Chargers 2mm 2mm DFN Package, Thermal Regulation, Charge Current Monitor Output LTC4065/LTC4065A Standalone Li-Ion Battery Chargers in 2 2 DFN 4.2V, ±0.6% Float Voltage, Up to 750mA Charge Current, 2mm 2mm DFN, “A” Version has ACPR Function. LTC4411/LTC4412 Low Loss PowerPath Controller in ThinSOT Automatic Switching Between DC Sources, Load Sharing, Replaces ORing Diode LTC4412HV High Voltage Power Path Controllers in ThinSOT VIN = 3V to 36V, More Efficient than Diode ORing, Automatic Switching Between DC Sources, Simplified Load Sharing, ThinSOT Package. Power Management LTC3405/LTC3405A 300mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.7V to 6V, VOUT = 0.8V, IQ = 20µA, ISD < 1µA, ThinSOT Package LTC3406/LTC3406A 600mA (IOUT), 1.5MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.6V, IQ = 20µA, ISD < 1µA, ThinSOT Package LTC3411 1.25A (IOUT), 4MHz, Synchronous Step-Down DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 0.8V, IQ = 60µA, ISD < 1µA, MS10 Package LTC3440 600mA (IOUT), 2MHz, Synchronous Buck-Boost DC/DC Converter 95% Efficiency, VIN = 2.5V to 5.5V, VOUT = 2.5V, IQ = 25µA, ISD < 1µA, MS Package LTC3455 Dual DC/DC Converter with USB Power Manager and Li-Ion Battery Charger Seamless Transition Between Power Sources: USB, Wall Adapter and Battery; 95% Efficient DC/DC Conversion LT3493 1.2A, 750kHz Step-Down Switching Regulator 88% Efficiency, VIN = 3.6V to 36V (40V Maximum), VOUT = 0.8V, ISD < 2µA, 2mm 3mm DFN Package LTC4055 USB Power Controller and Battery Charger Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 200m Ideal Diode, 4mm 4mm QFN16 Package LTC4066 USB Power Controller and Li-Ion Battery Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 50m Charger with Low-Loss Ideal Diode Ideal Diode, 4mm 4mm QFN24 Package LTC4085 USB Power Manager with Ideal Diode Controller and Li-Ion Charger Charges Single Cell Li-Ion Batteries Directly from a USB Port, Thermal Regulation, 200m Ideal Diode with <50m Option, 4mm 3mm DFN14 Package ThinSOT is a trademark of Linear Technology Corporation. 40891f 24 Linear Technology Corporation LT 1106 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com LINEAR TECHNOLOGY CORPORATION 2006